UCC28070PWR - Texas Instruments

+–

VIN VOUT

12V to 21V

L2 D2

COUT

CPC CPC

CZC CZC

RZC RZC

CSS

RRDM

RRT

CCDR

RDMX

RSYN

CREF

RPK1

RPK2

CZV

RZV

CPV

CAOA

CAOBPKLMT

GND

VAO

VINAC

VSENSE

CSA

CSB

CDR

GDB

GDA

IMO VCC

RSYNTH

VREF

DMAX

RDM

RIMO

To CSB

To CSA

From Ixfrms

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

Interleaving Continuous Conduction Mode PFC Controller

Check for Samples: UCC28070

1FEATURES APPLICATIONS

•Interleaved Average Current-Mode PWM •High-Efficiency Server and Desktop Power

Control with Inherent Current Matching Supplies

•Advanced Current Synthesizer Current •Telecom Rectifiers

Sensing for Superior Efficiency •White Goods and Industrial Equipment

•Highly-Linear Multiplier Output with Internal

Quantized Voltage Feed-Forward Correction DESCRIPTION

for Near-Unity PF The UCC28070 is an advanced power factor

correction device that integrates two pulse-width

•Programmable Frequency (30 kHz to 300 kHz) modulators (PWMs) operating 180°out of phase.

•Programmable Maximum Duty-Cycle Clamp This interleaved PWM operation generates

•Programmable Frequency Dithering Rate and substantial reduction in the input and output ripple

Magnitude for Enhanced EMI Reduction currents, and the conducted-EMI filtering becomes

easier and less expensive. A significantly improved

–Magnitude: 3 kHz to 30 kHz multiplier design provides a shared current reference

–Rate: Up to 30 kHz to two independent current amplifiers that ensures

•External Clock Synchronization Capability matched average current mode control in both PWM

outputs while maintaining a stable, low-distortion

•Enhanced Load and Line Transient Response sinusoidal input line current.

through Voltage Amplifier Output Slew-Rate

Correction The UCC28070 contains multiple innovations

including current synthesis and quantized voltage

•Programmable Peak Current Limiting feed-forward to promote performance enhancements

•Bias-Supply UVLO, Over-Voltage Protection, in PF, efficiency, THD, and transient response.

Open-Loop Detection, and PFC-Enable Features including frequency dithering, clock

Monitoring synchronization, and slew rate enhancement further

•External PFC-Disable Interface expand the potential performance enhancements.

•Open-Circuit Protection on VSENSE and The UCC28070 also contains a variety of protection

VINAC pins features including output over-voltage detection,

programmable peak-current limit, under-voltage

•Programmable Soft Start lockout, and open-loop protection.

•20-Lead TSSOP/SOIC Packages

Simplified Application Diagram

1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas

Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

Products conform to specifications per the terms of the Texas

Instruments standard warranty. Production processing does not

necessarily include testing of all parameters.

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

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ORDERING INFORMATION

PART NUMBER PACKAGE PACKING

UCC28070PW Plastic, 20-Pin TSSOP (PW) 70-Pc. Tube

UCC28070PWR Plastic, 20-Pin TSSOP (PW) 2000-Pc. Tape and Reel

UCC28070DW Plastic, 20-Pin SOIC (DW) 25-Pc. Tube

UCC28070DWR Plastic, 20-Pin SOIC (DW) 2000-Pc. Tape and Reel

ABSOLUTE MAXIMUM RATINGS(1) (2) (3) (4)

over operating free-air temperature range (unless otherwise noted)

PARAMETER LIMIT UNIT

Supply voltage: VCC 22 V

Supply current: IVCC 20 mA

Voltage: GDA, GDB −0.5 to VCC+0.3 V

Gate drive current –continuous: GDA, GDB +/−0.25 A

Gate drive current –pulsed: GDA, GDB +/−0.75

Voltage: DMAX, RDM, RT, CDR, VINAC, VSENSE, SS, VAO, IMO, CSA, CSB, −0.5 to +7 V

CAOA, CAOB, PKLMT, VREF

Current: RT, DMAX, RDM, RSYNTH −0.5 mA

Current: VREF, VAO, CAOA, CAOB, IMO 10

Operating junction temperature, TJ−40 to +125

Storage temperature, TSTG −65 to +150 °C

Lead temperature (10 seconds) 260

(1) These are stress limits. Stress beyond these limits may cause permanent damage to the device. Functional operation of the device at

these or any conditions beyond those indicated under RECOMMENDED OPERATING CONDITIONS is not implied. Exposure to

absolute maximum rated conditions for extended periods of time may affect device reliability.

(2) All voltages are with respect to GND.

(3) All currents are positive into the terminal, negative out of the terminal.

(4) In normal use, terminals GDA and GDB are connected to an external gate driver and are internally limited in output current.

UCC28070

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ELECTROSTATIC DISCHARGE (ESD) PROTECTIONRATING UNIT

Human Body Model (HBM) 2,000 V

Charged Device Model (CDM) 500

DISSIPATION RATINGS THERMAL IMPEDANCE TA= 25°C POWER

PACKAGE TA= 85°C POWER RATING

JUNCTION-TO-AMBIENT RATING

20-Pin TSSOP 125 °C/Watt (1) and (2) 800 mW (1) 320 mW (1)

20-Pin SOIC 95 °C/Watt (1) and (2) 1050 mW (1) 420 mW (1)

(1) Thermal resistance is a strong function of board construction and layout. Air flow reduces thermal resistance. This number is only a

general guide.

(2) Thermal resistance calculated with a low-K methodology.

RECOMMENDED OPERATING CONDITIONS

over operating free-air temperature range (unless otherwise noted)

PARAMETER MIN MAX UNIT

VCC Input Voltage (from a low-impedance source) VUVLO + 1 V 21 V

VREF Load Current 2 mA

VINAC Input Voltage Range 0 3

IMO Voltage Range 0 3.3 V

PKLMT, CSA, &CSB Voltage Range 0 3.6

RSYNTH Resistance (RSYN) 15 750 kΩ

RDM Resistance (RRDM) 30 330

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

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ELECTRICAL CHARACTERISTICS

over operating free-air temperature range −40°C<TA<125°C, TJ= TA, VCC = 12 V, GND = 0 V, RRT = 75 kΩ, RDMX = 68.1

kΩ, RRDM = RSYN = 100 kΩ, CCDR = 2.2 nF, CSS = CVREF = 0.1 μF, CVCC = 1 μF, IVREF = 0 mA (unless otherwise noted)

SYMBOL PARAMETER TEST CONDITIONS MIN TYP MAX UNITS

Bias Supply

VCCSHUNT VCC shunt voltage (1) IVCC = 10 mA 23 25 27 V

VCC current, disabled VSENSE = 0 V 7 mA

VCC current, enabled VSENSE = 3 V (switching) 9 12

VCC = 7 V 200 μA

VCC current, UVLO VCC = 9 V 4 6 mA

VUVLO UVLO turn-on threshold Measured at VCC (rising) 9.8 10.2 10.6

UVLO hysteresis Measured at VCC (falling) 1 V

VREF enable threshold Measured at VCC (rising) 7.5 8 8.5

Linear Regulator

VREF voltage, no load IVREF = 0 mA 5.82 6 6.18 V

Measured as the change in VREF,

VREF load rejection -12 12

(IVREF = 0 mA and −2 mA) mV

Measured as the change in VREF,

VREF line rejection -12 12

(VCC = 11V and 20 V, IVREF = 0 μA)

PFC Enable

VEN Enable threshold Measured at VSENSE (rising) 0.65 0.75 0.85 V

Enable hysteresis 0.15

External PFC Disable

Disable threshold Measured at SS (falling) 0.5 0.6 V

Hysteresis VSENSE >0.85 V 0.15

Oscillator

Output phase shift Measured between GDA and GDB 179 180 181 Degree

VDMAX,VRT,Timing regulation voltages Measured at DMAX, RT, &RDM 2.91 3 3.09 V

and VRDM RRT = 75 kΩ, RDMX = 68.1 kΩ,95 100 105

VRDM = 0 V, VCDR = 6 V

fPWM PWM switching frequency kHz

RRT = 24.9 kΩ, RDMX = 22.6 kΩ,270 290 330

VRDM = 0 V, VCDR = 6 V

RRT = 75 kΩ, RDMX = 68.1 kΩ,

DMAX Duty-cycle clamp 92% 95% 98%

VRDM = 0 V, VCDR = 6 V

RRT = 24.9 kΩ, RDMX = 22.6 kΩ,

Minimum programmable off-time 50 150 250 ns

VRDM = 0 V, VCDR = 6 V

RRDM = 316 kΩ, RRT = 75 kΩ234

Frequency dithering magnitude change

fDM in fPWM RRDM = 31.6 kΩ, RRT = 24.9 kΩ24 30 36 kHz

CCDR = 2.2 nF, RRDM = 100 kΩ3

Frequency dithering rate rate of

fDR change in fPWM CCDR = 0.3 nF, RRDM = 100 kΩ20

Dither rate current Measure at CDR (sink and source) ±10 μA

ICDR Dither disable threshold Measured at CCDR (rising) 5 5.25 V

(1) Excessive VCC input voltage and/or current damages the device. This clamp will not protect the device from an unregulated supply. If

an unregulated supply is used, a series-connected fixed positive voltage regulator such as a UA78L15A is recommended. See the

Absolute Maximum Ratings section for the limits on VCC voltage and current.

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

ELECTRICAL CHARACTERISTICS (continued)

over operating free-air temperature range −40°C<TA<125°C, TJ= TA, VCC = 12 V, GND = 0 V, RRT = 75 kΩ, RDMX = 68.1

kΩ, RRDM = RSYN = 100 kΩ, CCDR = 2.2 nF, CSS = CVREF = 0.1 μF, CVCC = 1 μF, IVREF = 0 mA (unless otherwise noted)

SYMBOL PARAMETER TEST CONDITIONS MIN TYP MAX UNITS

Clock Synchronization

VCDR SYNC enable threshold Measured at CDR (rising) 5 5.25 V

VCDR = 6 V, Measured from RDM (rising) to

SYNC propagation delay 50 100 ns

GDx (rising)

SYNC threshold (Rising) VCDR = 6 V, Measured at RDM 1.2 1.5 V

SYNC threshold (Falling) VCDR = 6 V, Measured at RDM 0.4 0.7

Positive pulse width 0.2 μs

SYNC pulses Maximum duty cycle (2) 50 %

Voltage Amplifier

VSENSE voltage In regulation, TA= 25°C 2.97 3 3.03 V

VSENSE voltage In regulation 2.94 3 3.06

VSENSE input bias current In regulation 250 500 nA

VAO high voltage VSENSE = 2.9 V 4.8 5 5.2 V

VAO low voltage VSENSE = 3.1 V 0.05 0.50

gMV VAO transconductance 2.8 V <VSENSE <3.2 V, VAO = 3 V 70 μS

VAO sink current, overdriven limit VSENSE = 3.5 V, VAO = 3 V 30

VAO source current, overdriven VSENSE = 2.5 V, VAO = 3 V, SS = 3 V −30 μA

VAO source current, VSENSE = 2.5 V, VAO = 3 V −130

overdriven limit + ISRC Measured as VSENSE (falling) / VSENSE

Slew-rate correction threshold 92 93 95 %

(regulation)

Slew-rate correction hysteresis Measured at VSENSE (rising) 3 9 mV

Measured at VAO, in addition to VAO

ISRC Slew-rate correction current −100 μA

source current.

Slew-rate correction enable threshold Measured at SS (rising) 4 V

VAO discharge current VSENSE = 0.5 V, VAO = 1 V 10 μA

Soft Start

ISS SS source current VSENSE = 0.9 V, SS = 1 V −10 μA

Adaptive source current VSENSE = 2.0 V, SS = 1 V −1.5 -2.5 mA

Adaptive SS disable Measured as VSENSE –SS -30 0 30 mV

SS sink current VSENSE = 0.5 V, SS = 0.2 V 0.5 0.9 mA

(2) Due to the influence of the synchronization pulse width on the programmability of the maximum PWM switching duty cycle (DMAX) it is

recommended to minimize the synchronization signal's duty cycle.

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

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ELECTRICAL CHARACTERISTICS (continued)

over operating free-air temperature range −40°C<TA<125°C, TJ= TA, VCC = 12 V, GND = 0 V, RRT = 75 kΩ, RDMX = 68.1

kΩ, RRDM = RSYN = 100 kΩ, CCDR = 2.2 nF, CSS = CVREF = 0.1 μF, CVCC = 1 μF, IVREF = 0 mA (unless otherwise noted)

SYMBOL PARAMETER TEST CONDITIONS MIN TYP MAX UNITS

Over Voltage

Measured as VSENSE (rising) / VSENSE

VOVP OVP threshold 104 106 108 %

(regulation)

OVP hysteresis Measured at VSENSE (falling) 100 mV

Measured between VSENSE (rising) and

OVP propagation delay 0.2 0.3 μs

GDx (falling)

Zero-Power

VZPWR Zero-power detect threshold Measured at VAO (falling) 0.65 0.75 V

Zero-power hysteresis 0.15

Multiplier

VAO ≥1.5 V, TA= 25°C 16 17 18

VAO = 1.2 V, TA= 25°C 14.5 17.0 19.5

kMULT Gain constant VAO ≥1.5 V 15 17 19 μA

VAO = 1.2 V 13 17 21

VINAC = 0.9 VPK, VAO = 0.8 V -0.2 0 0.2

IIMO Output current: zero VINAC = 0 V, VAO = 5 V -0.2 0 0.2

Quantized Voltage Feed Forward

VLVL1 Level 1 threshold (3) 0.6 0.7 0.8

VLVL2 Level 2 threshold 1

VLVL3 Level 3 threshold 1.2

VLVL4 Level 4 threshold 1.4

Measured at VINAC (rising) V

VLVL5 Level 5 threshold 1.65

VLVL6 Level 6 threshold 1.95

VLVL7 Level 7 threshold 2.25

VLVL8 Level 8 threshold 2.6

Current Amplifiers

CAOx high voltage 5.75 6 V

CAOx low voltage 0.1

gMC CAOx transconductance 100 μS

CAOx sink current, overdriven 50 μA

CAOx source current, overdriven −50

Input common mode range 0 3.6 V

RSYNTH = 6 V, TA= 25°C -4 -8 -13

Input offset Voltage mV

RSYNTH = 6 V 0 -8 -20

Input offset voltage 0 −8−20 mV

Measured as Phase A’s input offset minus

Phase mismatch -12 0 12

Phase B’s input offset

CAOx pull-down current VSENSE = 0.5 V, CAOx = 0.2 V 0.5 0.9 mA

(3) The Level 1 threshold represents the “zero-crossing detection”threshold above which VINAC must rise to initiate a new input half-cycle,

and below which VINAC must fall to terminate that half-cycle.

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

ELECTRICAL CHARACTERISTICS (continued)

over operating free-air temperature range −40°C<TA<125°C, TJ= TA, VCC = 12 V, GND = 0 V, RRT = 75 kΩ, RDMX = 68.1

kΩ, RRDM = RSYN = 100 kΩ, CCDR = 2.2 nF, CSS = CVREF = 0.1 μF, CVCC = 1 μF, IVREF = 0 mA (unless otherwise noted)

SYMBOL PARAMETER TEST CONDITIONS MIN TYP MAX UNITS

Current Synthesizer

VSENSE = 3 V, VINAC = 0 V 2.91 3 3.09

VRSYNTH Regulation voltage VSENSE = 3 V, VINAC = 2.85 V 0.10 0.15 0.20 V

Synthesizer disable threshold Measured at RSYNTH (rising) 5 5.25

VINAC input bias current 0.250 0.500 μA

Peak Current Limit

Peak current limit threshold PKLMT = 3.30 V, measured at CSx (rising) 3.27 3.3 3.33 V

Measured between CSx (rising) and GDx

Peak current limit propagation delay 60 100 ns

(falling) edges

PWM Ramp

VRMP PWM ramp amplitude 3.8 4.0 4.2 V

PWM ramp offset voltage TA= 25°C, RRT = 75 kΩ0.65 0.7

PWM ramp offset temperature −2 mV/ °C

coefficient

Gate Drive

GDA, GDB output voltage, high, VCC = 20 V, CLOAD = 1 nF 11.5 13 15

clamped V

GDA, GDB output voltage, High CLOAD = 1 nF 10 10.5

GDA, GDB output voltage, Low CLOAD = 1 nF 0.2 0.3

Rise time GDx 1 V to 9 V, CLOAD = 1 nF 18 30 ns

Fall time GDx 9 V to 1 V, CLOAD = 1 nF 12 25

GDA, GDB output voltage, UVLO VCC = 0 V, IGDA, IGDB = 2.5 mA 0.7 2 V

Thermal Shutdown

Thermal shutdown threshold 160 °C

Thermal shutdown recovery 140

GDB

CAOB

GND

VCC

GDA

DMAX

VAO

RDM

PKLMT

RSYNTH

CSA

VSENSE

VINAC

IMO

CDR

CSB VREF

CAOA

GDB

GND

VCC

GDA

VREF

CAOA

CAOB

VAO

VSENSE

VINAC

IMO

RSYNTH

CSB

CSA

PKLMT

DMAX

CDR

RDM

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

www.ti.com

DEVICE INFORMATION

SOIC-20 Top View, DW Package

TSSOP-20 Top View, PW Package

TERMINAL FUNCTIONS

NAME PIN # I/O DESCRIPTION

Dither Rate Capacitor. Frequency-dithering timing pin. An external capacitor to GND programs

CDR 1 I the rate of oscillator dither. Connect the CDR pin to the VREF pin to disable dithering.

Dither Magnitude Resistor. Frequency-dithering magnitude and external synchronization pin. An

external resistor to GND programs the magnitude of oscillator frequency dither. When frequency

RDM 2 I dithering is disabled (CDR >5 V), the internal master clock will synchronize to positive edges

(SYNC) presented on the RDM pin. Connect RDM to GND when dithering is disabled and synchronization

is not desired.

Voltage Amplifier Output. Output of transconductance voltage error amplifier. Internally

VAO 3 O connected to Multiplier input and Zero-Power comparator. Connect the voltage regulation loop

compensation components between this pin and GND.

Output Voltage Sense. Internally connected to the inverting input of the transconductance

voltage error amplifier in addition to the positive terminal of the Current Synthesis difference

VSENSE 4 I amplifier. Also connected to the OVP, PFC Enable, and slew-rate comparators. Connect to PFC

output with a resistor-divider network.

Scaled AC Line Input Voltage. Internally connected to the Multiplier and negative terminal of the

VINAC 5 I Current Synthesis difference amplifier. Connect a resistor-divider network between VIN, VINAC,

and GND identical to the PFC output divider network connected at VSENSE.

Multiplier Current Output. Connect a resistor between this pin and GND to set the multiplier

IMO 6 O gain.

Current Synthesis Down-Slope Programming. Connect a resistor between this pin and GND to

RSYNTH 7 I set the magnitude of the current synthesizer down-slope. Connecting RSYNTH to VREF will

disable current synthesis and connect CSA and CSB directly to their respective current amplifiers.

Phase B Current Sense Input. During the on-time of GDB, CSB is internally connected to the

CSB 8 I inverting input of Phase B’s current amplifier through the current synthesis stage.

Phase A Current Sense Input. During the on-time of GDA, CSA is internally connected to the

CSA 9 I inverting input of Phase A’s current amplifier through the current synthesis stage.

Peak Current Limit Programming. Connect a resistor-divider network between VREF and this

PKLMT 10 I pin to set the voltage threshold of the cycle-by-cycle peak current limiting comparators. Allows

adjustment for desired ΔILB.

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

TERMINAL FUNCTIONS (continued)

NAME PIN # I/O DESCRIPTION

Phase B Current Amplifier Output. Output of phase B’s transconductance current amplifier.

Internally connected to the inverting input of phase B’s PWM comparator for trailing-edge

CAOB 11 O modulation. Connect the current regulation loop compensation components between this pin and

GND.

Phase A Current Amplifier Output. Output of phase A’s transconductance current amplifier.

Internally connected to the inverting input of phase A’s PWM comparator for trailing-edge

CAOA 12 O modulation. Connect the current regulation loop compensation components between this pin and

GND.

6-V Reference Voltage and Internal Bias Voltage. Connect a 0.1-μF ceramic bypass capacitor

VREF 13 O as close as possible to this pin and GND.

Phase A’s Gate Drive. This limited-current output is intended to connect to a separate gate-drive

GDA 14 O device suitable for driving the Phase A switching component(s). The output voltage is typically

clamped to 13.5 V.

Bias Voltage Input. Connect a 0.1-μF ceramic bypass capacitor as close as possible to this pin

VCC 15 I and GND.

Device Ground Reference. Connect all compensation and programming resistor and capacitor

GND 16 I/O networks to this pin. Connect this pin to the system through a separate trace for high-current

noise isolation.

Phase B’s Gate Drive. This limited-current output is intended to connect to a separate

GDB 17 O gate-drivedevice suitable for driving the Phase B switching component(s). The output voltage is

typically clamped to 13.5 V.

Soft-Start and External Fault Interface. Connect a capacitor to GND on this pin to set the

soft-start slew rate based on an internally-fixed 10-μA current source. The regulation reference

voltage for VSENSE is clamped to VSS until VSS exceeds 3 V. Upon recovery from certain fault

SS 18 I conditions a 1-mA current source is present at the SS pin until the SS voltage equals the

VSENSE voltage. Pulling the SS pin below 0.6 V immediately disables both GDA and GDB

outputs.

Timing Resistor. Oscillator frequency programming pin. A resistor to GND sets the running

RT 19 I frequency of the internal oscillator.

Maximum Duty-Cycle Resistor. Maximum PWM duty-cycle programming pin. A resistor to GND

DMAX 20 I sets the PWM maximum duty-cycle based on the ratio of RDMX/RRT.

14 GDA

VCC

17 GDB

+S Q

PWM1

+S Q

PWM2

CLKB

OffB

IpeakB

12CAOA

11CAOB

Driver

8CSB

9CSA

CA2

CA1

CLKA

OffA

IpeakA

GmAmp

IpeakA

IpeakB

GND

PKLMT

RSYNTH

Current

Synthesizer

VINAC

VSENSE

OutA

OutB

GND

Fault

(Clamped at 13.5V)

VCC

(Clamped at 13.5V)

Fault

VSENSE

3 VAO

18 SS

ISS

Mult.

GmAmp -

+2.8V

VINAC5

6 IMO

Slew Rate

Correction

10uA

100uA

250nA

ReStart

+SS

IMO=

VVINAC * (VVAO – 1)

KVFF

* 17uA

Voltage

Feed-

Forward

KVFF

ReStart

10uA

Disable

ReStart

Ext.Disable

1mA

Adaptive SS

Control

Logic

DMAX

RDM/

SYNC

CLKA

CLKB

OffA

OffB

SYNC

Logic

CDR

Oscillator w/

Freq. Dither

SYNC

Enable

Dither

Disable

15VCC

10.2V

16GND

13VREF 6V Linear

Regulator

25V

9.2V

UVLO

S Q

0.75V

ReStart

oThermSD

160 On

140 Off

0.60V

3.18V

ZeroPwr 0.75V

VSENSE

0.75V

VAO

OVP

Ext.Disable 0.60V

3.08V

0.90V

Fault

250nA

VSENSE

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

www.ti.com

Functional Block Diagram

-60 -40 -20 0 60 80 120 140

TJ-Temperature- 0C

20 40 100

IVCC - Supply Current - mA

SUPPLY CURRENT

TEMPERATURE

IVCC,VCC=12V,enabled

IVCC,VCC=12V,disabled

-60 -10 90 140

TJ-Temperature- 0C

5.82

5.88

6.18

5.94

6.00

6.06

6.12

VREF - Reference Voltage - V

REFERENCEVOLTAGE

TEMPERATURE

VREF(IVREF =0mA)

-60 -10 90 140

TJ-Temperature- 0C

0.10

0.50

0.15

0.25

0.35

0.40

IVSENSE - Bias Current - mA

IVSENSE BIASCURRENT

TEMPERATURE

0.05

0.20

0.30

0.45

-60 -20 100 140

TJ- Temperature - 0C

2.94

2.96

3.06

2.98

3.00

3.02

3.04

VSENSE Regulation - V

VSENSE REGULATION

TEMPERATURE

-40 0 20 60 80 120

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

TYPICAL CHARACTERISTICS

Figure 1. Figure 2.

Figure 3. Figure 4.

0 1 5 6

VAO -Voltage AmplifierOutput-V

180

120

140

IMO - Multiplier Output Current - mA

MULTIPLIEROUTPUTCURRENT

VOLTAGE AMPLIFIEROUTPUT

100

160 Level1

Level2

Level3

Level4

Level5

Level6

Level7

Level8

QVFFLevel

-60 -20 100 140

TJ- Temperature - 0C

Multiplier Constant - mA

MULTIPIER CONSTANT

TEMPERATURE

1208060200-40

VAO = 1.5 V

VAO = 5.0 VVAO = 3.0 V

VAO = 1.2 V

-60 80 140

TJ- Temperature - 0C

-1.0

1.0

-0.5

Normalized Change in Switching Frequency - %

SWITCHING FREQUENCY (normalized change)

TEMPERATURE

0.8

-20

-0.8

-0.3

0.3

0.5

Typical Frequency= 30 kHz

RT = 249 k?

Typical Frequency= 100 kHz

RT = 75 k?

Typical Frequency= 290 kHz

RT = 24.9 k?

12010060200-40

-60 90 140

TJ-Temperature- 0C

0.00

0.10

0.50

0.15

0.25

0.35

IVINAC - Bias Current - mA

IVINAC BIASCURRENT

TEMPERATURE

0.45

-10

VINAC=0.2V

VINAC=2.85V

VINAC=2.5V

VINAC=1.0V

VINAC=2.0V

0.05

0.20

0.30

0.40

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

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TYPICAL CHARACTERISTICS (continued)

Figure 5. Figure 6.

Figure 7. Figure 8.

2.5 2.7 3.3 3.5

VSENSE - V

-140

-100

3.0

-80

-40

-20

IVAO - Voltage Amplifier Output Current - mA

VOLTAGE AMPLIFIERTRANSFERFUNCTION

VSENSE

2.6 2.8 2.9 3.1 3.2 3.4

-120

-60

-60 -20 100 140

TJ- Temperature - 0C

VAO - Voltage Amplifier Transconductance - mS

VOLTAGE AMPLIFIER TRANSCONDUCTANCE

TEMPERATURE

-40 0 20 60 80 120

-60 -20 100 140

TJ- Temperature - 0C

110

100

105

CAOx Transconductance - mS

CURRENT AMPLIFIER TRANSCONDUCTANCE

TEMPERATURE

-40 0 20 60 80 120

UCC28070

www.ti.com

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

TYPICAL CHARACTERISTICS (continued)

Figure 9. Figure 10.

Figure 11.

-60 -20 100 140

TJ-Temperature- 0C

-20

-15

-10

-5

CAx Input Offset - mV

CAxINPUTOFFSETVOLTAGE

TEMPERATURE(at0.8Vcommonmode)

-40 0 20 60 80 120

CAx-3s

CAx+3s

CAx AVG

-60 -20 100 140

TJ-Temperature- 0C

-15

-10

CA1 to CA2 Relative Offset Voltage - mV

CA1TOCA2RELATIVEOFFSET

TEMPERATURE(at0.8Vcommonmode)

-40 0 20 60 80 120

A-B-3s

A-B+3s

A-B AVG

-5

-60 -20 100 140

TJ-Temperature- 0C

-15

-10

CAx Input Offset - mV

CAxINPUTOFFSETVOLTAGE

TEMPERATURE(at2.0Vcommonmode)

-40 0 20 60 80 120

-5

CAx-3s

CAx+3s

CAx AVG

-20

-60 -20 100 140

TJ-Temperature- 0C

-15

-10

CA1 to CA2 Relative Offset Voltage - mV

CA1TOCA2RELATIVEOFFSET

TEMPERATURE(at2.0Vcommonmode)

-40 0 20 60 80 120

-5

A-B-3s

A-B+3s

A-B AVG

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

www.ti.com

TYPICAL CHARACTERISTICS (continued)

Figure 12. Figure 13.

Figure 14. Figure 15.

-60 -20 100 140

TJ-Temperature- 0C

-15

-10

CAx Input Offset - mV

CAxINPUTOFFSETVOLTAGE

TEMPERATURE(at3.6Vcommonmode)

-40 0 20 60 80 120

-5

-20

CAx-3s

CAx+3s

CAx AVG

-60 -20 100 140

TJ-Temperature- 0C

-15

-10

CA1 to CA2 Relative Offset Voltage - mV

CA1TOCA2RELATIVEOFFSET

TEMPERATURE(at3.6Vcommonmode)

-40 0 20 60 80 120

-5

A-B-3s

A-B+3s

A-B AVG

UCC28070

www.ti.com

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

TYPICAL CHARACTERISTICS (continued)

Figure 16. Figure 17.

16 1

O O

CRMS

I V

h p

æ ö

= -

ç ÷

ç ÷ ç ÷

è ø è ø

è ø

16 1

O O

CRMS

I V

h p

æ ö

= -

ç ÷

ç ÷ ç ÷

è ø è ø

è ø

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

www.ti.com

APPLICATION INFORMATION

THEORY OF OPERATION

Interleaving

One of the main benefits from the 180°interleaving of phases is significant reductions in the high-frequency

ripple components of both the input current and the current into the output capacitor of the PFC pre-regulator.

Compared to that of a single-phase PFC stage of equal power, the reduced ripple on the input current eases the

burden of filtering conducted-EMI noise and helps reduce the EMI filter and CIN sizes. Additionally, reduced

high-frequency ripple current into the PFC output capacitor, COUT, helps to reduce its size and cost. Furthermore,

with reduced ripple and average current in each phase, the boost inductor size can be smaller than in a

single-phase design [1].

Ripple current reduction due to interleaving is often referred to as “ripple cancellation”, but strictly speaking, the

peak-to-peak ripple is completely cancelled only at 50% duty-cycle in a 2-phase system. At duty-cycles other

than 50%, ripple reduction occurs in the form of partial cancellation due to the superposition of the individual

phase currents. Nevertheless, compared to the ripple currents of an equivalent single-phase PFC pre-regulator,

those of a 2-phase interleaved design are extraordinarily smaller [1]. Independent of ripple cancellation, the

frequency of the interleaved ripple, at both the input and output, is 2 x fPWM.

On the input, 180°interleaving reduces the peak-to-peak ripple amplitude to 1/2 or less of the ripple amplitude of

the equivalent single-phase current.

On the output, 180°interleaving reduces the rms value of the PFC-generated ripple current in the output

capacitor by a factor of slightly more than √2, for PWM duty-cycles >50%.

This can be seen in the following derivations, adapting the method by Erickson [2].

In a single-phase PFC pre-regulator, the total rms capacitor current contributed by the PFC stage at all

duty-cycles can be shown to be approximated by:

(1)

In a dual-phase interleaved PFC pre-regulator, the total rms capacitor current contributed by the PFC stage for D

>50% can be shown to be approximated by:

(2)

In these equations, IO= average PFC output load current, VO= average PFC output voltage, VM= peak of the

input ac-line voltage, and η= efficiency of the PFC stage at these conditions. It can be seen that the quantity

under the radical for iCrms2φis slightly smaller than 1/2 of that under the radical for iCrms1φ. The rms currents

shown contain both the low-frequency and the high-frequency components of the PFC output current.

Interleaving reduces the high-frequency component, but not the low-frequency component.

( ) ( )

7500

PWM

R k f kHz

W =

( )

2 1

DMX RT MAX

R R D= ´ ´ -

( ) ( )

937 5

RDM

R k f kHz

W =

( ) 66 7 RDM

CDR

R ( k )

C pF . f ( kHz )

æ ö

= ´ç ÷

è ø

UCC28070

www.ti.com

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Programming the PWM Frequency and Maximum Duty-Cycle Clamp

The PWM frequency and maximum duty-cycle clamps for both GDx outputs of the UCC28070 are set through

the selection of the resistors connected to the RT and DMAX pins, respectively. The selection of the RT resistor

(RRT) directly sets the PWM frequency (fPWM).

(3)

Once RRT has been determined, the DMAX resistor (RDMX) may be derived.

(4)

where DMAX is the desired maximum PWM duty-cycle.

Frequency Dithering (Magnitude and Rate)

Frequency dithering refers to modulating the switching frequency to achieve a reduction in conducted-EMI noise

beyond the capability of the line filter alone. The UCC28070 implements a triangular modulation method which

results in equal time spent at every point along the switching frequency range. This total range from minimum to

maximum frequency is defined as the dither magnitude, and is centered around the nominal switching frequency

fPWM set with RRT. For example, a dither magnitude of 20 kHz on a nominal fPWM of 100 kHz results in a

frequency range of 100 kHz ±10 kHz. Furthermore, the programmed duty-cycle clamp set by RDMX remains

constant at the programmed value across the entire range of the frequency dithering.

The rate at which fPWM traverses from one extreme to the other and back again is defined as the dither rate. For

example, a dither rate of 1 kHz would linearly modulate the nominal frequency from 110 kHz to 90 kHz to 110

kHz once every millisecond. A good initial design target for dither magnitude is ±10% of fPWM. Most boost

components can tolerate such a spread in fPWM. The designer can then iterate around there to find the best

compromise between EMI reduction, component tolerances, and loop stability.

The desired dither magnitude is set by a resistor from the RDM pin to GND, of value calculated by the following

equation:

(5)

Once the value of RRDM is determined, the desired dither rate may be set by a capacitor from the CDR pin to

GND, of value calculated by the following equation:

(6)

Frequency dithering may be fully disabled by forcing the CDR pin >5 V or by connecting it to VREF (6 V) and

connecting the RDM pin directly to GND. (If populated, the relatively high impedance of the RDM resistor may

allow system switching noise to couple in and interfere with the controller timing functions if not bypassed with a

low impedance path when dithering is disabled.)

If an external frequency source is used to synchronize fPWM and frequency dithering is desired, the external

frequency source must provide the dither magnitude and rate functions as the internal dither circuitry is disabled

to prevent undesired performance during synchronization. (See following section for more details.)

SYNC

PWM

( ) ( )

15000 1 1W = ´

SYNC

R k .

f kHz

= ´

SYNC SYNC SYNC

D t f

( ) ( )

15000 2 1

æ ö

W = ´ ´ - -

ç ÷

è ø

DMX MAX SYNC

SYNC

R k D D

f ( kHz )

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

www.ti.com

External Clock Synchronization

The UCC28070 has also been designed to be easily synchronized to almost any external frequency source. By

disabling frequency dithering (pulling CDR >5 V), the UCC28070’s SYNC circuitry is enabled permitting the

internal oscillator to be synchronized with pulses presented on the RDM pin. In order to ensure a precise 180

degree phase shift is maintained between the GDA and GDB outputs, the frequency (fSYNC) of the pulses

presented at the RDM pin needs to be at twice the desired fPWM. For example, if a 100-kHz switching frequency

is desired, the fSYNC should be 200 kHz.

(7)

In order to ensure the internal oscillator does not interfere with the SYNC function, RRT should be sized to set the

internal oscillator frequency at least 10% below the fSYNC.

(8)

It must be noted that the PWM modulator gain will be reduced by a factor equivalent to the scaled RRT due to a

direct correlation between the PWM ramp current and RRT. Adjustments to the current loop gains should be

made accordingly.

It must also be noted that the maximum duty cycle clamp programmability is affected during external

synchronization. The internal timing circuitry responsible for setting the maximum duty cycle is initiated on the

falling edge of the synchronization pulse. Therefore, the selection of RDMX becomes dependent on the

synchronization pulse width (tSYNC).

For use in RDMX equation immediately below. (9)

(10)

Consequently to minimize the impact of the tSYNC it is clearly advantageous to utilize the smallest synchronization

pulse width feasible.

NOTE

When external synchronization is used, a propagation delay of approximately 50 ns to 100

ns exists between internal timing circuits and the SYNC signal’s falling edge, which may

result in reduced off-time at the highest of switching frequencies. Therefore, RDMX should

be adjusted downward slightly by (TSYNC-0.1 μs)/TSYNC to compensate. At lower SYNC

frequencies, this delay becomes an insignificant fraction of the PWM period, and can be

neglected.

( )

A B

kR R

UCC28070

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Multi-phase Operation

External synchronization also facilitates using more than 2 phases for interleaving. Multiple UCC28070s can

easily be paralleled to add an even number of additional phases for higher-power applications. With appropriate

phase-shifting of the synchronization signals, even more input and output ripple current cancellation can be

obtained. (An odd number of phases can be accommodated if desired, but the ripple cancellation would not be

optimal.) For 4-, 6-, or any 2 x n-phases (where n = the number of UCC28070 controllers), each controller should

receive a SYNC signal which is 360/n degrees out of phase with each other. For a 4-phase application

interleaving with two controllers, SYNC1 should be 180°out of phase with SYNC2 for optimal ripple cancellation.

Similarly for a 6-phase system, SYNC1, SYNC2, and SYNC3 should be 120°out of phase with each other for

optimal ripple cancellation.

In a multi-phase interleaved system, each current loop is independent and treated separately, however there is

only one common voltage loop. To maintain a single control loop, all VSENSE, VINAC, SS, IMO and VAO

signals are paralleled, respectively between the n controllers. Where current-source outputs are combined (SS,

IMO, VAO), the calculated load impedances must be adjusted by 1/n to maintain the same performance as with

a single controller.

Figure 18 illustrates the paralleling of two controllers for a 4-phase 90°-interleaved PFC system.

VSENSE and VINAC Resistor Configuration

The primary purpose of the VSENSE input is to provide the voltage feedback from the output to the voltage

control loop. Thus, a traditional resistor-divider network needs to be sized and connected between the output

capacitor and the VSENSE pin to set the desired output voltage based on the 3-V regulation voltage on

VSENSE.

A unique aspect of the UCC28070 is the need to place the same resistor-divider network on the VIN side of the

inductor to the VINAC pin. This provides the scaled input voltage monitoring needed for the linear multiplier and

current synthesizer circuitry. It is not required that the actual resistance of the VINAC network be identical to the

VSENSE network, but it is necessary that the attenuation (kR) of the two divider networks be equivalent for

proper PFC operation.

(11)

In noisy environments, it may be beneficial for small filter capacitors to be applied to the VSENSE and VINAC

inputs to avoid the destabilizing effects of excessive noise on these inputs. If applied, the RC time-constant

should not exceed 100μs on the VSENSE input to avoid significant delay in the output transient response. The

RC time-constant should also not exceed 100 μs on the VINAC input to avoid degrading of the wave-shape

zero-crossings. Usually, a time constant of 3/fPWM is adequate to filter out typical noise on VSENSE and VINAC.

Some design and test iteration may be required to find the optimal amount of filtering required in a particular

application.

VSENSE and VINAC Open Circuit Protection

Both the VSENSE and VINAC pins have been designed with an internal 250-nA current sink to ensure that in the

event of an open circuit at either pin, the voltage is not left undefined, and the UCC28070 remains in a “safe”

operating mode.

+–

VIN

12V to 21V

COUT

CSS

RRT2

RDMX2

CREF

RS3

RS4

CAOA

CAOBPKLMT

GND

VAO

VINAC

VSENSE

CSA

CSB

CDR

GDB

GDA

IMO VCC

RSYNTH

VREF

DMAX

RDM

RIMO

To CSA2

To CSB2

12V to 21V

CP C

CZC

RZC RZC

RRT1

RDMX1

RPK1

RPK2

RS1

RS2

CAOA

CAOBPKLMT

GND

VAO

VINAC

VSENSE

CSA

CSB

CDR

GDB

GDA

IMO VCC

RSYNTH

VREF

DMAX

RDM

To CSB1

To CSA1

VREF1

From Ixfrms

RSYN1

VREF1

Vin

From Ixfrms VREF2

Synchronized

Clocks

w/ 180 o

Phase Shift

CSB1

CSA1

CSB2

CSA2

RSYN2

RZC RZC

CZC CZC

CZC

CPC

CREF

CZV

RZV

CPV

VOUT

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

www.ti.com

Figure 18. Simplified Four-Phase Application Diagram Using Two UCC28070

Waveform at

CSx input

Synthesized

down-slope

Current Synthesizer

output to CA

( ) ( )

( )

10 CT B R

SYN

N L H k

R k R

m´ ´ ´

W = W

UCC28070

www.ti.com

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

Current Synthesizer

One of the most prominent innovations in the UCC28070 design is the current synthesizer circuitry that

synchronously monitors the instantaneous inductor current through a combination of on-time sampling and

off-time down-slope emulation.

During the on-time of the GDA and GDB outputs, the inductor current is recorded at the CSA and CSB pins

respectively via the current transformer network in each output phase. Meanwhile, the continuous monitoring of

the input and output voltage via the VINAC and VSENSE pins permits the UCC28070 to internally recreate the

inductor current’s down-slope during each output’s respective off-time. Through the selection of the RSYNTH

resistor (RSYN), based on the equation below, the internal response may be adjusted to accommodate the wide

range of inductances expected across the wide array of applications.

During inrush surge events at power-up and ac drop-out recovery, VSENSE <VINAC, so the synthesized down

slope becomes zero. In this case, the synthesized inductor current will remain above the IMO reference and the

current loop drives the duty cycle to zero. This avoids excessive stress on the MOSFETS during the surge event.

Once VINAC falls below VSENSE the duty cycle increases until steady-state operation resumes.

Figure 19. Inductor Current’s Down Slope

(12)

Variables

•LB= Nominal Zero-Bias Boost Inductance (μH),

•RS= Sense Resistor (Ω),

•NCT = Current-sense Transformer turns ratio,

•kR= RB/(RA+RB) = the resistor-divider attenuation at the VSENSE and VINAC pins.

CSx

+IPEAKx

PKLMT

Current

Synthesizer

Externally Programmable Peak

Current Limit level (PKLMT)

3V Average Current-sense

Signal Range, plus Ripple

To Current

Amplifier

To Gate-Drive

Shut-down

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

www.ti.com

Programmable Peak Current Limit

The UCC28070 has been designed with a programmable cycle-by-cycle peak current limit dedicated to disabling

either GDA or GDB output whenever the corresponding current-sense input (CSA or CSB respectively) rises

above the voltage established on the PKLMT pin. Once an output has been disabled via the detection of peak

current limit, the output remains disabled until the next clock cycle initiates a new PWM period. The programming

range of the PKLMT voltage extends to upwards of 4 V to permit the full utilization of the 3-V average current

sense signal range, however it should be noted that the linearity of the current amplifiers begin to compress

above 3.6 V.

A resistor-divider network from VREF to GND can easily program the peak current limit voltage on PKLMT,

provided the total current out of VREF is less than 2 mA to avoid drooping of the 6-V VREF voltage. A load of

less than 0.5 mA is suggested, but if the resistance on PKLMT is very high, a small filter capacitor on PKLMT is

recommended to avoid operational problems in high-noise environments.

Figure 20. Externally Programmable Peak Current Limit

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

Linear Multiplier

The multiplier of the UCC28070 generates a reference current which represents the desired wave shape and

proportional amplitude of the ac input current. This current is converted to a reference voltage signal by the RIMO

resistor, which is scaled in value to match the voltage of the current-sense signals. The instantaneous multiplier

current is dependent upon the rectified, scaled input voltage VVINAC and the voltage-error amplifier output VVAO.

The VVINAC signal conveys three pieces of information to the multiplier:

1. The overall wave-shape of the input voltage (typically sinusoidal),

2. The instantaneous input voltage magnitude at any point in the line cycle,

3. The rms level of the input voltage.

The VVAO signal represents the total output power of the PFC pre-regulator.

A major innovation in the UCC28070 multiplier architecture is the internal quantized VRMS feed-forward (QVFF)

circuitry, which eliminates the requirement for external filtering of the VINAC signal and the subsequent slow

response to transient line variations. A unique circuit algorithm detects the transition of the peak of VVINAC

through seven thresholds and generates an equivalent VFF level centered within the eight QVFF ranges. The

boundaries of the ranges expand with increasing VIN to maintain an approximately equal-percentage delta

between levels. These eight QVFF levels are spaced to accommodate the full “universal”line range of 85 V-265

VRMS.

A great benefit of the QVFF architecture is that the fixed kVFF factors eliminate any contribution to distortion of the

multiplier output, unlike an externally-filtered VINAC signal which unavoidably contains 2nd-harmonic distortion

components. Furthermore, the QVFF algorithm allows for rapid response to both increasing and decreasing

changes in input rms voltage so that disturbances transmitted to the PFC output are minimized. 5% hysteresis in

the level thresholds help avoid “chattering”between QVFF levels for VVINAC voltage peaks near a particular

threshold or containing mild ringing or distortion. The QVFF architecture requires that the input voltage be largely

sinusoidal, and relies on detecting zero-crossings to adjust QVFF downward on decreasing input voltage.

Zero-crossings are defined as VVINAC falling below 0.7 V for at least 50 μs typically.

Table 1 reflects the relationship between the various VINAC peak voltages and the corresponding kVFF terms for

the multiplier equation.

Table 1. VINAC Peak Voltages

LEVEL VVINAC PEAK VOLTAGE kVFF (V2) VIN PEAK VOLTAGE (1)

8 2.60 V ≤VVINAC(pk) 3.857 >345 V

7 2.25 V ≤VVINAC(pk) <2.60 V 2.922 300 V to 345 V

6 1.95 V ≤VVINAC(pk) <2.25 V 2.199 260 V to 300 V

5 1.65 V ≤VVINAC(pk) <1.95 V 1.604 220 V to 260 V

4 1.40 V ≤VVINAC(pk) <1.65 V 1.156 187 V to 220 V

3 1.20 V ≤VVINAC(pk) <1.40 V 0.839 160 V to 187 V

2 1.00 V ≤VVINAC(pk) <1.20 V 0.600 133 V to 160 V

1 VVINAC(pk) ≤1.00 V 0.398 <133 V

(1) The VIN peak voltage boundary values listed above are calculated based on a 400-V PFC output voltage and the use of a matched

resistor-divider network (kR= 3 V/400 V = 0.0075) on VINAC and VSENSE (as required for current synthesis). When VOUT is designed

to be higher or lower than 400 V, kR= 3 V/VOUT, and the VIN peak voltage boundary values for each QVFF level adjust to VVINAC(pk)/kR.

( ) ( )

17 1

VINAC VAO

IMO

VFF

A V V

m ´ ´ -

1 10 OUT (max)

IN (max)

. P

1 414

IN (max)

IN ( rms ) IN ( pk ) IN ( rms )

RMS

I , I . I

= = ´Thus and

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

www.ti.com

The multiplier output current IIMO for any line and load condition can thus be determined by the equation

(13)

Because the kVFF value represents the scaled VRMS 2at the center of a level, VVAO will adjust slightly upwards or

downwards when VINACpk is either lower or higher than the center of the QVFF voltage range to compensate for

the difference. This is automatically accomplished by the voltage loop control when VIN varies, both within a level

and after a transition between levels.

The output of the voltage-error amplifier VAO is clamped at 5.0 V, which represents the maximum PFC output

power. This value is used to calculate the maximum reference current at the IMO pin, and sets a limit for the

maximum input power allowed (and, as a consequence, limits maximum output power).

Unlike a continuous VFF situation, where maximum input power is a fixed power at any VRMS input, the discrete

QVFF levels permit a variation in maximum input power within limited boundaries as the input VRMS varies within

each level.

The lowest maximum power limit occurs at the VINAC voltage of 0.76 V, while the highest maximum power limit

occurs at the increasing threshold from level-1 to level-2. This pattern repeats at every level transition threshold,

keeping in mind that decreasing thresholds are 95% of the increasing threshold values. Below VINAC = 0.76 V,

PIN is always less than PIN(max), falling linearly to zero with decreasing input voltage.

For example, to design for the lowest maximum power allowable, determine the maximum steady-state (average)

output power required of the PFC pre-regulator and add some additional percentage to account for line drop-out

recovery power (to recharge COUT while full load power is drawn) such as 10% or 20% of POUT(max). Then apply

the expected efficiency factor to find the lowest maximum input power allowable:

(14)

At the PIN(max) design threshold, VVINAC = 0.76 V, hence QVFF = 0.398 and input VAC = 73 VRMS (accounting for

2-V bridge-rectifier drop) for a nominal 400-V output system.

(15)

( )( )

0 76 5 1

17 130

0 398

IMO(max)

. V V V

I A A

m m

= ´ =

IMO IMO(max) IN ( pk )

R I I N

æ ö

´ = ´ ´

ç ÷

è ø

( )

2IN ( pk ) S

IMO

CT IMO(max)

I R

RN I

æ ö

æ ö´ ´

ç ÷

è ø

=´

( )( )

1 2

1 0 5 1

17 171

0 398

IMO( L L )

. V V V

I A A

m m

= ´ =

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

This IIN(pk) value represents the combined average current through the boost inductors at the peak of the line

voltage. Each inductor current is detected and scaled by a current-sense transformer (CT). Assuming equal

currents through each interleaved phase, the signal voltage at each current sense input pin (CSA and CSB) is

developed across a sense resistor selected to generate ~3 V based on (1/2) x IIN(pk) x RS/NCT, where RSis the

current sense resistor and NCT is the CT turns-ratio.

IIMO is then calculated at that same lowest maximum-power point, as

(16)

RIMO is selected such that:

(17)

Therefore:

(18)

At the increasing side of the level-1 to level-2 threshold, it should be noted that the IMO current would allow

higher input currents at low-line:

(19)

However, this current may easily be limited by the programmable peak current limiting (PKLMT) feature of the

UCC28070 if required by the power stage design.

The same procedure can be used to find the lowest and highest input power limits at each of the QVFF level

transition thresholds. At higher line voltages, where the average current with inductor ripple is traditionally below

the PKLMT threshold, the full variation of maximum input power will be seen, but the input currents will inherently

be below the maximum acceptable current levels of the power stage.

The performance of the multiplier in the UCC28070 has been significantly enhanced when compared to previous

generation PFC controllers, with high linearity and accuracy over most of the input ranges. The accuracy is at its

worst as VVAO approaches 1 V because the error of the (VVAO-1) subtraction increases and begins to distort the

IMO reference current to a greater degree.

UCC28070

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Enhanced Transient Response (VA Slew-Rate Correction)

Due to the low voltage loop bandwidth required to maintain proper PFC and ignore the slight 120-Hz ripple on

the output, the response of ordinary controllers to input voltage and load transients will also be slow. However,

the QVFF function effectively handles the line transient response with the exception of any minor adjustments

needed within a QVFF level. Load transients on the other hand can only be handled by the voltage loop, therefore,

the UCC28070 has been designed to improve its transient response by pulling up on the output of the voltage

amplifier (VAO) with an additional 100 μA of current in the event the VSENSE voltage drops below 93% of

regulation (2.79 V). During a soft-start cycle, when VSENSE is ramping up from the 0.75-V PFC Enable

threshold, the 100-μA correction current source is disabled to ensure the gradual and controlled ramping of

output voltage and current during a soft start.

Voltage Biasing (VCC and VREF)

The UCC28070 operates within a VCC bias supply range of 10 V to 21 V. An Under-Voltage Lock-Out (UVLO)

threshold prevents the PFC from activating until VCC >10.2 V, and 1 V of hysteresis assures reliable start-up

from a possibly low-compliance bias source. An internal 25-V zener-like clamp on VCC is intended only to protect

the device from brief energy-limited surges from the bias supply, and should NOT be used as a regulator with a

current-limited source.

At minimum, a 0.1-μF ceramic bypass capacitor must be applied from VCC to GND close to the device pins to

provide local filtering of the bias supply. Larger values may be required depending on ICC peak current

magnitudes and durations to minimize ripple voltage on VCC.

In order to provide a smooth transition out of UVLO and to make the 6-V voltage reference available as early as

possible, the VREF output is enabled when VCC exceeds 8 V typically.

The VREF circuitry is designed to provide the biasing of all internal control circuits and for limited use externally.

At minimum, a 22-nF ceramic bypass capacitor must be applied from VREF to GND close to the device pins to

ensure stability of the circuit. External load current on VREF should be limited to less than 2 mA, or degraded

regulation may result.

PFC Enable and Disable

The UCC28070 contains two independent circuits dedicated to disabling the GDx outputs based on the biasing

conditions of the VSENSE or SS pins. The first circuit which monitors the VVSENSE, is the traditional PFC Enable

that holds off soft-start and the overall PFC function until the output has pre-charged to ~25%. Prior to VVSENSE

reaching 0.75 V, almost all of the internal circuitry is disabled. Once VVSENSE reaches 0.75 V and VAO <0.75 V,

the oscillator, multiplier, and current synthesizer are enabled and the SS circuitry begins to ramp up the voltage

on the SS pin. The second circuit provides an external interface to emulate an internal fault condition to disable

the GDx output without fully disabling the voltage loop and multiplier. By externally pulling the SS pin below 0.6

V, the GDx outputs are immediately disabled and held low. Assuming no other fault conditions are present,

normal PWM operation resumes when the external SS pull-down is released. It must be noted that the external

pull-down needs to be sized large enough to override the internal 1.5-mA adaptive SS pull-up once the SS

voltage falls below the disable threshold. It is recommended that a MOSFET with less than 100-ΩRDS(on)

resistance be used to ensure the SS pin is held adequately below the disable threshold.

2 25

SS SS

. V

t C Am

æ ö

= ´ç ÷

è ø

VSENSE

SS SS

V V

t C Am

æ ö

= ´ç ÷

è ø

VSS

VVSENSE

VSS if no adaptive current

Time (s)

(V)

PFC externally

disabled due to

AC-line drop-out AC-Line recovers

and SS pin released

Reduced delay to regulation

UCC28070

www.ti.com

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

Adaptive Soft Start

In order to maintain a controlled power up, the UCC28070 has been designed with an adaptive soft-start function

that overrides the internal reference voltage with a controlled voltage ramp during power up. On initial power up,

once VVSENSE exceeds the 0.75-V enable threshold (VEN), the internal pull down on the SS pin is released, and

the 1.5-mA adaptive soft-start current source is activated. This 1.5-mA pull-up almost immediately pulls the SS

pin to 0.75 V (VVSENSE) to bypass the initial 25% of dead time during a traditional 0 V to Vregulation SS ramp.

Once the SS pin has reached the voltage on VSENSE, the 10-μA soft-start current (ISS) takes over. Thus,

through the selection of the soft-start capacitor (CSS), the effective soft-start time (tSS) may be easily programmed

based on the equation below.

(20)

Often, a system restart is desired following a brief shut-down. In such a case, VSENSE may still have substantial

voltage if VOUT has not fully discharged or if high line has peak charged COUT. To eliminate the delay caused by

charging CSS from 0 V up to the pre-charged VVSENSE with only the 10-μA current source and minimize any

further output voltage sag, the adaptive soft start uses a 1.5-mA current source to rapidly charge CSS to VVSENSE,

after which time the 10-μA source controls the VSS accent to the desired soft-start ramp rate. In such a case, tSS

is estimated as follows:

(21)

where VVSENSE0 is the voltage at VSENSE at the moment a soft start or restart is initiated.

NOTE

For soft start to be effective and avoid overshoot on VOUT, the SS ramp must be slower

than the voltage-loop control response. Choose CSS ≥CVZ to ensure this.

Figure 21. Soft-Start Ramp Rate

UCC28070

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PFC Start-Up Hold Off

An additional feature designed into the UCC28070 is the “Start-Up Hold Off”logic that prevents the device from

initiating a soft-start cycle until the VAO is below the zero-power threshold (0.75 V). This feature ensures that the

SS cycle will initiate from zero-power and zero duty-cycle while preventing the potential for any significant inrush

currents due to stored charge in the VAO compensation network.

Output Over-Voltage Protection (OVP)

Because of the high voltage output and a limited design margin on the output capacitor, output over-voltage

protection is essential for PFC circuits. The UCC28070 implements OVP through the continuous monitoring of

the VSENSE voltage. In the event VVSENSE rises above 106% of regulation (3.18 V), the GDx outputs are

immediately disabled to prevent the output voltage from reaching excessive levels. Meanwhile the CAOx outputs

are pulled low in order to ensure a controlled recovery starting from 0% duty-cycle after an OVP fault is released.

Once the VVSENSE voltage has dropped below 3.08 V, the PWM operation resumes normal operation.

Zero-Power Detection

In order to prevent undesired performance under no-load and near no-load conditions, the UCC28070

zero-power detection comparator is designed to disable both GDA and GDB output in the event the VAO voltage

falls below 0.75 V. The 150 mV of hysteresis ensures that the output remains disabled until the VAO has nearly

risen back into the linear range of the multiplier (VAO ≥0.9 V).

Thermal Shutdown

In order to protect the power supplies from silicon failures at excessive temperatures, the UCC28070 has an

internal temperature-sensing comparator that shuts down nearly all of the internal circuitry, and disables the GDA

and GDB outputs, if the die temperature rises above 160°C. Once the die temperature falls below 140°C, the

device brings the outputs up through a typical soft start.

Current

Synthesizer

CAOx

CSx

IMO

CPC

CZC

RZC

gmc = 100µS

CAx

RS CT

CA RMP SYNC B

Vout

v N

v V k s L

=D ´ ´ ´

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

Current Loop Compensation

The UCC28070 incorporates two identical and independent transconductance-type current-error amplifiers (one

for each phase) with which to control the shaping of the PFC input current waveform. The current-error amplifier

(CA) forms the heart of the embedded current control loop of the boost PFC pre-regulator, and is compensated

for loop stability using familiar principles [4, 5]. The output of the CA for phase-A is CAOA, and that for phase-B

is CAOB. Since the design considerations are the same for both, they are collectively referred to as CAOx,

where the "x" may be "A" or "B".

In a boost PFC pre-regulator, the current control loop comprises the boost power plant stage, the current sensing

circuitry, the wave-shape reference, the PWM stage, and the CA with compensation components. The CA

compares the average boost inductor current sensed with the wave-shape reference from the multiplier stage

and generates an output current proportional to the difference.

This CA output current flows through the impedance of the compensation network generating an output voltage,

VCAO, which is then compared with a periodic voltage ramp to generate the PWM signal necessary to achieve

PFC.

Figure 22. Current Error Amplifier With Type II Compensation

For frequencies above boost LC resonance and below fPWM, the small-signal model of the boost stage, which

includes current sensing, can be simplified to:

(22)

where LB= mid-value boost inductance, RS= CT sense resistor, NCT = CT turns ratio, VOUT = average output

voltage, ∆VRMP = 4Vpk-pk amplitude of the PWM voltage ramp, kSYNC = ramp reduction factor (if PWM frequency is

synchronized to an external oscillator; kSYNC = 1 otherwise), s = Laplace complex variable

An RZCCZC network is introduced on CAOx to obtain high gain for the low-frequency content of the inductor

current signal, but reduced flat gain above the zero frequency out to fPWM to attenuate the high-frequency

switching ripple content of the signal (thus averaging it).

RMP SYNC

LB CT

V k

g Rzc R

D ´

10 100

LB S

V N

Rzc S I Rm

£´ ´ D ´

CXO mc

RMP SYNC B

Vout N

f g Rzc

V k Lp

= ´

D ´ ´ ´

2PWM

Cpc f Rzcp

=´ ´

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

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The switching ripple voltage should be attenuated to less than 1/10 of the ΔVRMP amplitude so as to be

considered “negligible”ripple.

Thus, CAOx gain at fPWM is:

(23)

where ∆ILB is the maximum peak-to-peak ripple current in the boost inductor, and gmc is the transconductance of

the CA, 100 μS.

(24)

The current-loop cross-over frequency is then found by equating the open loop gain to 1 and solving for fCXO:

(25)

CCZ is then determined by setting fZC = fCXO = 1/(2πxRZCxCZC) and solving for CZC. At fZC = fCXO, a phase margin

of 45°is obtained at fCXO. Greater phase margin may be had by placing fZC <fCXO.

An additional high-frequency pole is generally added at fPWM to further attenuate ripple and noise at fPWM and

higher. This is done by adding a small-value capacitor, Cpc, across the RzcCzc network.

(26)

The procedure above is valid for fixed-value inductors.

NOTE

If a “swinging-choke”boost inductor (inductance decreases with increasing current) is

used, fCXO varies with inductance, so CZC should be determined at maximum inductance.

+VAO

VSENSE

CPV

CZV

RZV

gmv = 70µS

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

Voltage Loop Compensation

The outer voltage control loop of the dual-phase PFC controller functions the same as with a single-phase

controller, and compensation techniques for loop stability are standard [4]. The bandwidth of the voltage-loop

must be considerably lower than the twice-line ripple frequency (f2LF) on the output capacitor, to avoid

distortion-causing correction to the output voltage. The output of the voltage-error amplifier (VA) is an input to the

multiplier, to adjust the input current amplitude relative to the required output power. Variations on VAO within the

bandwidth of the current loops will influence the wave-shape of the input current. Since the low-frequency ripple

on COUT is a function of input power only, its peak-to-peak amplitude is the same at high-line as at low-line. Any

response of the voltage-loop to this ripple will have a greater distorting effect on high-line current than on low-line

current. Therefore, the allowable percentage of 3rd-harmonic distortion on the input current contributed by VAO

should be determined using high-line conditions.

Because the voltage-error amplifier (VA) is a transconductance type of amplifier, the impedance on its input has

no bearing on the amplifier gain, which is determined solely by the product of its transconductance (gmv) with its

output impedance (ZOV). Thus the VSENSE input divider-network values are determined separately, based on

criteria discussed in the VINAC section. Its output is the VAO pin.

Figure 23. Voltage Error Amplifier With Type II Compensation

The twice-line ripple voltage component of VSENSE must be sufficiently attenuated and phase-shifted at VAO to

achieve the desired level of 3rd-harmonic distortion of the input current wave-shape [4]. For every 1% of

3rd-harmonic input distortion allowable, the small-signal gain GVEA = VVAOpk / vSENSEpk = gmvxZOV at the twice-line

frequency should allow no more than 2% ripple over the full VAO voltage range. In the UCC28070, VVAO can

range from 1 V at zero load power to ~4.2 V(see note below) at full load power for a ΔVVAO = 3.2 V, so 2% of 3.2

V is 64-mV peak ripple.

NOTE

Although the maximum VAO voltage is clamped at 5 V, at full load VVAO may vary around

an approximate center point of 4.2 V to compensate for the effects of the quantized

feed-forward voltage in the multiplier stage (see Multiplier Section for details). Therefore,

4.2 V is the proper voltage to use to represent maximum output power when performing

voltage-loop gain calculations.

avg Cout avg

avg avg LF

Pin X Pin

vVout Vout f Coutp

= = ´ ´ ´

3 2

64 2

rd avg LF

OV ( f )

mv R avg

k mV Vout f Cout

Zg k Pin

p´ ´ ´ ´

=´ ´

3 2

64 2

mv R avg

rd avg LF

g k Pin

Cpv k mV Vout ( f ) Coutp

´ ´

=´ ´ ´ ´

( ) 1

avg Cout

VXO BST VEA R mv Cpv R

VAO avg

Pin X

Tv( f ) G G k g X k

V Vout

æ ö

= ´ ´ = ´ ´ ´ =

ç ÷

D ´

è ø

( )

mv R avg

VXO

VAO avg

g k Pin

V Vout Cpv Coutp

´ ´

=D ´ ´ ´ ´

2VXO

Rzv f Cpvp

=´

10 10

2VXO

Czv Cpv

f Rzvp

= » ´

UCC28070

SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

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The output capacitor maximum low-frequency zero-to-peak ripple voltage is closely approximated by:

(27)

where PIN(avg) is the total maximum input power of the interleaved-PFC pre-regulator, VOUT(avg) is the average

output voltage and COUT is the output capacitance.

VSENSEpk = vopkxkR, where kRis the gain of the resistor-divider network on VSENSE.

Thus, for k3rd% of allowable 3rd-harmonic distortion on the input current attributable to the VAO ripple,

(28)

This impedance on VAO is set by a capacitor (Cpv), where CPV = 1/( 2πf2LFxZOV(f2LF)) therefore,

(29)

The voltage-loop unity-gain cross-over frequency (fVXO) may now be solved by setting the open-loop gain equal

to 1:

(30)

so, (31)

The “zero-resistor”(RZV) from the zero-placement network of the compensation may now be calculated. Together

with CPV, RZV sets a pole right at fVXO to obtain 45°phase margin at the cross-over.

Thus, (32)

Finally, a zero is placed at or below fVXO/6 with capacitor CZV to provide high gain at dc but with a breakpoint far

enough below fVXO so as not to significantly reduce the phase margin. Choosing fVXO/10 allows one to

approximate the parallel combination value of CZV and CPV as CZV, and solve for CZV simply as:

(33)

By using a spreadsheet or math program, CZV, RZV, and CPV may be manipulated to observe their effects on fVXO

and phase margin and %-contribution to 3rd-harmonic distortion (see note below). Also, phase margin may be

checked as PIN(avg) level and system parameter tolerances vary.

NOTE

The percent of 3rd-harmonic distortion calculated in this section represents the

contribution from the f2LF voltage ripple on COUT only. Other sources of distortion, such as

the current-sense transformer, the current synthesizer stage, even distorted VIN, etc., can

contribute additional 3rd and higher harmonic distortion.

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

Advanced Design Techniques

Current Loop Feedback Configuration

(Sizing of the Current Transformer Turns Ratio and Sense Resistor (RS)

A current-sense transformer (CT) is typically used in high-power applications to sense inductor current while

avoiding significant losses in the sensing resistor. For average current-mode control, the entire inductor current

waveform is required; however low-frequency CTs are obviously impracticable. Normally, two high-frequency

CTs are used, one in the switching leg to obtain the up-slope current and one in the diode leg to obtain the

down-slope current. These two current signals are summed together to form the entire inductor current, but this

is not the case for the UCC28070.

A major advantage of the UCC28070 design is the current synthesis function, which internally recreates the

inductor current down-slope during the switching period off-time. This eliminates the need for the diode-leg CT in

each phase, significantly reducing space, cost and complexity. A single resistor programs the synthesizer down

slope, as previously discussed in the Current Synthesizer section.

A number of trade-offs must be made in the selection of the CT. Various internal and external factors influence

the size, cost, performance, and distortion contribution of the CT.

These factors include, but are not limited to:

•Turns-ratio (NCT)

•Magnetizing inductance (LM)

•Leakage inductance (LLK)

•Volt-microsecond product (Vμs)

•Distributed capacitance (Cd)

•Series resistance (RSER)

•External diode drop (VD)

•External current sense resistor (RS)

•External reset network

Traditionally, the turns-ratio and the current sense resistor are selected first. Some iterations may be needed to

refine the selection once the other considerations are included.

RSER

LLK

NCT

IDS iMRS

Reset

Network

CSx

UCC28070

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In general, 50 ≤NCT ≤200 is a reasonable range from which to choose. If NCT is too low, there may be high

power loss in RSand insufficient LM. If too high, there could be excessive LLK and Cd. (A one-turn primary

winding is assumed.)

Figure 24. Current Sense Transformer Equivalent Circuit

A major contributor to distortion of the input current is the effect of magnetizing current on the CT output signal

(iRS). A higher turns-ratio results in a higher LMfor a given core size. LMshould be high enough that the

magnetizing current (iM) generated is a very small percentage of the total transformed current. This is an

impossible criterion to maintain over the entire current range, because iMunavoidably becomes a larger fraction

of iRS as the input current decreases toward zero. The effect of iMis to “steal”some of the signal current away

from RS, reducing the CSx voltage and effectively understating the actual current being sensed. At low currents,

this understatement can be significant and CAOx increases the current-loop duty-cycle in an attempt to correct

the CSx input(s) to match the IMO reference voltage. This unwanted correction results in overstated current on

the input wave shape in the regions where the CT understatement is significant, such as near the ac line zero

crossings. It can affect the entire waveform to some degree under the high line, light-load conditions.

The sense resistor RSis chosen, in conjunction with NCT, to establish the sense voltage at CSx to be about 3 V

at the center of the reflected inductor ripple current under maximum load. The goal is to maximize the average

signal within the common-mode input range VCMCAO of the CAOx current-error amplifiers, while leaving room for

the peaks of the ripple current within VCMCAO. The design condition should be at the lowest maximum input power

limit as determined in the Multiplier Section. If the inductor ripple current is so high as to cause VCSx to exceed

VCMCAO, then RSor NCT or both must be adjusted to reduce peak VCSx, which could reduce the average sense

voltage center below 3 V. There is nothing wrong with this situation; but be aware that the signal is more

compressed between full- and no-load, with potentially more distortion at light loads.

The matter of volt-second balancing is important, especially with the widely varying duty-cycles in the PFC stage.

Ideally, the CT is reset once each switching period; that is, the off-time Vμs product equals the on-time Vμs

product. (Because a switching period is usually measured in microseconds, it is convenient to convert the

volt-second product to volt-microseconds to avoid sub-decimal numbers.) On-time Vμs is the time-integral of the

voltage across LMgenerated by the series elements RSER, LLK, D, and RS. Off-time Vμs is the time-integral of the

voltage across the reset network during the off-time. With passive reset, Vμs-off is unlikely to exceed Vμs-on.

Sustained unbalance in the on or off Vμs products will lead to core saturation and a total loss of the

current-sense signal. Loss of VCSx causes VCAOx to quickly rise to its maximum, programming a maximum

duty-cycle at any line condition. This, in turn causes the boost inductor current to increase without control, until

the system fuse or some component failure interrupts the input current.

( ) ( ) ( )

RS D RSER LK

on max ON max

V t V V V V

m= ´ + + +

ZRST

RRST

D D

RRST

CRST

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

It is vital that the CT has plenty of Vμs design-margin to accommodate various special situations where there to

be several consecutive maximum duty-cycle periods at maximum input current, such as during peak current

limiting.

Maximum Vμs(on) can be estimated by:

(34)

where all factors are maximized to account for worst-case transient conditions and tON(max) occurs during the

lowest dither frequency when frequency dithering is enabled. For design margin, a CT rating of ~5*Vμs(on)max

or higher is suggested. The contribution of VRS varies directly with the line current. However, VDmay have a

significant voltage even at near-zero current, so substantial Vμs(on) may accrue at the zero-crossings where the

duty-cycle is maximum. VRSER is the least contributor, and often can be neglected if RSER<<RS. VLK is developed

by the di/dt of the sensed current, and is not observable externally. However, its impact is considerable, given

the sub-microsecond rise-time of the current signal plus the slope of the inductor current. Fortunately, most of the

built-up Vμs across LMduring the on-time is removed during the fall-time at the end of the duty-cycle, leaving a

lower net Vμs(on) to be reset during the off-time. Nevertheless, the CT must, at the very minimum, be capable of

sustaining the full internal Vμs(on)max built up until the moment of turn-off within a switching period.

Vμs(off) may be generated with a resistor or zener diode, using the iMas bias current.

Figure 25. Possible Reset Networks

In order to accommodate various CT circuit designs and prevent the potentially destructive result due to CT

saturation, the UCC28070’s maximum duty-cycle needs to be programmed such that the resulting minimum

off-time accomplishes the required worst-case reset. (See the PWM Frequency and Duty-Cycle Clamp section of

the data sheet for more information on sizing RDMX) Be aware that excessive Cdin the CT can interfere with

effective resetting, because the maximum reset voltage is not reached until after 1/4-period of the CT

self-resonant frequency. A higher turns-ratio results in higher Cd[3], so a trade-off between NCT and DMAX must

be made.

The selected turns-ratio also affects LMand LLK, which vary proportionally to the square of the turns. Higher LMis

good, while higher LLK is not. If the voltage across LMduring the on-time is assumed to be constant (which it is

not, but close enough to simplify) then the magnetizing current is an increasing ramp.

This upward ramping current subtracts from iRS, which affects VCSx especially heavily at the zero-crossings and

light loads, as stated earlier. With a reduced peak at VCSx, the current synthesizer starts the down-slope at a

lower voltage, further reducing the average signal to CAOx and further increasing the distortion under these

conditions. If low input current distortion at very light loads is required, special mitigation methods may need to

be developed to accomplish that goal.

GDA

VCC

RTA

RSA

ROA

CSA

VCC

RTB

RSB

ROB

GDB

CTA

CTB

DPA1

DPA2

DPB1

DPB2

UCC28070

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Current Sense Offset and PWM Ramp for Improved Noise Immunity

To improve noise immunity at extremely light loads, a PWM ramp with a dc offset is recommended to be added

to the current sense signals. Electrical components RTA, RTB, ROA, ROB, CTA, CTB, DPA1, DPA2, DPB1, DPB1 CTA,

CTB form a PWM ramp that is activated and deactivated by the gate drive outputs of the UCC28070. Resistor

ROA and ROB add a dc offset to the CS resistors (RSA and RSB).

Figure 26. PWM Ramp and Offset Circuit

SA SB

R R=

( )

VCC OFF SA

OA OB

OFF

V V R

R R V

= =

( )

0 1

VCC S OFF DA SA

TA TB

S OFF

V (V . V ) V R

R R V . V

- ´ - +

= = ´ -

TA TB

TA S

C C R f

= = ´ ´

UCC28070

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SLUS794E –NOVEMBER 2007–REVISED APRIL 2011

When the inductor current becomes discontinuous the boost inductors ring with the parasitic capacitances in the

boost stages. This inductor current rings through the CTs causing a false current sense signal. Please refer to

the following graphical representation of what the current sense signal looks like when the inductor current goes

discontinuous.

NOTE

The inductor current and RS may vary from this graphical representation depending on

how much inductor ringing is in the design when the unit goes discontinuous.

Figure 27. False Current Sense Signal

To counter for the offset (VOFF) just requires adjusting resistors ROA and ROB to ensure that when the unit goes

discontinuous the current sense resistor is not seeing a positive current when it should be zero. Setting the offset

to 120 mV is a good starting point and may need to be adjusted based on individual design criteria.

(35)

(36)

A small PWM ramp that is equal to 10% of the maximum current sense signal (VS) less the offset can then be

added by properly selecting RTA, RTB, CTA and CTB.

(37)

(38)

UCC28070

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Recommended PCB Device Layout

Interleaved PFC techniques dramatically reduce input and output ripple current caused by the PFC boost

inductor, which allows the circuit to use smaller and less expensive filters. To maximize the benefits of

interleaving, the output filter capacitor should be located after the two phases allowing the current of each phase

to be combined together before entering the boost capacitor. Similar to other power management devices, when

laying out the PCB it is important to use star grounding techniques and to keep filter and high frequency bypass

capacitors as close to device pins and ground as possible. To minimize the possibility of interference caused by

magnetic coupling from the boost inductor, the device should be located at least 1 inch away from the boost

inductor. It is also recommended that the device not be placed underneath magnetic elements.

References

1. O’Loughlin, Michael, “An Interleaving PFC Pre-Regulator for High-Power Converters”, Texas Instruments,

Inc. 2006 Unitrode Power Supply Seminar, Topic 5

2. Erickson, Robert W., “Fundamentals of Power Electronics”, 1st ed., pp. 604-608 Norwell, MA: Kluwer

Academic Publishers, 1997

3. Creel, Kirby "Measuring Transformer Distributed Capacitance", White Paper, Datatronic Distribution, Inc.

website: http://www.datatronics.com/pdf/distributed_capacitance_paper.pdf

4. L. H. Dixon, "Optimizing the Design of a High Power Factor Switching Preregulator", Unitrode Power Supply

Design Seminar Manual SEM700, 1990. Texas Instruments Literature Number SLUP093

5. L. H. Dixon, "High Power Factor Preregulator for Off-Line Power Supplies", Unitrode Power Supply Design

Seminar Manual SEM600, 1988. Texas Instruments Literature Number SLUP087

REVISION HISTORY

Changes from Revision C (June 2009) to Revision D Page

•Changed 30 kHz to 300 kHz ................................................................................................................................................. 1

Changes from Revision D (June 2010) to Revision E Page

•Changed PWM switching frequency ..................................................................................................................................... 4

•Changed Figure 8 ............................................................................................................................................................... 12

PACKAGE OPTION ADDENDUM

www.ti.com 20-Apr-2011

Addendum-Page 1

PACKAGING INFORMATION

Orderable Device Status (1) Package Type Package

Drawing Pins Package Qty Eco Plan (2) Lead/

Ball Finish MSL Peak Temp (3) Samples

(Requires Login)

UCC28070DW ACTIVE SOIC DW 20 25 Green (RoHS

& no Sb/Br) CU NIPDAU Level-1-260C-UNLIM

UCC28070DWR ACTIVE SOIC DW 20 2000 Green (RoHS

& no Sb/Br) CU NIPDAU Level-1-260C-UNLIM

UCC28070PW ACTIVE TSSOP PW 20 70 Green (RoHS

& no Sb/Br) CU NIPDAU Level-1-260C-UNLIM

UCC28070PWG4 ACTIVE TSSOP PW 20 70 Green (RoHS

& no Sb/Br) CU NIPDAU Level-1-260C-UNLIM

UCC28070PWR ACTIVE TSSOP PW 20 2000 Green (RoHS

& no Sb/Br) CU NIPDAU Level-1-260C-UNLIM

UCC28070PWRG4 ACTIVE TSSOP PW 20 2000 Green (RoHS

& no Sb/Br) CU NIPDAU Level-1-260C-UNLIM

(1) The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability

information and additional product content details.

TBD: The Pb-Free/Green conversion plan has not been defined.

Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that

lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.

Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between

the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.

Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight

in homogeneous material)

(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information

provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and

continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.

TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

PACKAGE OPTION ADDENDUM

www.ti.com 20-Apr-2011

Addendum-Page 2

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

OTHER QUALIFIED VERSIONS OF UCC28070 :

•Automotive: UCC28070-Q1

NOTE: Qualified Version Definitions:

•Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device Package

Type Package

Drawing Pins SPQ Reel

Diameter

(mm)

Reel

Width

W1 (mm)

(mm) B0

(mm) K0

(mm) P1

(mm) W

(mm) Pin1

Quadrant

UCC28070DWR SOIC DW 20 2000 330.0 24.4 10.8 13.0 2.7 12.0 24.0 Q1

UCC28070PWR TSSOP PW 20 2000 330.0 16.4 6.95 7.1 1.6 8.0 16.0 Q1

PACKAGE MATERIALS INFORMATION

www.ti.com 14-Jul-2012

Pack Materials-Page 1

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

UCC28070DWR SOIC DW 20 2000 367.0 367.0 45.0

UCC28070PWR TSSOP PW 20 2000 367.0 367.0 38.0

PACKAGE MATERIALS INFORMATION

www.ti.com 14-Jul-2012

Pack Materials-Page 2

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