LT8300MPS5#TRPBF - LINEAR TECHNOLOGY

LT8300

8300f

Typical applicaTion

FeaTures DescripTion

100VIN Micropower Isolated

Flyback Converter with

150V/260mA Switch

The LT

8300 is a micropower high voltage isolated flyback

converter. By sampling the isolated output voltage directly

from the primary-side flyback waveform, the part requires

no third winding or opto-isolator for regulation. The output

voltage is programmed with a single external resistor. In-

ternal compensation and soft-start further reduce external

component count. Boundary mode operation provides a

small magnetic solution with excellent load regulation.

Low ripple Burst Mode operation maintains high efficiency

at light load while minimizing the output voltage ripple.

A 260mA, 150V DMOS power switch is integrated along

with all high voltage circuitry and control logic into a 5-lead

ThinSOT™ package.

The LT8300 operates from an input voltages range of 6V

to 100V and can deliver up to 2W of isolated output power.

The high level of integration and the use of boundary

and low ripple burst modes result in a simple to use, low

component count, and high efficiency application solution

for isolated power delivery.

L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks

and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the

property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497,

and 7471522.

5V Micropower Isolated Flyback Converter

applicaTions

n 6V to 100V Input Voltage Range

n 260mA, 150V Internal DMOS Power Switch

n Low Quiescent Current:

70µA in Sleep Mode

330µA in Active Mode

n Boundary Mode Operation at Heavy Load

n Low-Ripple Burst Mode

Operation at Light Load

n Minimum Load <0.5% (Typ) of Full Output

n VOUT Set with a Single External Resistor

n No Transformer Third Winding or Opto-Isolator

Required for Regulation

n Accurate EN/UVLO Threshold and Hysteresis

n Internal Compensation and Soft-Start

n 5-Lead TSOT-23 Package

n Isolated Telecom, Automotive, Industrial, Medical

Power Supplies

n Isolated Auxiliary/Housekeeping Power Supplies

Efficiency vs Load Current

LT8300

4:1

RFB

300µH 19µH

EN/UVLO

2.2µF

40.2k

VIN

36V TO 72V

VOUT+

1mA TO 300mA

VOUT–

GND

210k

•

47µF

8300 TA01a

LOAD CURRENT (mA)

EFFICIENCY (%)

100

8300 TA01b

300250200150100500

VIN = 48V

VIN = 72V

VIN = 36V

LT8300

8300f

pin conFiguraTionabsoluTe MaxiMuM raTings

SW (Note 2) ........................................................... 150V

VIN ......................................................................... 100V

EN/UVLO ................................................................... VIN

RFB ...................................................... VIN – 0.5V to VIN

Current into RFB ................................................... 200µA

Operating Junction Temperature Range (Notes 3, 4)

LT8300E, LT8300I ............................. –40°C to 125°C

LT8300H ............................................ –40°C to 150°C

LT8300MP ......................................... –55°C to 150°C

Storage Temperature Range .................. –65°C to 150°C

(Note 1)

EN/UVLO 1

GND 2

TOP VIEW

S5 PACKAGE

5-LEAD PLASTIC TSOT-23

RFB 3

5 VIN

4 SW

TJMAX = 150°C, θJA = 150°C/W

orDer inForMaTion

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LT8300ES5#PBF LT8300ES5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 125°C

LT8300IS5#PBF LT8300IS5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 125°C

LT8300HS5#PBF LT8300HS5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 150°C

LT8300MPS5#PBF LT8300MPS5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –55°C to 150°C

Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.

Consult LTC Marketing for information on non-standard lead based finish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/

LT8300

8300f

elecTrical characTerisTics

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 2: The SW pin is rated to 150V for transients. Depending on the

leakage inductance voltage spike, operating waveforms of the SW pin

should be derated to keep the flyback voltage spike below 150V as shown

in Figure 5.

Note 3: The LT8300E is guaranteed to meet performance specifications

from 0°C to 125°C operating junction temperature. Specifications over

the –40°C to 125°C operating junction temperature range are assured by

design, characterization and correlation with statistical process controls.

The l denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, VEN/UVLO = VIN unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT

VIN Input Voltage Range 6 100 V

VIN UVLO Threshold Rising

Falling 5.8

3.2 6 V

IQVIN Quiescent Current VEN/UVLO = 0.3V

VEN/UVLO = 1.1V

Sleep Mode (Switch Off)

Active Mode (Switch On)

1.2

200

330

2 µA

µA

EN/UVLO Shutdown Threshold For Lowest Off IQl0.3 0.75 V

EN/UVLO Enable Threshold Falling

Hysteresis

l1.199 1.223

0.016 1.270 V

IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.3V

VEN/UVLO = 1.1V

VEN/UVLO = 1.3V

–0.1

2.2

–0.1

2.5

0.1

2.8

0.1

µA

fMAX Maximum Switching Frequency 720 750 780 kHz

fMIN Minimum Switching Frequency 6 7.5 9 kHz

tON(MIN) Minimum Switch-On Time 160 ns

tOFF(MIN) Minimum Switch-Off Time 350 ns

tOFF(MAX) Maximum Switch-Off Time Backup Timer 200 µs

ISW(MAX) Maximum SW Current Limit l228 260 292 mA

ISW(MIN) Minimum SW Current Limit l34 52 70 mA

SW Over Current Limit To Initiate Soft-Start 520 mA

RDS(ON) Switch On-Resistance ISW = 100mA 10 Ω

ILKG Switch Leakage Current VIN = 100V, VSW = 150V 0.1 0.5 µA

IRFB RFB Regulation Current l98 100 102 µA

RFB Regulation Current Line Regulation 6V ≤ VIN ≤ 100V 0.001 0.01 %/V

tSS Soft-Start Timer 2.7 ms

The LT8300I is guaranteed over the full –40°C to 125°C operating junction

temperature range. The LT8300H is guaranteed over the full –40°C to

150°C operating junction temperature range. The LT8300MP is guaranteed

over the full –55°C to 150°C operating junction temperature range. High

junction temperatures degrade operating lifetimes. Operating lifetime is

derated at junction temperature greater than 125°C.

Note 4: The LT8300 includes overtemperature protection that is intended

to protect the device during momentary overload conditions. Junction

temperature will exceed 150°C when overtemperature protection is active.

Continuous operation above the specified maximum operating junction

temperature may impair device reliability.

LT8300

8300f

Typical perForMance characTerisTics

Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Waveforms

VIN Shutdown Current

VIN Quiescent Current,

Sleep Mode

VIN Quiescent Current,

Active Mode

Output Load and Line Regulation Output Temperature Variation

Switching Frequency

vs Load Current

TA = 25°C, unless otherwise noted.

LOAD CURRENT (mA)

OUTPUT VOLTAGE (V)

5.20

4.85

5.15

4.90

5.00

5.05

5.10

4.95

4.80

8300 G01

300250200150100500

VIN = 36V

VIN = 48V

VIN = 72V

FRONT PAGE APPLICATION

AMBIENT TEMPERATURE (°C)

OUTPUT VOLTAGE (V)

5.5

4.6

5.4

4.7

5.0

5.1

5.2

5.3

4.8

4.9

4.5

8300 G02

1507550 125100250–25–50

FRONT PAGE APPLICATION

VIN = 48V, IOUT = 200mA

LOAD CURRENT (mA)

FREQUENCY (kHz)

500

100

200

400

300

8300 G03

300250200150100500

FRONT PAGE APPLICATION

VIN = 48V

2µs/DIV

FRONT PAGE APPLICATION

VIN = 48V, IOUT = 300mA

ILPRI

100mA/DIV

VSW

50V/DIV

VOUT

50mV/DIV

8300 G04 2µs/DIV

FRONT PAGE APPLICATION

VIN = 48V, IOUT = 60mA

ILPRI

100mA/DIV

VSW

50V/DIV

VOUT

50mV/DIV

8300 G05 20µs/DIV

FRONT PAGE APPLICATION

VIN = 48V, IOUT = 1mA

ILPRI

100mA/DIV

VSW

50V/DIV

VOUT

50mV/DIV

8300 G06

VIN (V)

IQ (µA)

8300 G07

100806040200

TJ = –55°C

TJ = 150°C TJ = 25°C

VIN (V)

IQ (µA)

100

8300 G08

100806040200

TJ = 25°C

TJ = –55°C

TJ = 150°C

VIN (V)

IQ (µA)

380

300

320

360

340

280

8300 G09

100806040200

TJ = 25°C

TJ = –55°C

TJ = 150°C

LT8300

8300f

Typical perForMance characTerisTics

RDS(ON) Switch Current Limit Maximum Switching Frequency

Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time

EN/UVLO Enable Threshold EN/UVLO Hysteresis Current RFB Regulation Current

TA = 25°C, unless otherwise noted.

TEMPERATURE (°C)

EN/UVLO

(V)

1.240

1.205

1.210

1.225

1.230

1.235

1.215

1.220

1.200

8300 G10

1507550 125100250–25–50

TEMPERATURE (°C)

HYS

(µA)

8300 G11

1507550 125100250–25–50

TEMPERATURE (°C)

RFB

(µA)

105

101

102

103

104

100

8300 G12

1507550 125100250–25–50

TEMPERATURE (°C)

RESISTANCE (Ω)

8300 G13

1507550 125100250–25–50

TEMPERATURE (°C)

(mA)

300

100

150

200

250

8300 G14

1507550 125100250–25–50

MAXIMUM CURRENT LIMIT

MINIMUM CURRENT LIMIT

TEMPERATURE (°C)

FREQUENCY (kHz)

1000

200

400

600

800

8300 G15

1507550 125100250–25–50

TEMPERATURE (°C)

FREQUENCY (kHz)

8300 G16

1507550 125100250–25–50

TEMPERATURE (°C)

TIME (ns)

400

100

200

300

8300 G17

1507550 125100250–25–50

TEMPERATURE (°C)

TIME (ns)

400

100

200

300

8300 G18

1507550 125100250–25–50

LT8300

8300f

pin FuncTions

EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The

EN/UVLO pin is used to enable the LT8300. Pull the pin

below 0.3V to shut down the LT8300. This pin has an ac-

curate 1.223V threshold and can be used to program a VIN

undervoltage lockout (UVLO) threshold using a resistor

divider from VIN to ground. A 2.5µA current hysteresis

allows the programming of VIN UVLO hysteresis. If neither

function is used, tie this pin directly to VIN.

GND (Pin 2): Ground. Tie this pin directly to local ground

plane.

RFB (Pin 3): Input Pin for External Feedback Resistor.

Connect a resistor from this pin to the transformer pri-

mary SW pin. The ratio of the RFB resistor to the internal

trimmed 12.23k resistor, times the internal bandgap

reference, determines the output voltage (plus the effect

of any non-unity transformer turns ratio). Minimize trace

area at this pin.

SW (Pin 4): Drain of the 150V Internal DMOS Power

Switch. Minimize trace area at this pin to reduce EMI and

voltage spikes.

VIN (Pin 5): Input Supply. The VIN pin supplies current

to internal circuitry and serves as a reference voltage for

the feedback circuitry connected to the RFB pin. Locally

bypass this pin to ground with a capacitor.

LT8300

8300f

block DiagraM

8300 BD

–

OSCILLATOR

1:4

R Q

1.223V

25µA

M2M3

BOUNDARY

DETECTOR

DRIVER

–

RSENSE

0.3Ω

RREF

12.23kΩ

RFB

2.5µA

EN/UVLO

3 45

–

1.223V

REFERENCE

REGULATORS

VIN

GND

RFB SWVIN

VIN

NPS:1 DOUT

LSEC

LPRI

VOUT+

VOUT–

•

COUT

CIN

LT8300

8300f

operaTion

The LT8300 is a current mode switching regulator IC de-

signed specially for the isolated flyback topology. The key

problem in isolated topologies is how to communicate the

output voltage information from the isolated secondary

side of the transformer to the primary side for regulation.

Historically, opto-isolators or extra transformer windings

communicate this information across the isolation bound-

ary. Opto-isolator circuits waste output power, and the

extra components increase the cost and physical size of

the power supply. Opto-isolators can also cause system

issues due to limited dynamic response, nonlinearity, unit-

to-unit variation and aging over lifetime. Circuits employing

extra transformer windings also exhibit deficiencies, as

using an extra winding adds to the transformer’s physical

size and cost, and dynamic response is often mediocre.

The LT8300 samples the isolated output voltage through

the primary-side flyback pulse waveform. In this manner,

neither opto-isolator nor extra transformer winding is re-

quired for regulation. Since the LT8300 operates in either

boundary conduction mode or discontinuous conduction

mode, the output voltage is always sampled on the SW

pin when the secondary current is zero. This method im-

proves load regulation without the need of external load

compensation components.

The LT8300 is a simple to use micropower isolated flyback

converter housed in a 5-lead TSOT-23 package. The output

voltage is programmed with a single external resistor. By

integrating the loop compensation and soft-start inside, the

part further reduces the number of external components.

As shown in the Block Diagram, many of the blocks are

similar to those found in traditional switching regulators

including reference, regulators, oscillator, logic, current

amplifier, current comparator, driver, and power switch.

The novel sections include a flyback pulse sense circuit,

a sample-and-hold error amplifier, and a boundary mode

detector, as well as the additional logic for boundary

conduction mode, discontinuous conduction mode, and

low ripple Burst Mode operation.

Boundary Conduction Mode Operation

The LT8300 features boundary conduction mode operation

at heavy load, where the chip turns on the primary power

switch when the secondary current is zero. Boundary

conduction mode is a variable frequency, variable peak-

current switching scheme. The power switch turns on

and the transformer primary current increases until an

internally controlled peak current limit. After the power

switch turns off, the voltage on the SW pin rises to the

output voltage multiplied by the primary-to-secondary

transformer turns ratio plus the input voltage. When the

secondary current through the output diode falls to zero,

the SW pin voltage collapses and rings around VIN. A

boundary mode detector senses this event and turns the

power switch back on.

Boundary conduction mode returns the secondary current

to zero every cycle, so parasitic resistive voltage drops

do not cause load regulation errors. Boundary conduc-

tion mode also allows the use of smaller transformers

compared to continuous conduction mode and does not

exhibit sub-harmonic oscillation.

Discontinuous Conduction Mode Operation

As the load gets lighter, boundary conduction mode in-

creases the switching frequency and decreases the switch

peak current at the same ratio. Running at a higher switching

frequency up to several MHz increases switching and gate

charge losses. To avoid this scenario, the LT8300 has an

additional internal oscillator, which clamps the maximum

switching frequency to be less than 750kHz. Once the

switching frequency hits the internal frequency clamp,

the part starts to delay the switch turn-on and operates

in discontinuous conduction mode.

Low Ripple Burst Mode Operation

Unlike traditional flyback converters, the LT8300 has to

turn on and off at least for a minimum amount of time

and with a minimum frequency to allow accurate sampling

of the output voltage. The inherent minimum switch cur-

rent limit and minimum switch-off time are necessary to

guarantee the correct operation of specific applications.

As the load gets very light, the LT8300 starts to fold back

the switching frequency while keeping the minimum switch

current limit. So the load current is able to decrease while

still allowing minimum switch-off time for the sample-

and-hold error amplifier. Meanwhile, the part switches

between sleep mode and active mode, thereby reducing the

LT8300

8300f

operaTion

effective quiescent current to improve light load efficiency.

In this condition, the LT8300 operates in low ripple Burst

Mode. The typical 7.5kHz minimum switching frequency

Output Voltage

The RFB resistor as depicted in the Block Diagram is the

only external resistor used to program the output voltage.

The LT8300 operates similar to traditional current mode

switchers, except in the use of a unique flyback pulse

sense circuit and a sample-and-hold error amplifier, which

sample and therefore regulate the isolated output voltage

from the flyback pulse.

Operation is as follows: when the power switch M1 turns

off, the SW pin voltage rises above the VIN supply. The

amplitude of the flyback pulse, i.e., the difference between

the SW pin voltage and VIN supply, is given as:

VFLBK = (VOUT + VF + ISEC • ESR) • NPS

VF = Output diode forward voltage

ISEC = Transformer secondary current

ESR = Total impedance of secondary circuit

NPS = Transformer effective primary-to-secondary

turns ratio

The flyback voltage is then converted to a current IRFB by

the flyback pulse sense circuit (M2 and M3). This cur-

rent IRFB also flows through the internal trimmed 12.23k

RREF resistor to generate a ground-referred voltage. The

resulting voltage feeds to the inverting input of the sample-

and-hold error amplifier. Since the sample-and-hold error

amplifier samples the voltage when the secondary current

is zero, the (ISEC • ESR) term in the VFLBK equation can be

assumed to be zero.

The bandgap reference voltage VBG, 1.223V, feeds to the

non-inverting input of the sample-and-hold error ampli-

fier. The relatively high gain in the overall loop causes

the voltage across RREF resistor to be nearly equal to the

applicaTions inForMaTion

determines how often the output voltage is sampled and

also the minimum load requirement.

bandgap reference voltage VBG. The resulting relationship

between VFLBK and VBG can be expressed as:

VFLBK

RFB









•RREF =VBG

VFLBK =VBG

RREF









•RFB =IRFB •RFB

VBG = Bandgap reference voltage

IRFB = RFB regulation current = 100µA

Combination with the previous VFLBK equation yields an

equation for VOUT, in terms of the RFB resistor, transformer

turns ratio, and diode forward voltage:

VOUT =100µA •RFB

NPS









−VF

Output Temperature Coefficient

The first term in the VOUT equation does not have tempera-

ture dependence, but the output diode forward voltage VF

has a significant negative temperature coefficient (–1mV/°C

to –2mV/°C). Such a negative temperature coefficient pro-

duces approximately 200mV to 300mV voltage variation

on the output voltage across temperature.

For higher voltage outputs, such as 12V and 24V, the output

diode temperature coefficient has a negligible effect on the

output voltage regulation. For lower voltage outputs, such

as 3.3V and 5V, however, the output diode temperature

coefficient does count for an extra 2% to 5% output voltage

regulation. For customers requiring tight output voltage

regulation across temperature, please refer to other LTC

parts with integrated temperature compensation features.

LT8300

8300f

applicaTions inForMaTion

Selecting Actual RFB Resistor Value

The LT8300 uses a unique sampling scheme to regulate

the isolated output voltage. Due to the sampling nature,

the scheme contains repeatable delays and error sources,

which will affect the output voltage and force a re-evaluation

of the RFB resistor value. Therefore, a simple two-step

process is required to choose feedback resistor RFB.

Rearrangement of the expression for VOUT in the Output

Voltage section yields the starting value for RFB:

RFB =NPS •VOUT +VF

( )

100µA

VOUT = Output voltage

VF = Output diode forward voltage = ~0.3V

NPS = Transformer effective primary-to-secondary

turns ratio

Power up the application with the starting RFB value and

other components connected, and measure the regulated

output voltage, VOUT(MEAS). The final RFB value can be

adjusted to:

RFB(FINAL) =

OUT

VOUT(MEAS)

•RFB

Once the final RFB value is selected, the regulation accuracy

from board to board for a given application will be very

consistent, typically under ±5% when including device

variation of all the components in the system (assuming

resistor tolerances and transformer windings matching

within ±1%). However, if the transformer or the output

diode is changed, or the layout is dramatically altered,

there may be some change in VOUT.

Output Power

A flyback converter has a complicated relationship between

the input and output currents compared to a buck or a

boost converter. A boost converter has a relatively constant

maximum input current regardless of input voltage and a

buck converter has a relatively constant maximum output

current regardless of input voltage. This is due to the

continuous non-switching behavior of the two currents. A

flyback converter has both discontinuous input and output

currents which make it similar to a non-isolated buck-boost

converter. The duty cycle will affect the input and output

currents, making it hard to predict output power. In ad-

dition, the winding ratio can be changed to multiply the

output current at the expense of a higher switch voltage.

The graphs in Figures 1 to 4 show the typical maximum

output power possible for the output voltages 3.3V, 5V,

12V, and 24V. The maximum output power curve is the

calculated output power if the switch voltage is 120V dur-

ing the switch-off time. 30V of margin is left for leakage

inductance voltage spike. To achieve this power level at

a given input, a winding ratio value must be calculated

to stress the switch to 120V, resulting in some odd ratio

values. The curves below the maximum output power

curve are examples of common winding ratio values and

the amount of output power at given input voltages.

One design example would be a 5V output converter with

a minimum input voltage of 36V and a maximum input

voltage of 72V. A six-to-one winding ratio fits this design

example perfectly and outputs equal to 2.44W at 72V but

lowers to 1.87W at 36V.

The following equations calculate output power:



POUT = η•VIN •D•ISW(MAX) •0.5

η = Efficiency =85%

D=DutyCycle =VOUT +VF

( )

•NPS

VOUT +VF

( )

•NPS +VIN

ISW(MAX) = Maximum switch current limit = 260mA

LT8300

8300f

applicaTions inForMaTion

Figure 4. Output Power for 24V Output

Figure 1. Output Power for 3.3V Output Figure 2. Output Power for 5V Output

Figure 3. Output Power for 12V Output

INPUT VOLTAGE (V)

OUTPUT POWER (W)

3.5

2.5

1.5

3.0

2.0

1.0

0.5

040 8020 60

8300 F01

100

N = 12:1

MAXIMUM

OUTPUT

POWER

N = 8:1

N = 6:1

N = 4:1

INPUT VOLTAGE (V)

OUTPUT POWER (W)

3.5

2.5

1.5

3.0

2.0

1.0

0.5

040 8020 60

8300 F02

100

MAXIMUM

OUTPUT

POWER

N = 2:1

N = 4:1

N = 6:1

N = 8:1

INPUT VOLTAGE (V)

OUTPUT POWER (W)

3.5

2.5

1.5

3.0

2.0

1.0

0.5

040 8020 60

8300 F03

100

N = 4:1

MAXIMUM

OUTPUT

POWER

N = 3:1

N = 2:1

N = 1:1

INPUT VOLTAGE (V)

OUTPUT POWER (W)

3.5

2.5

1.5

3.0

2.0

1.0

0.5

040 8020 60

8300 F04

100

N = 1:1

N = 3:2

N = 1:2

MAXIMUM

OUTPUT

POWER

N = 2:1

Primary Inductance Requirement

The LT8300 obtains output voltage information from the

reflected output voltage on the SW pin. The conduction

of secondary current reflects the output voltage on the

primary SW pin. The sample-and-hold error amplifier needs

a minimum 350ns to settle and sample the reflected output

voltage. In order to ensure proper sampling, the second-

ary winding needs to conduct current for a minimum of

350ns. The following equation gives the minimum value

for primary-side magnetizing inductance:

LPRI ≥tOFF(MIN) •NPS •VOUT +VF

( )

ISW(MIN)

tOFF(MIN) = Minimum switch-off time = 350ns

ISW(MIN) = Minimum switch current limit = 52mA

In addition to the primary inductance requirement for

the minimum switch-off time, the LT8300 has minimum

switch-on time that prevents the chip from turning on

the power switch shorter than approximately 160ns. This

minimum switch-on time is mainly for leading-edge blank-

ing the initial switch turn-on current spike. If the inductor

current exceeds the desired current limit during that time,

oscillation may occur at the output as the current control

loop will lose its ability to regulate. Therefore, the following

equation relating to maximum input voltage must also be

followed in selecting primary-side magnetizing inductance:

LPRI ≥tON(MIN) •VIN(MAX)

ISW(MIN)

tON(MIN) = Minimum Switch-On Time = 160ns

LT8300

8300f

applicaTions inForMaTion

In general, choose a transformer with its primary mag-

netizing inductance about 20% to 40% larger than the

minimum values calculated above. A transformer with

much larger inductance will have a bigger physical size

and may cause instability at light load.

Selecting a Transformer

Transformer specification and design is perhaps the most

critical part of successfully applying the LT8300. In addition

to the usual list of guidelines dealing with high frequency

isolated power supply transformer design, the following

information should be carefully considered.

Linear Technology has worked with several leading mag-

netic component manufacturers to produce pre-designed

flyback transformers for use with the LT8300. Table 1

shows the details of these transformers.

Turns Ratio

Note that when choosing the RFB resistor to set output

voltage, the user has relative freedom in selecting a trans-

former turns ratio to suit a given application. In contrast,

the use of simple ratios of small integers, e.g., 4:1, 2:1,

1:1, provides more freedom in settling total turns and

mutual inductance.

Table 1. Predesigned Transformers — Typical Specifications

TRANSFORMER

PART NUMBER

LPRI

(µH)

LLEAKAGE

(µH) NP:NS:NB VENDOR TARGET APPLICATIONS

750312367 400 4.5 8:1 Würth Elektronik 48V to 3.3V/0.51A, 24V to 3.3V/0.37A, 12V to 3.3V/0.24A

750312557 300 2.5 6:1 Würth Elektronik 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A

48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A

750312365 300 1.8 4:1 Würth Elektronik 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A

750312558 300 1.75 2:1:1 Würth Elektronik 48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA

48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA

750312559 300 2 1:1 Würth Elektronik 48V to 24V/67mA, 24V to 24V/50mA, 12V to 24V/33mA

750311019 400 5 6:1:2 Würth Elektronik 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A

48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A

750311558 300 1.5 4:1:1 Würth Elektronik 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A

750311660 350 3 2:1:0.33 Würth Elektronik 48V to 12V/0.134A, 24V to 12V/0.1A, 12V to 12V/0.066A

48V to 15V/0.124A, 24V to 15V/0.088A, 12V to 15V/0.056A

750311838 350 3 2:1:1 Würth Elektronik 48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA

48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA

750311659 300 2 1:1:0.2 Würth Elektronik 48V to 24V/67mA, 24V to 24V/50mA, 12V to 24V/33mA

10396-T026 300 2.5 6:1:2 Sumida 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A

48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A

10396-T024 300 2 4:1:1 Sumida 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A

10396-T022 300 2 2:1:0.33 Sumida 48V to 12V/0.134A, 24V to 12V/0.1A, 12V to 12V/0.066A

48V to 15V/0.124A, 24V to 15V/0.088A, 12V to 15V/0.056A

10396-T028 300 2.5 2:1:1 Sumida 48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA

48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA

L10-0116 500 7.3 6:1 BH Electronics 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A

48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A

L10-0112 230 3.38 4:1 BH Electronics 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A

L11-0067 230 2.16 4:1 BH Electronics 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A

* All the transformers are rated for 1.5kV Isolation.

LT8300

8300f

Typically, choose the transformer turns ratio to maximize

available output power. For low output voltages (3.3V or

5V), a larger N:1 turns ratio can be used with multiple

primary windings relative to the secondary to maximize the

transformer’s current gain (and output power). However,

remember that the SW pin sees a voltage that is equal

to the maximum input supply voltage plus the output

voltage multiplied by the turns ratio. In addition, leakage

inductance will cause a voltage spike (VLEAKAGE) on top of

this reflected voltage. This total quantity needs to remain

below the 150V absolute maximum rating of the SW pin

to prevent breakdown of the internal power switch. To-

gether these conditions place an upper limit on the turns

ratio, NPS, for a given application. Choose a turns ratio

low enough to ensure:

NPS <

150V −V

IN(MAX)

−V

LEAKAGE

VOUT +VF

For lower output power levels, choose a smaller N:1 turns

ratio to alleviate the SW pin voltage stress. Although a

1:N turns ratio makes it possible to have very high output

voltages without exceeding the breakdown voltage of the

internal power switch, the multiplied parasitic capacitance

through turns ratio coupled with the relatively resistive

150V internal power switch may cause the switch turn-on

current spike ringing beyond 160ns leading-edge blanking,

thereby producing light load instability in certain applica-

tions. So any 1:N turns ratio should be fully evaluated

before its use with the LT8300.

The turns ratio is an important element in the isolated

feedback scheme, and directly affects the output voltage

accuracy. Make sure the transformer manufacturer speci-

fies turns ratio accuracy within ±1%.

applicaTions inForMaTion

Saturation Current

The current in the transformer windings should not exceed

its rated saturation current. Energy injected once the core is

saturated will not be transferred to the secondary and will

instead be dissipated in the core. When designing custom

transformers to be used with the LT8300, the saturation

current should always be specified by the transformer

manufacturers.

Winding Resistance

Resistance in either the primary or secondary windings

will reduce overall power efficiency. Good output voltage

regulation will be maintained independent of winding re-

sistance due to the boundary/discontinuous conduction

mode operation of the LT8300.

Leakage Inductance and Snubbers

Transformer leakage inductance on either the primary or

secondary causes a voltage spike to appear on the primary

after the power switch turns off. This spike is increasingly

prominent at higher load currents where more stored en-

ergy must be dissipated. It is very important to minimize

transformer leakage inductance.

When designing an application, adequate margin should

be kept for the worst-case leakage voltage spikes even

under overload conditions. In most cases shown in Figure

5, the reflected output voltage on the primary plus VIN

should be kept below 120V. This leaves at least 30V margin

for the leakage spike across line and load conditions. A

larger voltage margin will be required for poorly wound

transformers or for excessive leakage inductance.

In addition to the voltage spikes, the leakage inductance

also causes the SW pin ringing for a while after the power

switch turns off. To prevent the voltage ringing falsely trig-

ger boundary mode detector, the LT8300 internally blanks

the boundary mode detector for approximately 250ns. Any

remaining voltage ringing after 250ns may turn the power

switch back on again before the secondary current falls

to zero. So the leakage inductance spike ringing should

be limited to less than 250ns.

LT8300

8300f

applicaTions inForMaTion

A snubber circuit is recommended for most applications.

Two types of snubber circuits shown in Figure 6 that can

protect the internal power switch include the DZ (diode-

Zener) snubber and the RC (resistor-capacitor) snubber. The

DZ snubber ensures well defined and consistent clamping

voltage and has slightly higher power efficiency, while the

RC snubber quickly damps the voltage spike ringing and

provides better load regulation and EMI performance.

Figure 5 shows the flyback waveforms with the DZ and

RC snubbers.

For the DZ snubber, proper care must be taken when

choosing both the diode and the Zener diode. Schottky

diodes are typically the best choice, but some PN diodes

can be used if they turn on fast enough to limit the leak-

age inductance spike. Choose a diode that has a reverse-

voltage rating higher than the maximum SW pin voltage.

The Zener diode breakdown voltage should be chosen to

balance power loss and switch voltage protection. The best

compromise is to choose the largest voltage breakdown.

Use the following equation to make the proper choice:

VZENER(MAX) ≤ 150V – VIN(MAX)

For an application with a maximum input voltage of 72V,

choose a 68V Zener diode, the VZENER(MAX) of which is

around 72V and below the 78V maximum.

The power loss in the clamp will determine the power rat-

ing of the Zener diode. Power loss in the clamp is highest

at maximum load and minimum input voltage. The switch

current is highest at this point along with the energy stored

in the leakage inductance. A 0.5W Zener will satisfy most

applications when the highest VZENER is chosen.

Figure 5. Maximum Voltages for SW Pin Flyback Waveform

Figure 6. Snubber Circuits

8300 F05

VSW

tOFF > 350ns

VLEAKAGE

tSP < 250ns

VSW VSW

TIME

No Snubber with DZ Snubber with RC Snubber

tOFF > 350ns

VLEAKAGE

tSP < 250ns

TIME

tOFF > 350ns

VLEAKAGE

tSP < 250ns

TIME

<150V

<120V

<150V

<120V

<150V

<120V

8300 F06b8300 F06a

DZ Snubber RC Snubber

Lℓ

•

Lℓ

•

LT8300

8300f

applicaTions inForMaTion

Tables 2 and 3 show some recommended diodes and

Zener diodes.

Table 2. Recommended Zener Diodes

PART

VZENER

(V)

POWER

(W) CASE VENDOR

MMSZ5266BT1G 68 0.5 SOD-123 On Semi

MMSZ5270BT1G 91 0.5 SOD-123

CMHZ5266B 68 0.5 SOD-123 Central

Semiconductor

CMHZ5267B 75 0.5 SOD-123

BZX84J-68 68 0.5 SOD323F NXP

BZX100A 100 0.5 SOD323F

Table 3. Recommended Diodes

PART I (A)

VREVERSE

(V) CASE VENDOR

BAV21W 0.625 200 SOD-123 Diodes Inc.

BAV20W 0.625 150 SOD-123

The recommended approach for designing an RC snubber

is to measure the period of the ringing on the SW pin when

the power switch turns off without the snubber and then

add capacitance (starting with 100pF) until the period of

the ringing is 1.5 to 2 times longer. The change in period

will determine the value of the parasitic capacitance, from

which the parasitic inductance can be determined from

the initial period, as well. Once the value of the SW node

capacitance and inductance is known, a series resistor can

be added to the snubber capacitance to dissipate power

and critically dampen the ringing. The equation for deriving

the optimal series resistance using the observed periods

( tPERIOD and tPERIOD(SNUBBED)) and snubber capacitance

(CSNUBBER) is:

CPAR =CSNUBBER

tPERIOD(SNUBBED)

tPERIOD











−1

LPAR =tPERIOD2

CPAR •4π2

RSNUBBER =LPAR

CPAR

Figure 7. Undervoltage Lockout (UVLO)

LT8300

GND

EN/UVLO

RUN/STOP

CONTROL

(OPTIONAL)

8300 F07

Note that energy absorbed by the RC snubber will be

converted to heat and will not be delivered to the load.

In high voltage or high current applications, the snubber

may need to be sized for thermal dissipation.

Undervoltage Lockout (UVLO)

A resistive divider from VIN to the EN/UVLO pin imple-

ments undervoltage lockout (UVLO). The EN/UVLO pin

falling threshold is set at 1.223V with 16mV hysteresis.

In addition, the EN/UVLO pin sinks 2.5µA when the volt-

age at the pin is below 1.223V. This current provides user

programmable hysteresis based on the value of R1. The

programmable UVLO thresholds are:

VIN(UVLO+)=1.239V •(R1

R2)

R2 +2.5µA •R1

VIN(UVLO−)=1.223V •(R1+R2)

Figure 7 shows the implementation of external shutdown

control while still using the UVLO function. The NMOS

grounds the EN/UVLO pin when turned on, and puts the

LT8300 in shutdown with quiescent current less than 2µA.

LT8300

8300f

Minimum Load Requirement

The LT8300 samples the isolated output voltage from the

primary-side flyback pulse waveform. The flyback pulse

occurs once the primary switch turns off and the secondary

winding conducts current. In order to sample the output

voltage, the LT8300 has to turn on and off at least for a

minimum amount of time and with a minimum frequency.

The LT8300 delivers a minimum amount of energy even

during light load conditions to ensure accurate output volt-

age information. The minimum energy delivery creates a

minimum load requirement, which can be approximately

estimated as:

ILOAD(MIN) =LPRI •ISW(MIN)

2•fMIN

2•VOUT

LPRI = Transformer primary inductance

ISW(MIN) = Minimum switch current limit = 52mA

fMIN = Minimum switching frequency = 7.5kHz

The LT8300 typically needs less than 0.5% of its full output

power as minimum load. Alternatively, a Zener diode with its

breakdown of 20% higher than the output voltage can serve

as a minimum load if pre-loading is not acceptable. For a 5V

output, use a 6V Zener with cathode connected to the output.

Output Short Protection

When the output is heavily overloaded or shorted, the

reflected SW pin waveform rings longer than the internal

blanking time. After the 350ns minimum switch-off time,

the excessive ring falsely trigger the boundary mode

detector and turn the power switch back on again before

the secondary current falls to zero. Under this condition,

the LT8300 runs into continuous conduction mode at

750kHz maximum switching frequency. Depending on the

VIN supply voltage, the switch current may run away and

exceed 260mA maximum current limit. Once the switch

current hits 520mA over current limit, a soft-start cycle

initiates and throttles back both switch current limit and

switch frequency. This output short protection prevents the

switch current from running away and limits the average

output diode current.

applicaTions inForMaTion

Design Example

Use the following design example as a guide to design

applications for the LT8300. The design example involves

designing a 12V output with a 120mA load current and an

input range from 36V to 72V.

VIN(MIN) = 36V, VIN(NOM) = 48V, VIN(MAX) = 72V,

VOUT = 12V, IOUT = 120mA

Step 1: Select the Transformer Turns Ratio.

NPS <150V

−

VIN(MAX)

−

VLEAKAGE

VOUT +VF

VLEAKAGE = Margin for transformer leakage spike = 30V

VF = Output diode forward voltage = ~0.3V

Example:

NPS <150V

−

72V

−

30V

12V

0.3V =3.9

The choice of transformer turns ratio is critical in deter-

mining output current capability of the converter. Table 4

shows the switch voltage stress and output current capa-

bility at different transformer turns ratio.

Table 4. Switch Voltage Stress and Output Current Capability

vs Turns Ratio

NPS

VSW(MAX) at

VIN(MAX) (V)

IOUT(MAX) at

VIN(MIN) (mA) DUTY CYCLE (%)

1:1 84.3 84 15-25

2:1 96.6 135 25-41

3:1 108.9 168 34-51

Since both NPS = 2 and NPS = 3 can meet the 120mA output

current requirement, NPS = 2 is chosen in this example

to allow more margin for transformer leakage inductance

voltage spike.

LT8300

8300f

applicaTions inForMaTion

Step 2: Determine the Primary Inductance.

Primary inductance for the transformer must be set above

a minimum value to satisfy the minimum switch-off and

switch-on time requirements:

LPRI ≥tOFF(MIN) •NPS •VOUT +VF

( )

ISW(MIN)

LPRI ≥tON(MIN) •VIN(MAX)

ISW(MIN)

tOFF(MIN) = 350ns

tON(MIN) = 160ns

ISW(MIN) = 52mA

Example:

LPRI ≥350ns •2•(12V +0.3V)

52mA =166µH

LPRI ≥160ns •72V

52mA

=222µH

Most transformers specify primary inductance with a toler-

ance of ±20%. With other component tolerance considered,

choose a transformer with its primary inductance 20% to

40% larger than the minimum values calculated above.

LPRI = 300µH is then chosen in this example.

Once the primary inductance has been determined, the

maximum load switching frequency can be calculated as:

fSW =

tON +tOFF

LPRI •ISW

VIN +LPRI •ISW

NPS •(VOUT +VF)

ISW =VOUT •IOUT •2

η•VIN •D

Example:

D=(12V

0.3V)•2

(12V +0.3V) •2+48V =0.34

ISW =12V •0.12A •2

0.85 •48V •0.34 =0.21A

fSW =260kHz

The transformer also needs to be rated for the correct

saturation current level across line and load conditions. A

saturation current rating larger than 400mA is necessary

to work with the LT8300. The 10396-T022 from Sumida

is chosen as the flyback transformer.

Step 3: Choose the Output Diode.

Two main criteria for choosing the output diode include

forward current rating and reverse voltage rating. The

maximum load requirement is a good first-order guess

as the average current requirement for the output diode.

A conservative metric is the maximum switch current limit

multiplied by the turns ratio,

IDIODE(MAX) = ISW(MAX) • NPS

Example:

IDIODE(MAX) = 0.52A

Next calculate reverse voltage requirement using maxi-

mum VIN:

VREVERSE =VOUT +VIN(MAX)

NPS

Example:

VREVERSE =12V +

72V

2=48V

The SBR0560S1 (0.5A, 60V diode) from Diodes Inc. is

chosen.

LT8300

8300f

Step 4: Choose the Output Capacitor.

The output capacitor should be chosen to minimize the

output voltage ripple while considering the increase in size

and cost of a larger capacitor. Use the equation below to

calculate the output capacitance:

COUT =LPRI •ISW2

2•VOUT •∆VOUT

Example:

Design for output voltage ripple less than 1% of VOUT,

i.e., 120mV.

COUT =300µH •(0.21A)2

2•12V •0.12V =4.6µF

Remember ceramic capacitors lose capacitance with ap-

plied voltage. The capacitance can drop to 40% of quoted

capacitance at the maximum voltage rating. So a 10uF, 16V

rating ceramic capacitor is chosen.

Step 5: Design Snubber Circuit.

The snubber circuit protects the power switch from leakage

inductance voltage spike. A DZ snubber is recommended

for this application because of lower leakage inductance

and larger voltage margin. The Zener and the diode need

to be selected.

The maximum Zener breakdown voltage is set according

to the maximum VIN:

VZENER(MAX) ≤ 150V – VIN(MAX)

Example:

VZENER(MAX) ≤ 150V – 72V = 78V

applicaTions inForMaTion

A 68V Zener with a maximum of 72V will provide optimal

protection and minimize power loss. So a 68V, 0.5W Zener

from On Semiconductor (MMSZ5266BT1G) is chosen.

Choose a diode that is fast and has sufficient reverse

voltage breakdown:

VREVERSE > VSW(MAX)

VSW(MAX) = VIN(MAX) + VZENER(MAX)

Example:

VREVERSE > 144V

A 150V, 0.6A diode from Diodes Inc. (BAV20W) is chosen.

Step 6: Select the RFB Resistor.

Use the following equation to calculate the starting value

for RFB:

RFB =NPS •(VOUT +VF)

100µA

Example:

RFB =2•(12V

0.3V)

100µA =246k

Depending on the tolerance of standard resistor values,

the precise resistor value may not exist. For 1% standard

values, a 243k resistor in series with a 3.01k resistor

should be close enough. As discussed in the Application

Information section, the final RFB value should be adjusted

on the measured output voltage.

LT8300

8300f

applicaTions inForMaTion

Step 7: Select the EN/UVLO Resistors.

Determine the amount of hysteresis required and calculate

R1 resistor value:

VIN(HYS) = 2.5µA • R1

Example:

Choose 2.5V of hysteresis,

R1 = 1M

Determine the UVLO thresholds and calculate R2 resistor

value:

VIN(UVLO+)=

1.239V •(R1+R2)

R2 +2.5µA •R1

Example:

Set VIN UVLO rising threshold to 34.5V,

R2 = 40.2k

VIN(UVLO+) = 34.1V

VIN(UVLO–) = 31.6V

Step 8: Ensure minimum load.

The theoretical minimum load can be approximately

estimated as:

ILOAD(MIN) =300µH •(52mA)2•7.5kHz

2•12V =0.25mA

Remember to check the minimum load requirement in real

application. The minimum load occurs at the point where

the output voltage begins to climb up as the converter

delivers more energy than what is consumed at the out-

put. The real minimum load for this application is about

0.6mA, 0.5% of 120mA maximum load. In this example,

a 20k resistor is selected as the minimum load.

LT8300

8300f

Typical applicaTions

LT8300

6:1

RFB

300µH 8µH

EN/UVLO

2.2µF

40.2k

VIN

36V TO 72V

VOUT+

1mA TO 330mA

VOUT–

GND

316k

T1: WÜRTH 750312557

D1: DIODES INC. SBR2A30P1

•

47µF

8300 TA02

5V Micropower Isolated Flyback Converter

12V Micropower Isolated Flyback Converter

LT8300

2:1

RFB

300µH 75µH

EN/UVLO

2.2µF

40.2k

VIN

36V TO 72V

VOUT+

12V

0.6mA TO 120mA

VOUT–

GND

243k

T1: SUMIDA 10396-TO22

D1: DIODES INC. SBR0560S1

•

10µF

8300 TA03

LT8300

8300f

Typical applicaTions

LT8300

8:1

RFB

400µH 6µH

EN/UVLO

2.2µF

40.2k

VIN

36V TO 72V

VOUT+

3.3V

2mA TO 440mA

VOUT–

GND

287k

T1: WÜRTH 750312367

D1: NXP PMEG2020EH

•

100µF

8300 TA05

24V Micropower Isolated Flyback Converter

3.3V Micropower Isolated Flyback Converter

LT8300

1:1

RFB

300µH 300µH

EN/UVLO

2.2µF

40.2k

VIN

36V TO 72V

VOUT+

24V

0.3mA TO 60mA

VOUT–

GND

243k

T1: WÜRTH 750311559

D1: DIODES DFLS 1200-7

•

4.7µF

8300 TA04

LT8300

8300f

Typical applicaTions

VIN to (VIN + 10V) Micropower Converter

VIN to (VIN – 10V) Micropower Converter

LT8300 D1

RFB

330µH

EN/UVLO

1µF

118k

VIN

15V TO 80V VOUT+

10V

100mA

VOUT–

GND

102k

L1: COILTRONICS DR73-331-R

D1: DIODES INC. SBR1U150SA

Z1: CENTRAL CMDZ12L

8300 TA07

4.7µF

LT8300 D1

RFB

330µH

EN/UVLO

1µF

118k

VIN

15V TO 80V

VOUT+

10V

50mA

VOUT–

GND

102k

L1: COILTRONICS DR73-331-R

D1: DIODES INC. SBR1U150SA

Z1: CENTRAL CMDZ12L

8300 TA06

4.7µF

LT8300

8300f

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

package DescripTion

S5 Package

5-Lead Plastic TSOT-23

(Reference LTC DWG # 05-08-1635 Rev B)

1.50 – 1.75

(NOTE 4)

2.80 BSC

0.30 – 0.45 TYP

5 PLCS (NOTE 3)

DATUM ‘A’

0.09 – 0.20

(NOTE 3) S5 TSOT-23 0302 REV B

PIN ONE

2.90 BSC

(NOTE 4)

0.95 BSC

1.90 BSC

0.80 – 0.90

1.00 MAX 0.01 – 0.10

0.20 BSC

0.30 – 0.50 REF

NOTE:

1. DIMENSIONS ARE IN MILLIMETERS

2. DRAWING NOT TO SCALE

3. DIMENSIONS ARE INCLUSIVE OF PLATING

4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR

5. MOLD FLASH SHALL NOT EXCEED 0.254mm

6. JEDEC PACKAGE REFERENCE IS MO-193

3.85 MAX

0.62

MAX

0.95

REF

RECOMMENDED SOLDER PAD LAYOUT

PER IPC CALCULATOR

1.4 MIN

2.62 REF

1.22 REF

Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.

LT8300

8300f

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com

 LINEAR TECHNOLOGY CORPORATION 2012

LT 0812 • PRINTED IN USA

relaTeD parTs

Typical applicaTion

3.3V Isolated Converter (Conforming to DEF-STAN61-5)

PART NUMBER DESCRIPTION COMMENTS

LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch,

MSOP-16(12)

LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No Opto Flyback , MSOP-16 with High Voltage Spacing

LT3798 Off-Line Isolated No Opto-Coupler Flyback Controller

with Active PFC VIN and VOUT Limited Only by External Components

LT3573/LT3574/LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch

LT3757/LT3759/LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive

LT3957/LT3958 40V/100V Flyback/Boost Converters Monolithic with Integrated 5A/3.3A Switch

LTC3803/LTC3803-3/

LTC3803-5 200kHz/300kHz Flyback Controllers in SOT-23 VIN and VOUT Limited by External Components

LTC3805/LTC3805-5 Adjustable Frequency Flyback Controllers VIN and VOUT Limited by External Components

VIN (V)

IVIN (µA)

400

200

300

100

8300 TA08b

323028262420 2218

LT8300

1:1 D1

RFB

150µH 150µH

EN/UVLO

1µF

93.1k

VIN

18V TO 32V

VOUT+

3.3V

0mA TO 20mA

VOUT–

GND

42.2k D1: DIODES INC. SBR0560S1-7

L1: DRQ73-151-R

Z1: CENTRAL CMDZ4L7

•

1µF 1µF

8300 TA08a

LT3009-3.3

GND

SHDN

IN OUT

Input Current with No Load