MIC3230YMLTR - Micrel, Inc.

MIC3230/1/2

Constant Current Boost Controller for

Driving High Power LEDs

Bringing the Power to Light™

Bringing the Power to Light is a trademark of Micrel, Inc.

MicroLead Frame and MLF are registered trademark of Amkor Technologies.

Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com

General Description

The MIC3230/1/2 are constant current boost switching

controllers specifically designed to power one or more

strings of high power LEDs. The MIC3230/1/2 have an

input voltage range from 6V to 45V and are ideal for a

variety of solid state lighting applications.

The MIC3230/1/2 utilizes an external power device which

offers a cost conscious solution for high power LED

applications. The powerful drive circuitry can deliver up to

70W to the LED system. Power consumption has been

minimized through the implementation of a 250mV

feedback voltage reference providing an accuracy of ±3%.

The MIC323x family is dimmable via a pulse width

modulated (PWM) input signal and also features an enable

pin for low power shutdown.

Multiple MIC3230 ICs can be synchronized to a common

operating frequency. The clocks of these synchronized

devices can be used together in order to help reduce noise

and errors in a system.

An external resistor sets the adjustable switching

frequency of the MIC3230/1. The switching frequency can

be between 100kHz and 1MHz. Setting the switching

frequency provides the mechanism by which a design can

be optimized for efficiency (performance) and size of the

external components (cost). The MIC323x family of LED

drivers also offer the following protection features: Over

voltage protection (OVP), thermal shutdown and under-

voltage lock-out (UVLO).

The MIC3231 offers a dither feature to assist in the

reduction of EMI. This is particularly useful in sensitive

EMI applications, and provides for a reduction or

emissions by approximately 10dB.

The MIC3232 is a 400kHz fixed frequency device offered

in a small MSOP-10 package. The MIC3230/1 are offered

in both the EPAD TSSOP-16 package and the

3mm × 3mm MLF®-12 package.

Data sheets and support documentation can be found on

Micrel’s web site at: www.micrel.com.

Features

• 6V to 45V input supply range

• Capable of driving up to 70W

• Ultra low EMI via dithering on the MIC3231

• Programmable LED drive current

• Feedback voltage = 250mV ±3%

• Programmable switching frequency (MIC3230/1) or

400kHz fixed frequency operation (MIC3232)

• PWM Dimming and separate enable shutdown

• Frequency synchronization with other MIC3230s

• Protection features:

Over Voltage Protection (OVP)

Over temperature protection

Under-voltage Lock-out (UVLO)

• Packages:

IADJ IS65

1VIN

PWMD

COMP

10 VDD

DRV

PGND

OVP

IADJ IS

VIN

PWMD

COMP

1 VDD

DRV

PGND

OVP

FS EPAD SYNC/NC6

1N/C

VIN

PWMD

COMP

IADJ

AGND

16 N/C

VDD

DRV

PGND

OVP

SYNC/NC

N/C

8EPAD

MIC3232

MSOP-10 MIC3230/1

MLF-12 MIC3230/1

TSSOP-16

• –40°C to +125°C junction temperature range

Applications

• Street Lighting

• Solid State Lighting

• General Illumination

• Architectural Lighting

• Constant Current Power Supplies

_________________________________________________________________________________________________

January 2009

M9999-011409-A

(408) 955-1690

Micrel, Inc. MIC3230/1/2

January 2009 2 M9999-011409-A

Typical Application

47µH D1

100k

4.33k

COUT

4.7µF

100V

100k

RADJ

1/4W

RFS

16.5k CCOMP

10nF

CIN

4.7µF/50v

RSLC

VFB = 0.25V RCS

1/2W

Analog ground Power ground

VOUTVIN

PWMD

ENABLE

Synch to other MIC3230

ILED Return

LED 1

LED X

COMP

PWMD

VDD

AGND PGND

EPAD

OVP

DRV

VIN

IADJ

MIC3230/31

SYNC

10µF

10V

Figure 1. Typical Application of the MIC3230 LED Driver

Product Option Matrix

MIC3230 MIC3231 MIC3232

Input Voltage 6V to 45V 6V to 45V 6V to 45V

Synchronization Yes No No

Dither No Yes No

Frequency Range Adj from 100kHz to 1MHz Adj from 100kHz to 1MHz Fixed Freq. = 400kHz

Package EPAD TSSOP-16

3mm × 3mm MLF®-12

EPAD TSSOP-16

3mm × 3mm MLF®-12 MSOP-10

Ordering Information

Part Number Temperature Range Package Lead Finish

MIC3230YTSE –40° to +125°C EPAD TSSOP-16 Pb-Free

MIC3230YML –40° to +125°C 3mm x 3mm MLF®-12L Pb-Free

MIC3231YTSE –40° to +125°C EPAD TSSOP-16 Pb-Free

MIC3231YML –40° to +125°C 3mm x 3mm MLF®-12L Pb-Free

MIC3232YMM –40° to +125°C MSOP-10 Pb-Free

Micrel, Inc. MIC3230/1/2

January 2009 3 M9999-011409-A

Pin Configur ation

IADJ IS65

1VIN

PWMD

COMP

10 VDD

DRV

PGN

OVP

IADJ IS

VIN

PWMD

COMP

1 VDD

DRV

PGND

OVP

FS EPAD SYNC/NC6

1N/C

VIN

PWMD

COMP

IADJ

AGND

16 N/C

VDD

DRV

PGND

OVP

SYNC/NC

N/C

8EPAD

MSOP-10 (MM)

MIC3232

3mmx3mmMLF®-12L (ML)

MIC3230, MIC3231

See Product Option Matrix for selection

TSSOP-16 (TSE)

MIC3230, MIC3231

See Product Option Matrix for selection

Pin Description

Pin Number

3x3MLF

Pin Number

TSSOP-16L

Pin Number

MSOP-10L

Pin Name Pin Function

-- 1 -- NC No Connect

1 2 1 VIN Input Voltage (power) 6V to 45V

2 3 2 EN

Enable Control (Input). Logic High (1.5V) enables

the regulator. Logic Low (0.4V) shuts down the

regulator. Connect a 100k resistor from EN to VIN.

3 4 3 PWMD

PWM input. High signal terminates the output

power. Low Signal starts up the output power.

4 5 4 COMP Compensation (output): for external compensation

5 6 5 IADJ Feedback (input)

6 7 -- FS

Frequency Select (input). Connected to a Resistor

to determine the operating frequency

-- 8 -- AGND Analog Ground

-- 9 -- NC No Connect

7 10 -- SYNC

Sync (output). Connect to another MIC3230 to

synchronize multiple converters.

8 11 6 IS

Current Sense (input). Connected to external

current sense resistor which in turn is connected to

the source of the external FET as well as an external

slope compensation resistor

9 12 7 OVP

OVP divider connection (output). Connect the top of

the divider string to the output. If the load is

disconnected, the output voltage will rise until OVP

reaches 1.25V and then will regulate around this

point

10 13 8 PGND Power Ground

11 14 9 DRV

Drive Output: connect to the gate of external FET

(output)

12 15 10 VDD

VDD Filter for internal power rail. Do not connect an

external load to this pin. Connect 10µF to GND.

-- 16 -- NC No Connect

-- -- -- EPAD Connect to AGND

Micrel, Inc. MIC3230/1/2

January 2009 4 M9999-011409-A

Absolute Maximum Ratings(1)

Supply Voltage (VIN).....................................................+48V

Enable Pin Voltage........................................... -0.3V to +6V

IADJ Voltage ..................................................................+6V

Lead Temperature (soldering, #sec.)......................... 260°C

Storage Temperature (Ts)..........................-65°C to +150°C

ESD Rating(3)..................... MIC3230= 1500V HB, 100VMM

.........................................MIC3232= 2kV HB, 100VMM

.................................... MIC3231= 1500V HB, 150VMM

Operating Ratings(2)

Supply Voltage (VIN)......................................... +6V to +45V

Junction Temperature (TJ)........................ –40°C to +125°C

Junction Thermal Resistance

MSOP-10 (θJA) ..............................................130.5°C/W

EPAD TSSOP-16 (θJA) ...................................36.5°C/W

3mmx3mm MLF®-12L (θJA).............................60.7°C/W

Electrical Characteristics(4)

VIN = 12V; VEN = 3.6V; L = 47µH; C = 4.7µF; TJ = 25°C, Bold values indicate –40°C TJ  +125°C, unless noted.

Symbol Parameter Condition Min Typ Max Units

VIN Supply Voltage Range 6 45 V

UVLO Under Voltage Lockout 3.5 4.9 5.5 V

IVIN Quiescent Current VFB > 275mV (to ensure device is not

switching)

3.2 10 mA

ISD Shutdown Current VEN = 0V 30 µA

Room temperature (3%) 242.5 250 257.5 mV

VIADJ Feedback Voltage (at IADJ)

–40°C TJ  +125°C (5%) 237.5 250 262.5 mV

IADJ Feedback Input Current VFB = 250mV 1.2 3 µA

Line Regulation VIN = 12V to 24V 2 %

Load Regulation VOUT to 2 × VOUT 2 %

DMAX Maximum Duty Cycle MIC3230 & MIC3232

MIC3231 90

88 %

VEN Enable Threshold Turn ON

Turn OFF 1.5 1.15

1.1

0.4 V

IEN Enable Pin Current VEN = 3.3V

REN = 100k

17 30 µA

VPWM PWMD Threshold Turn ON

Turn OFF 1.5 0.75

0.7

0.4 V

fPWMD PWMD Frequency Range Note 5 (L = 47µH; C = 4.7µF) 0 500 Hz

fSW Programmable Oscillator

Frequency

RFREQ = 82.5k

RFREQ = 21k

RFREQ = 8.25k

360

109

400

950

440

kHz

fSW Fixed Frequency Option (MIC3232YMM) 360 400 440 kHz

FDITHER Low EMI (MIC3231) Frequency dither shift from nominal ±12 %

VSENS Current Limit Threshold

Voltage

RSENSE = 390 0.315 0.45 0.585 V

ISENSE I

SENSE peak current out RSENSE = 390 250 µA

VOVP Over Voltage Protection 1.203 1.24 1.277 V

Driver Impedance SINK

SOURCE

2.4

3.5 



Micrel, Inc. MIC3230/1/2

January 2009 5 M9999-011409-A

VDRH Driver Voltage High VIN = 12V 7 9 11 V

TJ Over-Temperature

Threshold Shutdown

150 °C

Thermal Shutdown Hysteresis 5 °C

Notes:

1. Exceeding the absolute maximum rating may damage the device.

2. The device is not guaranteed to function outside its operating rating.

3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.

4. Specification for packaged product only.

5. Guaranteed by design

Micrel, Inc. MIC3230/1/2

January 2009 6 M9999-011409-A

Typical Characteristics

Micrel, Inc. MIC3230/1/2

January 2009 7 M9999-011409-A

11.8

11.85

11.9

11.95

12.05

12.1

12.15

12.2

0 25 50 75 100 125 150

OUTPUT VOLTAGE (V)

LOAD (mA)

Load Regulation

VIN =3.6V

Micrel, Inc. MIC3230/1/2

January 2009 8 M9999-011409-A

Functional Description

A constant output current converter is the preferred

method for driving LEDs. Small variations in current have

a minimal effect on the light output, whereas small

variations in voltage have a significant impact on light

output. The MIC323x family of LED drivers are specifically

designed to operate as constant current LED Drivers and

the typical application schematic is shown in Figure 1.

The MIC323x family are designed to operate as a boost

controller, where the output voltage is greater than the

input voltage. This configuration allows for the design of

multiple LEDs in series to help maintain color and

brightness. The MIC323x family can also be configured as

a SEPIC controller, where the output voltage can be either

above or below the input voltage.

The MIC3230/1/2 have a very wide input voltage range,

between 6V and 45V, to help accommodate for a diverse

range of input voltage applications. In addition, the LED

current can be programmed to a wide range of values

through the use of an external resistor. This provides

design flexibility in adjusting the current for a particular

application need.

The MIC3230/1/2 features a low impedance gate driver

capable of switching large MOSFETs. This low

impedance helps provide higher operating efficiency.

The MIC323x family can control the brightness of the

LEDs via its PWM dimming capability. Applying a PWM

signal (up to 500Hz) to the PWMD pin allows for control of

the brightness of the LED.

Each member of the MIC323x family employs peak current

mode control. Peak current mode control offers

advantages over voltage mode control in the following

manner. Current mode control can achieve a superior line

transient performance compared to voltage mode control

and through small signal analysis (not shown here),

current mode control is easier to compensate than voltage

mode control, thus allowing for a less complex control loop

stability design. Figure 2 shows the functional block

diagram.

Control

RSLC

RCS

VDD OVP PWMD

GND

Q1 C5

Clock Out

VRAMP Out

SYNCSYNC

IADJ

RADJ

LDOVIN Internal VBIASVDD

10k

0.25V

Ai = 1.4V

VRAMP IN

ISLOPE COMP OUT

ISLOPE COMP

Leading Edge

Blanking

0.45V

250µa/T

VOUT

COMP

VIN

LED

DRV

VOUT

Figure 2. MIC3230 Functional Block Diagram

Micrel, Inc. MIC3230/1/2

January 2009 9 M9999-011409-A

Power Topology

Constant Output Current Con troller

The MIC323x family are peak current mode boost

controllers designed to drive high power LEDs. Unlike a

standard constant output voltage controller, the MIC323x

family has been designed to provide a constant output

current. The MIC323x family is designed for a wide input

voltage range, from 6V to 45V. In the boost configuration,

the output can be set from VIN up to 100V.

As a peak current mode controller, the MIC323x family

provides the benefits of superior line transient response as

well as an easier to design compensation.

This family of LED drivers features a built-in soft-start

circuitry in order to prevent start-up surges. Other

protection features include:

• Current Limit (ILIMIT) - Current sensing for over current

and overload protection

• Over Voltage Protection (OVP) - Output over voltage

protection to prevent operation above a safe upper

limit

• Under Voltage Lockout (UVLO) – UVLO designed to

prevent operation at very low input voltages

Setting the LED Current

The current through the LED string is set via the value

chosen for the current sense resistor, RADJ. This value can

be calculated using Equation 1:

Eq. (1)

ADJ

LED R

I25.0

Another important parameter to be aware of in the boost

controller design, is the ripple current. The amount of

ripple current through the LED string is equal to the output

ripple voltage divided by the LED AC resistance (RLED –

provided by the LED manufacturer) plus the current sense

resistor (RADJ). The amount of allowable ripple through the

LED string is dependent upon the application and is left to

the designer’s discretion. This equation is shown in

Equation 2:

Eq. (2) )( ADJLED

OUT

LED RR

IRIPPLE

≈Δ

Where

OUT

LED

OUT CTDI

VRIPPLE

××

Reference Voltage

The voltage feedback loop of the MIC323x uses an

internal reference voltage of 0.25V with an accuracy of

±3%. The feedback voltage is the voltage drop across the

current setting resistor (RADJ) as shown in Figure 1. When

in regulation the voltage at IADJ will equal 0.25V.

Output Over Voltage Protection (OVP)

The MIC323x provides an OVP circuitry in order to help

protect the system from an overvoltage fault condition.

This OVP point can be programmed through the use of

external resistors (R8 and R9 in Figure 1). A reference

value of 1.245V is used for the OVP. Equation 3 can be

used to calculate the resistor value for R9 to set the OVP

point.

Eq. (3) 1)245.1/(

9−

OVP

LED Dimming

The MIC323x family of LED drivers can control the

brightness of the LED string via the use of pulse width

modulated (PWM) dimming. A PWM input signal of up to

500Hz can be applied to the PWM DIM pin (see Figure 1)

to pulse the LED string ON and OFF. It is recommended

to use PWM dimming signals above 120Hz to avoid any

recognizable flicker by the human eye. PWM dimming is

the preferred way to dim a LED in order to prevent

color/wavelength shifting, as occurs with analog dimming.

The output current level remains constant during each

PWMD pulse.

Oscillator and Switching Frequency Selection

The MIC323x family features an internal oscillator that

synchronizes all of the switching circuits internal to the IC.

This frequency is adjustable on the MIC3230 and MIC3231

and fixed at 400kHz in the MIC3232.

In the MIC3230/1, the switching frequency can be set by

choosing the appropriate value for the resistor, R1

according to Equation 4:

Eq. (4)

035.1

)(

7526

)( ⎟

⎟

⎠

⎞

⎜

⎝

⎛

=Ω kHzF

SYNC (MIC3230 Only)

Multiple MIC3230 ICs can be synchronized by connecting

their SYNC pins together. When synchronized, the

MIC3230 with the highest frequency (master) will override

the other MIC3230s (slaves). The internal oscillator of the

master IC will override the oscillator of the slave part(s)

and all MIC3230 will be synchronized to the same master

switching frequency.

The SYNC pin is designed to be used only by other

MIC3230s and is available on the MIC3230 only. If the

SYNC pin is being unused, it is to be left floating (open).

In the MIC3231, the SYNC pin is to be left floating (open).

Micrel, Inc. MIC3230/1/2

January 2009 10 M9999-011409-A

Dithering (MIC3231 Only)

The MIC3231 has a feature which dithers the switching

frequency by ±12%. The purpose of this dithering is to

help achieve a spread spectrum of the conducted EMI

noise. This can allow for an overall reduction in noise

emission by approximately 10dB.

Internal Gate Driver

External FETs are driven by the MIC323x’s internal low

impedance gate drivers. These drivers are biased from the

VDD and have a source resistance of 2 and a sink

resistance of 3.5.

VDD

VDD is an internal linear regulator powered by VIN and VDD

is the bias supply for the internal circuitry of the MIC323x.

A 10µF ceramic bypass capacitor is required at the VDD pin

for proper operation. This pin is for filtering only and

should not be utilized for operation.

Current Limit

The MIC323x family features a current limit protection

feature to prevent any current runaway conditions. The

current limit circuitry monitors current on a pulse by pulse

basis. It limits the current through the inductor by sensing

the voltage across RCS. When 0.45V is present at the IS

pin, the pulse is truncated. The next pulse continues as

normally until the IS pin reaches 0.45V and it is truncated

once again. This will continue until the output load is

decreased.

Select RCS using Equation 5:

Eq. (5)

()

LIMITPK

MINMAX L

INOUT

CS I

DVV

45.0

×−

Slope Compensation

The MIC323x is a peak current mode controller and

requires slope compensation. Slope compensation is

required to maintain internal stability across all duty cycles

and prevent any unstable oscillations. The MIC323x uses

slope compensation that is set by an external resistor,

RSLC. The ability to set the proper slope compensation

through the use of a single external component results in

design flexibility. This slope compensation resistor, RSLC,

can be calculated using Equation 6:

Eq. (6)

(

)

CSINOUT

SLC FAL

RVV

RMINMAX

××

×−

250

where VIN_MAX and VOUT_MAX can be selected to system

specifications.

Current Sense IS

The IS pin monitors the rising slope of the inductor current

(m1 in Figure 5) and also sources a ramp current

(250µA/T) that flows through RSLC that is used for slope

compensation. This ramp of 250µA per period, T,

generates a ramped voltage across RSLC and is labeled VA

in Figure 3. The signal at the IS pin is the sum of VCS + VA

(as shown in Figure 3). The current sense circuitry and

block diagram is displayed in Figure 4. The IS pin is also

used as the current limit (see the previous section on

Current Limit).

Figure 3. Slope compensation waveforms

Soft Start

The boost switching convertor features a soft start in order

to power up in a controlled manner, thereby limiting the

inrush current from the line supply. Without this soft start,

the inrush current could be too high for the supply. To

prevent this, a soft start delay can be set using the

compensation capacitor (CCOMP in Figure 1). For switching

to begin, the voltage on the compensation cap must reach

about 0.7V. Switching starts with the minimum duty cycle

and increases to the final duty cycle. As the duty cycle

increases, VOUT will increase from VIN to it’s final value. A

6µA current source charges the compensation capacitor

and the soft start time can be calculated in Equation 7:

Eq. (7) μA

TY_STATECOMP_STEADCOMP

SOFTSTART 6

≈

VCOMP_STEADY_STATE is usually between 0.7V to 3V, but can

be as high as 5V.

Eq. (8)

(

)

PKASTATESTEADYCOMP VcsVAiV PK

×=

Where: TDR

VSLC

RAMP

APK ×××= and

LCS RIV PK ×

Ai = 1.4 V/V

D = Duty cycle (0 to1)

T = period

A 10nF ceramic capacitor will make this system stable at

all operating conditions.

Micrel, Inc. MIC3230/1/2

January 2009 11 M9999-011409-A

Eq. (10)

()

PPIN

RMSINAVEIN

II −=

Leading Edge Blanking

Large transient spikes due to the reverse recovery of the

diode may be present at the leading edge of the current

sense signal. (Note: drive current can also cause such

spikes) For this reason a switch is employed to blank the

first 100ns of the current sense signal. See Figure 6.

Eq. (11) 2

PPIN

AVEINPEAKIN

II +=

Note: If IIN_PP is small then IIN_AVE nearly equals IIN_RMS

Eq. (9)

OUTOUT

RMSIN Veff

I×

+RSLC–

0.45V

CCOMP

COMP

RCOMP = 10k

PWM Comparator

IADJ

VA = IRAMP × RSLP

Current Limit

Clock

250µa/T

VIN

DRV

VCS

VCS = IL × RCS

RCS

–

Figure 4. Current sense circui t (An explanation of the IS pin)

Cloc

PWM

IFET

IDIODE

IL_AVE = IIN_AVE

IL_PK = IL_AVE + 1/2 IL_PP

IOUT

(1-D)T

IL_PP

IFET_RMS

IL_AVE = IIN_AVE

Figure 5. Current Waveforms

Micrel, Inc. MIC3230/1/2

January 2009 12 M9999-011409-A

Figure 6. IS pin and VRCS (Ch1 = Switch Node, Ch2 = IS pin,

Ref1 = VCS)

Design Procedure for a LED Driver

Symbol Parameter Min Nom Max Units

Input

VIN Input Voltage 8 12 14 V

IIN Input current 2 A

Output

LEDs Number of LEDs 5 6 7

VF Forward voltage of LED 3.2 3.5 4.0 V

VOUT Output voltage 16 21 28 V

ILED LED current 0.33 0.35 0.37 A

IPP Required I Ripple 40 mA

PWMD PWM Dimming 0 100 %

OVP

Output over voltage

protection 30 V

System

FSW Switching frequency 500kHz

eff Efficiency 80 %

VDIODE

Forward drop of schottky

diode 0.6 V

Table 2. Design example parameters

Micrel, Inc. MIC3230/1/2

January 2009 13 M9999-011409-A

47µH D1

100k

4.33k

COUT

4.7µF

100V

100k

RADJ

1/4W

RFS

16.5k CCOMP

10nF

CIN

4.7µF/50v

RSLC

VFB = 0.25V RCS

1/2W

Analog ground Power ground

VOUTVIN

PWMD

ENABLE

Synch to other MIC3230

ILED Return

LED 1

LED X

COMP

PWMD

VDD

AGND PGND

EPAD

OVP

DRV

VIN

IADJ

MIC3230/31

SYNC

10µF

10V

Figure 7. Design Example Schematic

Design Example

In this example, we will be designing a boost LED driver

operating off a 12V input. This design has been created

to drive six LEDs at 350mA with a ripple of about 12%.

We are designing for 80% efficiency at a switching

frequency of 500kHz.

Select RFS

To operate at a switching frequency of 500kHz, the RFS

resistor must be chosen using Equation 3.

()

Ω==Ω kkRFS 6.16

500

7526 035.1

Use the closest standard value resistor of 16.5k.

Select RADJ

Having chosen the LED drive current to be 350mA in this

example, the current can be set by choosing the RADJ

resistor from Equation 1:

Ω== 71.0

35.0

25.0

RADJ

The power dissipation in this resistor is:

()

mWRIRP ADJADJ 87*

2==

Use a resistor rated at ¼ watt or higher. Choose the

closest value from a resistor manufacture.

Operating Duty Cycle

The operating duty cycle can be calculated using

Equation 12 provided below:

Eq. (12)

diode

VVout

VVineffVout

D+

+×−

=)(

These can be calculated for the nominal (typical)

operating conditions, but should also be understood for

the minimum and maximum system conditions as listed

below.

schottkynom

schottkynomnom

nom VVout

VVineffVout

D+

+×−

=)(

schottky

VVout

VVineffVout

D+

+×−

max

minmax

max

)(

schottky

VVout

VVineffVout

D+

+×−

min

maxmin

min

)(

Therefore DNOM =56% DMAX = 78% and DMIN = 33%

Inductor Selection

First, it is necessary to calculate the RMS input current

(nominal, min and max) for the system given the

operating conditions listed in the design example table.

This minimum value of the RMS input current is

necessary to ensure proper operation. Using Equation

9, the following values have been calculated:

rmsA

Veff

OUTOUT

RMSIN _64.1

min_

max_max_

max__ =

Micrel, Inc. MIC3230/1/2

January 2009 14 M9999-011409-A

rmsA

Veff

nomIN

nomOUTnomOUT

nomRMSIN _78.0

__ =

rmsA

Veff

OUTOUT

RMSIN _48.0

max_

min_min_

min__ =

Iout is the same as ILED

Selecting the inductor current (peak-to-peak), IL_PP, to be

between 20% to 50% of IIN_RMS_nom, in this case 40%, we

obtain:

PPnomrmsinnomPPin AII −

=== 31.078.0*4.04.0 ____

(see the current waveforms in Figure 5).

It can be difficult to find large inductor values with high

saturation currents in a surface mount package. Due to

this, the percentage of the ripple current may be limited

by the available inductor. It is recommended to operate

in the continuous conduction mode. The selection of L

described here is for continuous conduction mode.

Eq. (13)

PPin

TDV

××

Using the nominal values, we get:

31.0

256.012 =

××

Select the next higher standard inductor value of 47µH.

Going back and calculating the actual ripple current

gives:

Eq. (13a) PP

nomnomIN

PPin A

usv

TDV

I29.0

256.012

××

The average input current is different than the RMS input

current because of the ripple current. If the ripple current

is low, then the average input current nearly equals the

RMS input current. In the case where the average input

current is different than the RMS, Equation 10 shows the

following:

Eq. (13b)

()

(

)

max__max__

PPIN

RMSINAVEIN

II −=

()()

AI AVEIN 64.112/29.064.1 22

max__ ≈−=

The Maximum Peak input current IL_PK can found using

equation 11:

AIII PPLAVEINPKL 78.15.0 max__max__max__ =

The saturation current (ISAT) at the highest operating

temperature of the inductor must be rated higher than

this.

The power dissipated in the inductor is:

Eq. (13c)

DCRIP RMSinINDUCTOR ×= 2

max__

Current Limit and Slope Compensati on

Having calculated the IL_pk above, We can set the current

limit 20% above this maximum value:

AAI Limit

pkL 9.16.12.1

_=×=

The internal current limit comparator reference is set at

0.45V, therefore when , the IC enters

current limit.

45.0

PINIS

Eq. (14)

(

)

PKA VcsV PK +

45.0

Where is the peak of the waveform and

is the peak of the Vcs waveform

Vcs

Eq. (14a) CSpkLSLCRAMP RIDRI Limit

××

45.0

To calculate the value of the slope compensation

resistance, RSLC, we can use Equation 5:

(

)

CSINOUT

SLC FAL

RVV

RMINMAX

××

×−

250

First we must calculate RCS, which is given below in

Equation 15:

Eq. (15)

()

Limit

pkL

MINMAX

DVINVOUT

max

45.0

×−

Therefore;

()()

Ω=

×−

kHzH

RCS 179

9.1

50047

50.0828

45.0

Using a standard value 150m resistor for RCS, we

obtain the following for RSLC:

(

)

Ω=

××

Ω×−

=511

50025047

150828

kHzAH

RSLC

μμ

Use the next higher standard value if this not a standard

value. In this example 511 is a standard value.

Check: Because we must use a standard value for Rcs

and RSLC; may be set at a different level (if the

calculated value isn’t a standard value) and we must

calculate the actual value (remember

is the same as ).

Limit

pkL

Limit

pkL

Limit

pkin _

Limit

pkL

I_I

Rearranging Equation 14a to solve for :

Limit

pkL

SLCRAMP

pkin R

DRI

ILimit

)45.0(

××−

ILimit

actualin 34.2

150.

)75.051125045.0(

××−

This is higher than the initial

limit because we have to use standard values for RCS

AI PKL 9.12.1 max__ =×

Micrel, Inc. MIC3230/1/2

January 2009 15 M9999-011409-A

and for RSLC. If is too high than use a higher

value for RCS. The calculated value of RCS for a 1.9A

current limit was 179m. In this example, we have

chosen a lower value which results in a higher current

limit. If we use a higher standard value the current limit

will have a lower value. The designer does not have the

same choices for small valued resistors as with larger

valued resistors. The choices differ from resistor

manufacturers. If too large a current sense resistor is

selected, the maximum output power may not be able to

be achieved at low input line voltage levels. Make sure

the inductor will not saturate at the actual current limit

Limit

actualin

()

A51178.0 Ω××

Limit

actualin

VPINIS 250

Perform a check at IIN=2.34Apk.

VmA 45.015034.2 =Ω×+

Maximum Power dissipated in RCS is;

Eq. (17)

RMS ×

RR IP CSCS =_

Eq. (18) ⎟

⎟

⎠

⎞

⎜

⎝

⎛+== _

max_ RMSFET

RMS 12

max__max_

PPL

AVEIN

rmsA

RMS _44.1

26.0

64.178.0

⎟

⎠

⎞

⎜

⎝

⎛+=ICS

wattP CS

R31.015.25.1 2=×=

Use a 1/2 Watt resistor for RCS.

Output Capacitor

In this LED driver application, the ILED ripple current is a

more important factor compared to that of the output

ripple voltage (although the two are directly related). To

find the COUT for a required ILED ripple use the following

calculation:

For an output ripple 20% of

ripple

ILED nom

ILED

mAILEDripple 7035.02.0 =×=

Eq. (19) )(*

_totalLEDadjripple

nomnom

out RRILED

TDILED

C+

Find the equivalent ac resistance from the

datasheet of the LED. This is the inverse slope of the

ILED vs. VF curve i.e.:

acLED

Eq. (20) ILED

acLED Δ

In this example use for each LED.

Ω= 1.0

_acLED

If the LEDs are connected in series, multiply

s, we obtain the following:

Ω= 1.0

_acLED

R by the total number of LEDs. In this

example of 6 LED

=Ω×= 6.01.06

_totalLED

RRILED

TDILED

totalLEDadjripple

nomnom

out 1.4

)(*

Use the next highest standard value, which is 4.7uF.

is shown in Figure 5. For superior

There is a trade off between the output ripple and th

rising edge of the PWMD pulse. This is because

between PWM dimming pulses, the converter stops

pulsing and COUT will start to discharge. The amount that

COUT will discharge depends on the time between PWM

Dimming pluses. At the next PWMD pulse COUT has to

be charged up to the full output voltage VOUT before the

desired LED current flows.

Input Capacitor

The input current

performance, ceramic capacitors should be used

because of their low equivalent series resistance (ESR).

The input ripple current is equal to the ripple in the

inductor plus the ripple voltage across the input

capacitor, which is the ESR of CIN times the inductor

ripple. The input capacitor will also bypass the EMI

generated by the converter as well as any voltage spikes

generated by the inductance of the input line. For a

required VIN_RIPPLE:

Eq. (21)

()

kHzmV

SWRIPPLEIN

PPIN

4.1

500508

28.0

××

This is the minimum value that should be used. The

e, the FET has to hold off an output

input capacitor should also be rated for the maximum

RMS input current. To protect the IC from inductive

spikes or any overshoot, a larger value of input

capacitance may be required and it is recommended that

ceramic capacitors be used. In this design example a

value of 4.7µF ceramic capacitor was selected.

MOSFET Selection

In this design exampl

voltage maximum of 30V. It is recommended to use an

80% de-rating value on switching FETs, so a minimum

of a 38V FET should be selected. In this design

example, a 75V FET has been selected.

The switching FET power losses are the sum of the

conduction loss and the switching loss:

Eq. (22) FETCONDFETFET PPP __ +SWITCH

The conduct ion loss of the FET is when the FET is

, where

turned on. The conduction power loss of the FET is

found by the following equation:

Eq. (23) FETCONDFET IP =__ DSONRMS R×

Micrel, Inc. MIC3230/1/2

January 2009 16 M9999-011409-A

⎟

⎠

⎞

⎜

⎝

⎛+= 12

__ PPL

AVEINRMSFET

IDI

The switching losses occur during the switching

transitions of the FET. The transition times, ttransition, are

ere are the times when the FET is turning off and on. Th

two transition times per period, T. It is important not to

confuse T (the period) with the transition time, ttransition.

Eq. (24) Fsw

Eq. (25)

SWtransitionOUTAVEFETSWITCHFET FtVIP

××= max_max_max__max__

max_transition

t: To find

Eq. (26) Igatedrv

ttransition ≈

max_

is the total gate c rge of the external

MOSFET provided by the MOSFET manufacturer and

charge at a

where Qg ha

the Qg ould chosen at a VGS10V. This is not an

exact value, but is more of an estimate of max_transition

The FET manufacturers’ provide a gate

specified VGS voltage:

C_=

FET V@

This is the FET’s input capacitance. Select a FET with

RDS(on) and QG such that the external power is below

harge=68nC (typical)

The perature. As the

erature

at 125°C is:

ation 23:

about 0.7W for a SO-8 or about 1W for a PowerPak

(FET package). The Vishay Siliconix Si7148DP in a

PowerPak SO-8 package is one good choice. The

internal gate driver in the MIC3230/1/2 is 2A. From the

Si7148DP data sheet:

RDS(on)_25°C=0.0145

Total gate C

)is a function of tem

(

(tempR onDS )

in the FET increases so does the temp RDS(on).

To find )(

)( tempR onDS use Equation 27, or simply

o o

double the ) for )125(

)( CR onDS .

Eq. (27) (onDS

25(

)( CR onDS

)007.1()25()( )25(

))(

o−

×= Temp

onDS CRtemp

The (

RDS )(

)temp

Ω≈

−

∗mCRDSon 30)7)125( )25125( o

×= 00.1(0145.0

From Equ mWmP CONDFET 623064.1 2

_=Ω×=

From Equation 26: ns

Igatedrv

ttransition 34

68 ==≈

AVE 64.1

max__ =

IFET

VVOUT 28

max_ =

From Equation 25:

WattskHznsVA 78.0500342864.1PSWITCHFET max__

××

From Equation 22

WWmW 84.078.062 =+PFET

This about the limit for a part on a circuit board without

having to use any additional heat sinks.

is best used here because of the lower

d the low reverse recovery time. The

Rectifier Diode

A Schottky Diode

forward voltage an

voltage stress on the diode is the max VOUT and

therefore a diode with a higher rating than max VOUT

should be used. An 80% de-rating is recommended

here as well.

Eq (28)

⎟

⎠

⎞⎛ 2

_diode

⎜

⎝

+−= 12

)1( _

max__max_

PPL

AVEINRMS ID

Eq. (29) WP

IVP

diode

RMSdiodeSCHOTTKYdiode

81.0

max__

≈

MIC3230 power losses

IC3230are:

is the total gate charge of the

The power losses in the M

Eq.(30) FVQP gategateMIC ××=

3230 VinIQ×+

where gat

Qexternal

ati

MOSFE gate

V is the gate drive voltage of the

MIC3230. Fis the switching frequency. Q

I is the

quiescent cu rent of the MIC3230 found in th lectrical

characteriz on table. mAIQ2.3=. VIN is the voltage at

the VIN pin of the MIC3230. From Eq.(30)

mAkHznFPMIC 142.35001268

3230

e e

W45.0

Micrel, Inc. MIC3230/1/2

January 2009 17 M9999-011409-A

OVP-Over voltage protection 2. Even though the RRC is very short (tens of

nanoseconds) the peak currents are high (multiple

amperes). The high RRC causes a voltage drop on the

ground trace of the PCB and if the converter control IC is

referenced to this voltage drop, the output regulation will

suffer.

Set OVP higher than the maximum output voltage by at

least one volt. To find the resistor divider values for

OVP use Equation 3 and set the OVP=30V and

R8=100k:

Ω=

−

×Ω

R33.4

245.130

245.1100

3. It is important to connect the IC’s reference to the

same point as the output capacitors to avoid the voltage

drop caused by RRC. This is also called a star

connection or single point grounding.

PCB Layout

1. All typologies of DC-to-DC converters have a reverse

recovery current (RRC) of the flyback or (freewheeling)

diode. Even a Schottky diode, which is advertised as

having zero RRC, it really is not zero. The RRC of the

freewheeling diode in a boost converter is even greater

than in the Buck converter. This is because the output

voltage is higher than the input voltage and the diode

has to charge up to –VOUT during each on-time pulse and

then discharge to VF during the off-time.

4. Feedback trace: The high impedance traces of the

FB should be short.

Micrel, Inc. MIC3230/1/2

January 2009 18 M9999-011409-A

Package Information

10-Pin MSOP (MM)

Micrel, Inc. MIC3230/1/2

January 2009 19 M9999-011409-A

12-Pin 3mm × 3mm MLF® (ML)

Micrel, Inc. MIC3230/1/2

January 2009 20 M9999-011409-A

16-Pin Exposed Pad TSSOP (TSE)

MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA

TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com

The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its

use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.

Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product

reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical impla

into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A

Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully

indemnify Micrel for any damages resulting from such use or sale.

can nt