Product Folder Order Now Support & Community Tools & Software Technical Documents LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 LM5118 Wide Voltage Range Buck-Boost Controller 1 Features 3 Description * * * The LM5118 wide voltage range Buck-Boost switching regulator controller features all of the functions necessary to implement a highperformance, cost-efficient Buck-Boost regulator using a minimum of external components. The BuckBoost topology maintains output voltage regulation when the input voltage is either less than or greater than the output voltage, making it especially suitable for automotive applications. The LM5118 operates as a buck regulator while the input voltage is sufficiently greater than the regulated output voltage and gradually transitions to the buck-boost mode as the input voltage approaches the output. This dual-mode approach maintains regulation over a wide range of input voltages with optimal conversion efficiency in the buck mode and a glitch-free output during mode transitions. This easy-to-use controller includes drivers for the high-side buck MOSFET and the lowside boost MOSFET. The regulators control method is based upon current mode control using an emulated current ramp. Emulated current mode control reduces noise sensitivity of the pulse-width modulation circuit, allowing reliable control of the very small duty cycles necessary in high input voltage applications. Additional protection features include current limit, thermal shutdown and an enable input. The device is available in a power-enhanced, 20-pin HTSSOP package featuring an exposed die attach pad to aid thermal dissipation. 1 * * * * * * * * * * * Ultra-Wide Input Voltage Range From 3 V to 75 V Emulated Peak Current Mode Control Smooth Transition Between Step-Down and StepUp Modes Switching Frequency Programmable to 500 KHz Oscillator Synchronization Capability Internal High Voltage Bias Regulator Integrated High and Low-Side Gate Drivers Programmable Soft-Start Time Ultra-Low Shutdown Current Enable Input Wide Bandwidth Error Amplifier 1.5% Feedback Reference Accuracy Thermal Shutdown Package: 20-Pin HTSSOP (Exposed Pad) Create a Custom Design Using the LM5118 With the WEBENCH(R) Power Designer 2 Applications Industrial Buck-Boost Supplies Device Information(1) PART NUMBER PACKAGE LM5118 BODY SIZE (NOM) HTSSOP (20) 6.50 mm x 4.40 mm (1) For all available packages, see the orderable addendum at the end of the data sheet. Simplified Schematic Efficiency vs VIN and IOUT, VOUT = 12 V 100 VIN xx x xx x xxxxxxx xx xx VCC 95 HB LM5118 VOUT HS SS CS RAMP 90 x 85 x x 1 AMP x x x x 2 AMP x 80 VOUT FB GND x CSG LO RT EFFICIENCY (%) 3 AMP HO COMP 75 0 10 20 30 40 50 60 70 VIN (V) 1 An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications, intellectual property matters and other important disclaimers. PRODUCTION DATA. LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Table of Contents 1 2 3 4 5 6 Features .................................................................. Applications ........................................................... Description ............................................................. Revision History..................................................... Pin Configuration and Functions ......................... Specifications......................................................... 6.1 6.2 6.3 6.4 6.5 6.6 7 1 1 1 2 4 6 Absolute Maximum Ratings ...................................... 6 ESD Ratings.............................................................. 6 Recommended Operating Conditions....................... 6 Thermal Information .................................................. 6 Electrical Characteristics........................................... 7 Typical Characteristics ............................................ 10 Detailed Description ............................................ 12 7.1 Overview ................................................................. 12 7.2 Functional Block Diagram ....................................... 12 7.3 Feature Description................................................. 13 7.4 Device Functional Modes........................................ 20 8 Application and Implementation ........................ 23 8.1 Application Information............................................ 23 8.2 Typical Application ................................................. 23 9 Power Supply Recommendations...................... 35 9.1 Thermal Considerations .......................................... 35 9.2 Bias Power Dissipation Reduction .......................... 36 10 Layout................................................................... 38 10.1 Layout Guidelines ................................................. 38 10.2 Layout Example .................................................... 38 11 Device and Documentation Support ................. 39 11.1 11.2 11.3 11.4 Device Support...................................................... Trademarks ........................................................... Electrostatic Discharge Caution ............................ Glossary ................................................................ 39 39 39 39 12 Mechanical, Packaging, and Orderable Information ........................................................... 39 4 Revision History NOTE: Page numbers for previous revisions may differ from page numbers in the current version. Changes from Revision I (August 2014) to Revision J Page * Moved the automotive device to a standalone data sheet (SNVSAX9) ................................................................................ 1 * Moved the storage temp parameters to the Absolute Maximum Ratings .............................................................................. 6 * Changed junction temperature range from: 150C max to: -40C to 150C ....................................................................... 6 * Changed Handling Ratings table to ESD Ratings .................................................................................................................. 6 * Moved the junction temperature ranges to the condition statement ...................................................................................... 7 * Removed the VOLL minimum value ......................................................................................................................................... 8 * Removed the VOLH minimum value......................................................................................................................................... 8 Changes from Revision H (October 2013) to Revision I Page * Changed data sheet flow and layout to conform with new TI standards. Added the following sections: Application and Implementation; Power Supply Recommendations; Layout; Device and Documentation Support; Mechanical, Packaging, and Ordering Information .................................................................................................................................... 1 * Deleted footnote "These pins must not be raised above VIN." .............................................................................................. 6 Changes from Revision G (February, 2013) to Revision H Page * Deleted VIN Range on Functional Block Diagram................................................................................................................. 12 * Changed Figure 19............................................................................................................................................................... 24 * Changed Inductor Selection, L1 Text ................................................................................................................................... 27 * Changed Equation 15........................................................................................................................................................... 27 * Changed Equation 16 .......................................................................................................................................................... 27 * Added Efficiency Parameter ................................................................................................................................................. 27 * Changed II(PEAK) Value .......................................................................................................................................................... 28 * Changed I2(PEAK) Value.......................................................................................................................................................... 28 * Changed R13 = RSENSEText ................................................................................................................................................ 28 * Added Equation 19 .............................................................................................................................................................. 28 2 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 * Added Equation 20 .............................................................................................................................................................. 28 * Changed Equation 21 .......................................................................................................................................................... 28 * Changed Equation 22 .......................................................................................................................................................... 28 * Changed R13(BUCK) Value ..................................................................................................................................................... 28 * Changed R13(BUCK-BOOST) Value ............................................................................................................................................ 28 * Added Inductor Current Limit Calculation Section................................................................................................................ 29 * Added Equation 24 .............................................................................................................................................................. 29 * Added Equation 26 .............................................................................................................................................................. 29 * Changed Equation 29 .......................................................................................................................................................... 29 * Changed ESRMAX Value ....................................................................................................................................................... 29 * Deleted Previous Equation 17 .............................................................................................................................................. 30 * Deleted Previous Equation 18 .............................................................................................................................................. 30 * Changed C1 - C5 = Input Capacitor Text............................................................................................................................. 30 * Changed R8, R9 Text........................................................................................................................................................... 31 * Changed Equation 38 .......................................................................................................................................................... 31 * Changed R1, R3, C21 Text .................................................................................................................................................. 31 * Changed Equation 40 .......................................................................................................................................................... 32 * Changed DMIN to DMAX .......................................................................................................................................................... 32 * Changed DMAX Value ............................................................................................................................................................ 32 * Changed DC Gain(MOD) Value............................................................................................................................................... 32 * Changed ESRZERO Value...................................................................................................................................................... 32 * Changed R4, C18, C17 Text ................................................................................................................................................ 33 * Changed Figure 21............................................................................................................................................................... 34 * Changed Figure 22............................................................................................................................................................... 34 * Changed Figure 23............................................................................................................................................................... 34 * Added Figure 24 ................................................................................................................................................................... 34 * Added Figure 25 ................................................................................................................................................................... 34 * Added Figure 26 ................................................................................................................................................................... 34 Changes from Revision F (February 2013) to Revision G * Page Changed layout of National Data Sheet to TI format ........................................................................................................... 24 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 3 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com 5 Pin Configuration and Functions PWP Package 20-Pin HTSSOP (Top View) VIN 1 20 HS UVLO 2 19 HO RT 3 18 HB EN 4 17 VCCX RAMP 5 16 VCC AGND 6 15 LO SS 7 14 PGND FB 8 13 CSG COMP 9 12 CS VOUT 10 11 SYNC Pin Descriptions PIN NO. NAME TYPE (1) DESCRIPTION 1 VIN P/I 2 UVLO I Input supply voltage. If the UVLO pin is below 1.23 V, the regulator will be in standby mode (VCC regulator running, switching regulator disabled). When the UVLO pin exceeds 1.23 V, the regulator enters the normal operating mode. An external voltage divider can be used to set an undervoltage shutdown threshold. A fixed 5-A current is sourced out of the UVLO pin. If a current limit condition exists for 256 consecutive switching cycles, an internal switch pulls the UVLO pin to ground and then releases. 3 RT I The internal oscillator frequency is set with a single resistor between this pin and the AGND pin. The recommended frequency range is 50 kHz to 500 kHz. 4 EN I If the EN pin is below 0.5 V, the regulator will be in a low power state drawing less than 10 A from VIN. EN must be raised above 3 V for normal operation. 5 RAMP I Ramp control signal. An external capacitor connected between this pin and the AGND pin sets the ramp slope used for emulated current mode control. 6 AGND G Analog ground. 7 SS I Soft Start. An external capacitor and an internal 10-A current source set the rise time of the error amp reference. The SS pin is held low when VCC is less than the VCC undervoltage threshold (< 3.7 V), when the UVLO pin is low (< 1.23 V), when EN is low (< 0.5 V) or when thermal shutdown is active. 8 FB I Feedback signal from the regulated output. Connect to the inverting input of the internal error amplifier. 9 COMP O Output of the internal error amplifier. The loop compensation network should be connected between COMP and the FB pin. 10 VOUT I Output voltage monitor for emulated current mode control. Connect this pin directly to the regulated output. 11 SYNC I Sync input for switching regulator synchronization to an external clock. 12 CS I Current sense input. Connect to the diode side of the current sense resistor. 13 CSG I Current sense ground input. Connect to the ground side of the current sense resistor. 14 PGND G Power Ground. 15 LO O Boost MOSFET gate drive output. Connect to the gate of the external boost MOSFET. 16 VCC P/I/O 17 VCCX P/I (1) 4 Output of the bias regulator. Locally decouple to PGND using a low ESR/ESL capacitor located as close to the controller as possible. Optional input for an externally supplied bias supply. If the voltage at the VCCX pin is greater than 3.9 V, the internal VCC regulator is disabled and the VCC pin is internally connected to VCCX pin supply. If VCCX is not used, connect to AGND. G = Ground, I = Input, O = Output, P = Power Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Pin Descriptions (continued) PIN NO. NAME TYPE (1) DESCRIPTION 18 HB I High-side gate driver supply used in bootstrap operation. The bootstrap capacitor supplies current to charge the high-side MOSFET gate. This capacitor should be placed as close to the controller as possible and connected between HB and HS. 19 HO O Buck MOSFET gate drive output. Connect to the gate of the high-side buck MOSFET through a short, low inductance path. 20 HS I Buck MOSFET source pin. Connect to the source terminal of the high-side buck MOSFET and the bootstrap capacitor. -- EP -- Solder to the ground plane under the IC to aid in heat dissipation. Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 5 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com 6 Specifications 6.1 Absolute Maximum Ratings over operating free air temperature range (unless otherwise noted) (1) MIN MAX UNIT VIN, EN, VOUT to GND -0.3 76 V VCC, LO, VCCX, UVLO to GND -0.3 15 V HB to HS -0.3 15 V HO to HS -0.3 HB + 0.3 V HS to GND -4 76 V CSG, CS to GND -0.3 0.3 V RAMP, SS, COMP, FB, SYNC, RT to GND -0.3 7 V Junction temperature -40 150 C Storage temperature, Tstg -55 150 C (1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. 6.2 ESD Ratings VALUE V(ESD) (1) (2) Electrostatic discharge Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1) 2000 Charged-device model (CDM), per JEDEC specification JESD22-C101 (2) 1000 UNIT V JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process. 6.3 Recommended Operating Conditions over operating free air temperature range (unless otherwise noted) (1) MIN MAX (2) VIN UNIT 3 75 VCC, VCCX 4.75 14 V Junction temperature -40 +125 C (1) (2) V The Recommended Operating Conditions indicate conditions for which the device is intended to be functional, but does not ensure specific performance limits. For specifications and test conditions see Electrical Characteristics. 5-V VIN is required to initially start the controller. 6.4 Thermal Information LM5118 THERMAL METRIC (1) PWP (HTSSOP) UNIT 20 PINS RJA Junction-to-ambient thermal resistance 40 C/W RJC(bot) Junction-to-case (bottom) thermal resistance 4 C/W (1) 6 For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report. Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 6.5 Electrical Characteristics Unless otherwise specified, the following conditions apply: VIN = 48 V, VCCX = 0 V, EN = 5 V, RT = 29.11 k, No load on LO and HO. Typical values apply for TJ = 25C; minimum and maximum values apply over the full junction temperature range for operation, -40C to +125C. (1) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT VIN SUPPLY IBIAS VIN operating current VCCX = 0 V 4.5 5.5 mA IBIASX VIN operating current VCCX = 5 V 1 1.85 mA ISTDBY VIN shutdown current EN = 0 V 1 10 A VCC(REG) VCC regulation VCCX = 0 V 6.8 7 7.2 V VCC(REG) VCC regulation VCCX = 0 V, VIN = 6 V 5 5.25 5.5 VCC sourcing current limit VCC = 0 21 35 VCCX switch threshold VCCX rising 3.68 3.85 VCC REGULATOR VCCX switch hysteresis ICCX = 10 mA VCCX switch leakage VCCX = 0 V VCCCX pulldown resistance VCCX = 3 V VCC undervoltage lockout voltage VCC rising V 5 12 0.5 1 A V 70 3.52 VCC undervoltage hysteresis HB DC bias current 4.02 0.2 VCCX switch RDS(ON) 3.7 k 3.86 0.21 HB-HS = 15 V 205 VC LDO mode turnoff V mA V V 260 10 A V EN INPUT VIL max VIH min EN input low threshold EN input high threshold 0.5 V 3 V EN input bias current VEN = 3 V -1 1 A EN input bias current VEN = 0.5 V -1 1 A EN input bias current VEN = 75 V UVLO standby threshold UVLO rising 50 A UVLO THRESHOLDS 1.191 UVLO threshold hysteresis UVLO pullup current source UVLO = 0 V UVLO pulldown RDS(ON) 1.231 1.271 V 0.105 V 5 A 100 200 10.5 13.5 SOFT START SS current source SS = 0V SS to FB offset FB = 1.23 V SS output low voltage Sinking 100 A, UVLO = 0 V FB reference voltage Measured at FB pin, FB = COMP FB input bias current FB = 2 V 7.5 A 150 mV 7 mV ERROR AMPLIFIER VREF COMP sink/source current AOL DC gain fBW Unity bain bandwidth (1) 1.212 1.23 1.248 20 200 3 V nA mA 80 dB 3 MHz Minimum and maximum limits are 100% production tested at 25C. Limits over the operating temperature range are ensured through correlation using Statistical Quality Control (SQC) methods. Limits are used to calculate Average Outgoing Quality Level (AOQL). Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 7 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Electrical Characteristics (continued) Unless otherwise specified, the following conditions apply: VIN = 48 V, VCCX = 0 V, EN = 5 V, RT = 29.11 k, No load on LO and HO. Typical values apply for TJ = 25C; minimum and maximum values apply over the full junction temperature range for operation, -40C to +125C.(1) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT 305 400 495 ns PWM COMPARATORS tHO(OFF) Forced HO off-time TON(MIN) Minimum HO on-time COMP to comparator offset 70 ns 200 mV OSCILLATOR (RT PIN) fSW1 Frequency 1 RT = 29.11 k 178 200 224 kHz fSW2 Frequency 2 RT = 9.525 k 450 515 575 kHz SYNC Sync threshold falling 1.3 V CURRENT LIMIT VCS(TH) Cycle-by-cycle sense voltage threshold (CS-CSG) RAMP = 0 buck mode VCS(THX) Cycle-by-cycle sense voltage threshold (CS-CSG) RAMP = 0 buck-boost mode CS bias current CS = 0 V CSG bias current CSG = 0 V -103 -125 -147 -218 -255 -300 45 60 45 60 Current limit fault timer 256 mV mV A A cycles RAMP GENERATOR IR1 RAMP current 1 VIN = 60 V, VOUT = 10 V 245 305 365 A IR2 RAMP current 2 VIN = 12 V, VOUT = 12 V 95 115 135 A IR3 RAMP current 3 VIN = 5 V, VOUT = 12 V 65 80 95 A VOUT bias current VOUT = 48 V 245 LO low-state output voltage ILO = 100 mA 0.14 LO high-state output voltage ILO = -100 mA VOHL = VCC-VLO 0.25 LO rise time C-load = 1 nF, VCC = 8 V 16 ns LO fall time C-load = 1 nF, VCC = 8 V 14 ns IOHL Peak LO source current VLO = 0 V, VCC = 8 V 2.2 A IOLL Peak LO sink current VLO = VCC = 8 V 2.7 A A LOW-SIDE (LO) GATE DRIVER VOLL VOHL 0.23 V V HIGH-SIDE (HO) GATE DRIVER VOLH HO low-state output voltage IHO = 100 mA HO high-state output voltage IHO = -100 mA, VOHH = VHB-VOH HO rise time C-load = 1 nF, VCC = 8 V 14 ns HO fall time C-load = 1 nF, VCC = 8 V 12 ns IOHH Peak HO source current VHO = 0V, VCC = 8 V 2.2 A IOLH Peak HO sink current VHO = VCC = 8 V 3.5 A 3 V VOHH HB-HS undervoltage lockout 8 Submit Documentation Feedback 0.135 0.25 0.21 V V Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Electrical Characteristics (continued) Unless otherwise specified, the following conditions apply: VIN = 48 V, VCCX = 0 V, EN = 5 V, RT = 29.11 k, No load on LO and HO. Typical values apply for TJ = 25C; minimum and maximum values apply over the full junction temperature range for operation, -40C to +125C.(1) PARAMETER TEST CONDITIONS MIN TYP MAX 69% 75% 80% UNIT BUCK-BOOST CHARACTERISTICS Buck-boost mode Buck duty cycle (2) THERMAL TSD Thermal shutdown temperature Thermal shutdown hysteresis (2) 165 C 25 C When the duty cycle exceeds 75%, the LM5118 controller gradually phases into the Buck-Boost mode. Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 9 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com 6.6 Typical Characteristics -100 100 CURRENT LIMIT THRESHOLD (mV) xx x xx x xxxxxxx xx xx 95 EFFICIENCY (%) 3 AMP x 90 x 85 x x 1 AMP x x x x 2 AMP x 80 75 0 10 20 30 40 50 60 -125 -150 -175 -200 -225 -250 -275 65 70 70 75 VIN (V) 85 95 100 105 110 90 VOUT/VIN DC (%) Figure 1. Efficiency vs VIN and IOUT, VOUT = 12 V Figure 2. Current Limit Threshold vs VOUT/VIN VOUT = 12 V 10 10 8 8 6 6 VCC (V) VCC (V) 80 4 4 2 2 0 0 0 2 4 6 8 10 0 12 10 20 VIN (V) 30 40 50 IVCC (mA) Figure 3. VCC vs VIN Figure 4. VCC vs IVCC 50 150 40 120 30 90 4 3.5 60 10 30 CURRENT (A) 20 HO Sink PHASE () GAIN (dB) 3 2.5 LO Sink 2 1.5 LO Source 1 0 0 HO Source 0.5 -10 1E+04 1E+05 1E+06 -30 1E+07 0 0 FREQUENCY (Hz) 2 3 4 5 6 7 8 OUTPUT VOLTAGE (V) Figure 5. Error Amplifier Gain/Phase 10 1 Figure 6. LO and HO Peak Gate Current vs Output Voltage VCC = 8 V Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Typical Characteristics (continued) 600 500 FOSC (kHz) 400 300 200 100 0 0 20 40 60 80 100 120 140 160 RT (k:) Figure 7. Oscillator Frequency vs RT Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 11 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com 7 Detailed Description 7.1 Overview The LM5118 high voltage switching regulator features all of the functions necessary to implement an efficient high voltage buck or buck-boost regulator using a minimum of external components. The regulator switches smoothly from buck to buck-boost operation as the input voltage approaches the output voltage, allowing operation with the input greater than or less than the output voltage. This easy to use regulator integrates highside and low-side MOSFET drivers capable of supplying peak currents of 2 A. The regulator control method is based on current mode control using an emulated current ramp. Peak current mode control provides inherent line feed-forward, cycle-by-cycle current limiting and ease of loop compensation. The use of an emulated control ramp reduces noise sensitivity of the pulse-width modulation circuit, allowing reliable processing of very small duty cycles necessary in high input voltage applications. The operating frequency is user programmable from 50 kHz to 500 kHz. An oscillator synchronization pin allows multiple LM5118 regulators to self synchronize or be synchronized to an external clock. Fault protection features include current limiting, thermal shutdown, and remote shutdown capability. An undervoltage lockout input allows regulator shutdown when the input voltage is below a user selected threshold, and a low state at the enable pin will put the regulator into an extremely low current shutdown state. The device is available in the HTSSOP-20EP package featuring an exposed pad to aid in thermal dissipation. 7.2 Functional Block Diagram 17 LM5118 Vin VIN 1 VIN 4 EN VCCX VCC 16 7V REGULATOR VCC UVLO R8 C8 C2 C1 3.9V 3V 5 A THERMAL SHUTDOWN SHUTDOWN AND STANDBY MODE CONTROL R1 2 HB 18 C13 DISABLE HB UVLO UVLO R2 C12 1.23V HICCUP MODE FAULT TIMER Vin 10 A 7 SS PWM C4 Q1 CLK 5V S Q R Q HO 19 HS 20 L1 LEVEL SHIFT D2 VOUT 1.23V C1 1 I-LIMIT 8 FB ERROR AMP C5 C6 Vth( buck) = 1.25V Vth ( buck- boost) = 2.50V + R4 9 D1 CLK 10 Rs V/A A=10 C9 R7 CS 12 TRACK and HOLD C1 0 Rs CSG 13 COMP R5 VOUT 10 Vin 11 SYNC RAMP GENERATOR OSCILLATOR 3 RT LO 15 IRAMP (buck-boost) = 5 A x Vin + 50 A AGND 5 RAMP Q2 IRAMP (buck) = (5A x (Vin -Vout )) + 50 A CLK R3 R6 IRAMP BUCK-BOOST MODE CONTROL 6 14 PGND C3 12 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 7.3 Feature Description A buck-boost regulator can maintain regulation for input voltages either higher or lower than the output voltage. The challenge is that buck-boost power converters are not as efficient as buck regulators. The LM5118 has been designed as a dual-mode controller whereby the power converter acts as a buck regulator while the input voltage is above the output. As the input voltage approaches the output voltage, a gradual transition to the buck-boost mode occurs. The dual-mode approach maintains regulation over a wide range of input voltages, while maintaining the optimal conversion efficiency in the normal buck mode. The gradual transition between modes eliminates disturbances at the output during transitions. Figure 8 shows the basic operation of the LM5118 regulator in the buck mode. In buck mode, transistor Q1 is active and Q2 is disabled. The inductor current ramps in proportion to the VIN - VOUT voltage difference when Q1 is active and ramps down through the recirculating diode D1 when Q1 is off. The first order buck mode transfer function is VOUT/VIN = D, where D is the duty cycle of the buck switch, Q1. VIN Buck Switch Current HB Q1 HO D2 VOUT HS D1 LM5118 Diode Current CS CSG Q2 (OFF) LO Figure 8. Buck Mode Operation Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 13 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Feature Description (continued) Figure 9 shows the basic operation of buck-boost mode. In buck-boost mode both Q1 and Q2 are active for the same time interval each cycle. The inductor current ramps up (proportional to VIN) when Q1 and Q2 are active and ramps down through the recirculating diode during the off time. The first order buck-boost transfer function is VOUT/VIN = D/(1-D), where D is the duty cycle of Q1 and Q2. VIN Buck Switch Current HB Q1 HO D2 VOUT HS D1 LM5118 Diode Current CS CSG Q2 LO Figure 9. Buck-Boost Mode Operation 100 90 70 60 50 40 30 DUTY CYCLE (%) 80 20 10 18 17 16 15 14 13 12 0 VIN (V) Figure 10. Mode Dependence on Duty Cycle (VOUT = 12 V) 7.3.1 UVLO An undervoltage lockout pin is provided to disable the regulator when the input is below the desired operating range. If the UVLO pin is below 1.13 V, the regulator enters a standby mode with the outputs disabled, but with VCC regulator operating. If the UVLO input exceeds 1.23 V, the regulator will resume normal operation. A voltage divider from the input to ground can be used to set a VIN threshold to disable the regulator in brownout conditions or for low input faults. 14 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Feature Description (continued) If a current limit fault exists for more than 256 clock cycles, the regulator will enter a hiccup mode of current limiting and the UVLO pin will be pulled low by an internal switch. This switch turns off when the UVLO pin approaches ground potential allowing the UVLO pin to rise. A capacitor connected to the UVLO pin will delay the return to a normal operating level and thereby set the off-time of the hiccup mode fault protection. An internal 5A pullup current pulls the UVLO pin to a high state to ensure normal operation when the VIN UVLO function is not required and the pin is left floating. 7.3.2 Oscillator and Sync Capability The LM5118 oscillator frequency is set by a single external resistor connected between the RT pin and the AGND pin. The RT resistor should be located very close to the device and connected directly to the pins of the IC. To set a desired oscillator frequency (f), the necessary value for the RT resistor can be calculated from Equation 1: 9 RT = 6.4 x 10 3 - 3.02 x 10 f (1) The SYNC pin can be used to synchronize the internal oscillator to an external clock. The external clock must be of higher frequency than the free-running frequency set by the RT resistor. A clock circuit with an open-drain output is the recommended interface from the external clock to the SYNC pin. The clock pulse duration should be greater than 15 ns. Multiple LM5118 devices can be synchronized together simply by connecting the SYNC pins together as in Figure 11. In this configuration, all of the devices are synchronized to the highest frequency device. Figure 12 shows the SYNC input and output features of the LM5118. The internal oscillator circuit drives the SYNC pin with a strong pull down or weak pullup inverter. When the SYNC pin is pulled low, either by the internal oscillator or an external clock, the ramp cycle of the oscillator is terminated and forced 400 ns off-time is initiated before a new oscillator cycle begins. If the SYNC pins of several LM5118 IC's are connected together, the IC with the highest internal clock frequency will pull all the connected SYNC pins low and terminate the oscillator ramp cycles of the other ICs. The LM5118 with the highest programmed clock frequency will serve as the master and control the switching frequency of all the devices with lower oscillator frequencies. LM5118 SYNC LM5118 SYNC UP TO FIVE LM5118 DEVICES Figure 11. Sync From Multiple Devices Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 15 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Feature Description (continued) SYNC 100 PA I 1/RT 1.23V Q S Q R DEADTIME ONE - SHOT Figure 12. Simplified Oscillator and Block Diagram With Sync I/O Circuit 7.3.3 Error Amplifier and PWM Comparator The internal high gain error amplifier generates an error signal proportional to the difference between the regulated output voltage and an internal precision reference (1.23 V). The output of the error amplifier is connected to the COMP pin. Loop compensation components, typically a type II network shown in are connected between the COMP and FB pins. This network creates a low frequency pole, a zero, and a noise reducing high frequency pole. The PWM comparator compares the emulated current sense signal from the RAMP generator to the error amplifier output voltage at the COMP pin. The same error amplifier is used for operation in buck and buck-boost mode. v Buck: ( 5 PA x ( Vin - Vout ) + 50 PA ) x Emulated Ramp Buck - Boost: ( 5 PA x Vin + 50 PA ) x ton Cramp ton Cramp Pedestal Level = 10 x Rs (volts/amp) t Ton Figure 13. Composition of Emulated Current Signal 16 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Feature Description (continued) 7.3.4 Ramp Generator The ramp signal of a pulse-width modulator with current mode control is typically derived directly from the buck switch drain current. This switch current corresponds to the positive slope portion of the inductor current signal. Using this signal for the PWM ramp simplifies the control loop transfer function to a single pole response and provides inherent input voltage feed-forward compensation. The disadvantage of using the buck switch current signal for PWM control is the large leading edge spike due to circuit parasitics. The leading edge spike must be filtered or blanked to avoid early termination of the PWM pulse. Also, the current measurement may introduce significant propagation delays. The filtering, blanking time and propagation delay limit the minimal achievable pulse width. In applications where the input voltage may be relatively large in comparison to the output voltage, controlling a small pulse width is necessary for regulation. The LM5118 uses a unique ramp generator which does not actually measure the buck switch current but instead creates a signal representing or emulating the inductor current. The emulated ramp provides signal to the PWM comparator that is free of leading edge spikes and measurement or filtering delays. The current reconstruction is comprised of two elements, a sample-andhold pedestal level and a ramp capacitor which is charged by a controlled current source. Refer to Figure 13 for details. The sample-and-hold pedestal level is derived from a measurement of the recirculating current through a current sense resistor in series with the recirculating diode of the buck regulator stage. A small value current sensing resistor is required between the recirculating diode anode and ground. The CS and CSG pins should be Kelvin connected directly to the sense resistor. The voltage level across the sense resistor is sampled and held just prior to the onset of the next conduction interval of the buck switch. The current sensing and sample-and-hold provide the DC level of the reconstructed current signal. The sample and hold of the recirculating diode current is valid for both buck and buck-boost modes. The positive slope inductor current ramp is emulated by an external capacitor connected from the RAMP pin to the AGND and an internal voltage controlled current source. In buck mode, the ramp current source that emulates the inductor current is a function of the VIN and VOUT voltages per Equation 2: IRAMP (buck) = 5 PA x (VIN - VOUT) + 50 PA V (2) In buck-boost mode, the ramp current source is a function of the input voltage VIN, per Equation 3: IRAMP (buck - boost) = 5 PA x VIN + 50 PA V (3) Proper selection of the RAMP capacitor (CRAMP) depends upon the value of the output inductor (L) and the current sense resistor (RS). For proper current emulation, the sample and hold pedestal value and the ramp amplitude must have the same relative relationship to the actual inductor current. That is: RS x A = CRAMP = gm x L CRAMP gm x L A x RS where * * gm is the ramp generator transconductance (5 A/V) A is the current sense amplifier gain (10V/V) (4) The ramp capacitor should be located very close to the device and connected directly to the RAMP and AGND pins. Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 17 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Feature Description (continued) The relationship between the average inductor current and the pedestal value of the sampled inductor current can cause instability in certain operating conditions. This instability is known as sub-harmonic oscillation, which occurs when the inductor ripple current does not return to its initial value by the start of the next switching cycle. Sub-harmonic oscillation is normally characterized by observing alternating wide and narrow pulses at the switch node. Adding a fixed slope voltage ramp (slope compensation) to the current sense signal prevents this oscillation. The 50 A of offset current provided from the emulated current source adds enough slope compensation to the ramp signal for output voltages less than or equal to 12 V. For higher output voltages, additional slope compensation may be required. In such applications, the ramp capacitor can be decreased from the nominal calculated value to increase the ramp slope compensation. The pedestal current sample is obtained from the current sense resistor (Rs) connected to the CS and CSG pins. It is sometimes helpful to adjust the internal current sense amplifier gain (A) to a lower value in order to obtain the higher current limit threshold. Adding a pair of external resistors RG in a series with CS and CSG as in Figure 14 reduces the current sense amplifier gain A according to Equation 5: 10k A= 1k + RG (5) 7.3.5 Current Limit In the buck mode the average inductor current is equal to the output current (IOUT). In buck-boost mode the average inductor current is approximately equal to: Iout x 1 + VOUT VIN (6) Consequently, the inductor current in buck-boost mode is much larger especially when VOUT is large relative to VIN. The LM5118 provides a current monitoring scheme to protect the circuit from possible over-current conditions. When set correctly, the emulated current sense signal is proportional to the buck switch current with a scale factor determined by the current sense resistor. The emulated ramp signal is applied to the current limit comparator. If the peak of the emulated ramp signal exceeds 1.25 V when operating in the buck mode, the PWM cycle is immediately terminated (cycle-by-cycle current limiting). In buck-boost mode the current limit threshold is increased to 2.50 V to allow higher peak inductor current. To further protect the external switches during prolonged overload conditions, an internal counter detects consecutive cycles of current limiting. If the counter detects 256 consecutive current limited PWM cycles, the LM5118 enters a low power dissipation hiccup mode. In the hiccup mode, the output drivers are disabled, the UVLO pin is momentarily pulled low, and the soft-start capacitor is discharged. The regulator is restarted with a normal soft-start sequence once the UVLO pin charges back to 1.23 V. The hiccup mode off-time can be programmed by an external capacitor connected from UVLO pin to ground. This hiccup cycle will repeat until the output overload condition is removed. In applications with low output inductance and high input voltage, the switch current may overshoot due to the propagation delay of the current limit comparator and control circuitry. If an overshoot should occur, the sampleand-hold circuit will detect the excess recirculating diode current. If the sample-and-hold pedestal level exceeds the internal current limit threshold, the buck switch will be disabled and will skip PWM cycles until the inductor current has decayed below the current limit threshold. This approach prevents current runaway conditions due to propagation delays or inductor saturation since the inductor current is forced to decay before the buck switch is turned on again. 18 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Feature Description (continued) CURRENT LIMIT COMPARATOR CURRENT SENSE AMPLIFIER Vth TRACK and HOLD - 10k 1k CS RG + RS + IRAMP 1k 10k CSG IL RG 0.2V RAMP RESET RAMP CRAMP Figure 14. Current Limit and Ramp Circuit 7.3.6 Maximum Duty Cycle Each conduction cycle of the buck switch is followed by a forced minimum off-time of 400 ns to allow sufficient time for the recirculating diode current to be sampled. This forced off-time limits the maximum duty cycle of the controller. The actual maximum duty cycle will vary with the operating frequency as follows: DMAX = 1 - f x 400 x 10-9 where * f is the oscillator frequency in Hz (7) 1 MAX DUTY CYCLE 0.95 0.9 0.85 0.8 0.75 0 100 200 300 400 500 600 FREQUENCY (kHz) Figure 15. Maximum Duty Cycle vs Frequency Limiting the maximum duty cycle will limit the maximum boost ratio (VOUT/VIN) while operating in buck-boost mode. For example, from Figure 15, at an operating frequency of 500 kHz, DMAX is 80%. Using the buck-boost transfer function. Vout D= Vin + Vout (8) With D = 80%, solving for VOUT results in: VOUT = 4 x VIN (9) Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 19 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Feature Description (continued) With a minimum input voltage of 5 V, the maximum possible output voltage is 20 V at f = 500 kHz. The buckboost step-up ratio can be increased by reducing the operating frequency which increases the maximum duty cycle. 7.3.7 Soft Start The soft-start feature allows the regulator to gradually reach the initial steady-state operating point, thus reducing start-up stresses and surges. The internal 10-A soft-start current source gradually charges an external soft-start capacitor connected to the SS pin. The SS pin is connected to the positive input of the internal error amplifier. The error amplifier controls the pulse-width modulator such that the FB pin approximately equals the SS pin as the SS capacitor is charged. Once the SS pin voltage exceeds the internal 1.23-V reference voltage, the error amp is controlled by the reference instead of the SS pin. The SS pin voltage is clamped by an internal amplifier at a level of 150 mV above the FB pin voltage. This feature provides a soft-start controlled recovery in the event a severe overload pulls the output voltage (and FB pin) well below normal regulation but does not persist for 256 clock cycles. Various sequencing and tracking schemes can be implemented using external circuits that limit or clamp the voltage level of the SS pin. The SS pin acts as a non-inverting input to the error amplifier anytime SS voltage is less than the 1.23-V reference. In the event a fault is detected (overtemperature, VCC undervoltage, hiccup current limit), the soft-start capacitor will be discharged. When the fault condition is no longer present, a new softstart sequence will begin. 7.3.8 HO Output The LM5118 contains a high-side, high-current gate driver and associated high voltage level shift. This gate driver circuit works in conjunction with an internal diode and an external bootstrap capacitor. A 0.1-F ceramic capacitor, connected with short traces between the HB pin and HS pin is recommended for most circuit configurations. The size of the bootstrap capacitor depends on the gate charge of the external FET. During the off time of the buck switch, the HS pin voltage is approximately -0.5 V and the bootstrap capacitor is charged from VCC through the internal bootstrap diode. When operating with a high PWM duty cycle, the buck switch will be forced off each cycle for 400 ns to ensure that the bootstrap capacitor is recharged. 7.3.9 Thermal Protection Internal Thermal Shutdown circuitry is provided to protect the integrated circuit in the event the maximum junction temperature is exceeded. When activated, typically at 165C, the controller is forced into a low power reset state, disabling the output driver and the bias regulator. This protection is provided to prevent catastrophic failures from accidental device overheating. 7.4 Device Functional Modes Figure 10 shows how duty cycle effects the operational mode and is useful for reference in the following discussions. Initially, only the buck switch is active and the buck duty cycle increases to maintain output regulation as VIN decreases. When VIN is approximately equal to 15.5 V, the boost switch begins to operate with a low duty cycle. If VIN continues to fall, the boost switch duty cycle increases and the buck switch duty cycle decreases until they become equal at VIN = 13.2 V. 7.4.1 Buck Mode Operation: VIN > VOUT The LM5118 buck-boost regulator operates as a conventional buck regulator with emulated current mode control while VIN is greater than VOUT and the buck mode duty cycle is less than 75%. In buck mode, the LO gate drive output to the boost switch remains low. 20 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Device Functional Modes (continued) 7.4.2 Buck-Boost Mode Operation: VIN VOUT When VIN decreases relative to VOUT, the duty cycle of the buck switch will increase to maintain regulation. Once the duty cycle reaches 75%, the boost switch starts to operate with a very small duty cycle. As VIN is further decreased, the boost switch duty cycle increases until it is the same as the buck switch. As VIN is further decreased below VOUT, the buck and boost switch operate together with the same duty cycle and the regulator is in full buck-boost mode. This feature allows the regulator to transition smoothly from buck to buck-boost mode. Note that the regulator can be designed to operate with VIN less than 4 V, but VIN must be at least 5 V Figure 16 presents a timing illustration of the gradual transition from buck to buck-boost mode when the input voltage ramps downward over a few switching cycles. Figure 16. Buck (HO) and Boost (LO) Switch Duty Cycle vs. Time, Illustrating Gradual Mode Change With Decreasing Input Voltage 7.4.3 High Voltage Start-Up Regulator The LM5118 contains a dual-mode, high voltage linear regulator that provides the VCC bias supply for the PWM controller and the MOSFET gate driver. The VIN input pin can be connected directly to input voltages as high as 75 V. For input voltages below 10 V, an internal low dropout switch connects VCC directly to VIN. In this supply range, VCC is approximately equal to VIN. For VIN voltages greater than 10 V, the low dropout switch is disabled and the VCC regulator is enabled to maintain VCC at approximately 7 V. A wide operating range of 4 V to 75 V (with a start-up requirement of at least 5 V) is achieved through the use of this dual mode regulator. The output of the VCC regulator is current limited to 35 mA, typical. Upon power up, the regulator sources current into the capacitor connected to the VCC pin. When the voltage at the VCC pin exceeds the VCC undervoltage threshold of 3.7 V and the UVLO input pin voltage is greater than 1.23 V, the gate driver outputs are enabled and a soft-start sequence begins. The gate driver outputs remain enabled until VCC falls below 3.5 V or the voltage at the UVLO pin falls below 1.13 V. Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 21 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Device Functional Modes (continued) In many applications, the regulated output voltage or an auxiliary supply voltage can be applied to the VCCX pin to reduce the IC power dissipation. For output voltages between 4 V and 15 V, VOUT can be connected directly to VCCX. When the voltage at the VCCX pin is greater than 3.85 V, the internal VCC regulator is disabled and an internal switch connects VCCX to VCC, reducing the internal power dissipation. In high voltage applications, take extrac care to ensure the VIN pin voltage does not exceed the absolute maximum voltage rating of 76 V. During line or load transients, voltage ringing on the VIN line that exceeds the absolute maximum rating can damage the IC. Both careful PCB layout and the use of quality bypass capacitors located close to the VIN and GND pins are essential. Vin 10V 7V 3.7V Vcc 3.5V VIN and VCC Internal Enable Signal Figure 17. VIN and VCC Sequencing 7.4.4 Enable The LM5118 contains an enable function which provides a very low input current shutdown mode. If the EN pin is pulled below 0.5 V, the regulator enters shutdown mode, drawing less than 10 A from the VIN pin. Raising the EN input above 3 V returns the regulator to normal operation. The EN pin can be tied directly to the VIN pin if this function is not needed. It must not be left floating. A 1-M pullup resistor to VIN can be used to interface with an open-collector or open-drain control signal. 22 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 8 Application and Implementation NOTE Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI's customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality. 8.1 Application Information The LM5118 high voltage switching regulator features all of the functions necessary to implement an efficient high voltage buck or buck-boost regulator using a minimum of external components. A buck-boost regulator can maintain regulation for input voltages either higher or lower than the output voltage. 8.2 Typical Application VIN VCC HB LM5118 HO VOUT HS SS CS RAMP CSG LO RT VOUT FB GND COMP Figure 18. Simplified Application Schematic Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 23 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Typical Application (continued) Figure 19. 12-V, 3-A Typical Application Schematic 24 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Typical Application (continued) 8.2.1 Design Requirements The procedure for calculating the external components is illustrated with the following design example. The designations used in the design example correlate to the Figure 19. The design specifications are: * VOUT = 12 V * VIN = 5 V to 42 V * f = 300 kHz * Minimum load current (CCM operation) = 600 mA * Maximum load current = 3 A 8.2.2 Detailed Design Procedure 8.2.2.1 Custom Design With WEBENCH(R) Tools Click here to create a custom design using the LM5118 device with the WEBENCH(R) Power Designer. 1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements. 2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial. 3. Compare the generated design with other possible solutions from Texas Instruments. The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time pricing and component availability. In most cases, these actions are available: * Run electrical simulations to see important waveforms and circuit performance * Run thermal simulations to understand board thermal performance * Export customized schematic and layout into popular CAD formats * Print PDF reports for the design, and share the design with colleagues Get more information about WEBENCH tools at www.ti.com/WEBENCH. 8.2.2.2 R7 = RT RT sets the oscillator switching frequency. Generally speaking, higher operating frequency applications will use smaller components, but have higher switching losses. An operating frequency of 300 kHz was selected for this example as a reasonable compromise for both component size and efficiency. The value of RT can be calculated as follows: 9 RT = 6.4 x 10 3 - 3.02 x 10 f (10) therefore, R7 = 18.3 k Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 25 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Typical Application (continued) Ipk+ L1 Current Io Buck Iripple Io/(1-D) B-B Ipk- Iripple 1/Fs Figure 20. Inductor Current Waveform 26 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Typical Application (continued) 8.2.2.3 Inductor Selection, L1 The inductor value is determined based upon the operating frequency, load current, ripple current and the input and output voltages. Refer to Figure 20 for details. To keep the circuit in continuous conduction mode (CCM), the maximum ripple current IRIPPLE should be less than twice the minimum load current. For the specified minimum load of 0.6 A, The maximum ripple current is 1.2 Ap-p. Also, the minimum value of L must be calculated both for a buck and buck-boost configurations. The final value of inductance will generally be a compromise between the two modes. It is desirable to have a larger value inductor for buck mode, but the saturation current rating for the inductor must be large for buck-boost mode, resulting in a physically large inductor. Additionally, large value inductors present buck-boost mode loop compensation challenges which will be discussed in the Error Amplifier Configuration section. For the design example, the inductor values in both modes are calculated as: VOUT (VIN(MAX) VOUT ) L1 Buck Mode VIN(MAX) u f u IRIPPLE (11) L1 VIN(MIN) (VOUT ) (VOUT VIN(MIN) ) u f u IRIPPLE Buck-Boost Mode where * * * * * * VOUT is the output voltage VIN(MAX) is the maximum input voltage f is the switching frequency IRIPPLE is the selected inductor peak to peak ripple current (1.2 A selected for this example) VIN(MIN) is the minimum input voltage (12) The resulting inductor values are: * L1 = 28 H, Buck Mode * L1 = 9.8 H Buck-Boost mode A 10-H inductor was selected which is a compromise between these values, while favoring the buck-boost mode. As illustrated in the compensation section below, the inductor value should be as low as possible to move the buck-boost right-half-plane zero to a higher frequency. The ripple current is then rechecked with the selected inductor value using Equation 11 and Equation 12, IRIPPLE(BUCK) = 3.36 A IRIPPLE(BUCK-BOOST) = 1.17 A (13) (14) Because the inductor selected is lower than calculated for the Buck mode, the minimum load current for CCM in buck mode is 1.68 A at maximum VIN. With a 10-H inductor, the worst case peak inductor currents can be estimated for each case, assuming a 20% inductor value tolerance and assuming 80% efficiency of the converter. IRIPPLE(BUCK) I I1(PEAK) OUT K 2(1 LTOL ) (15) I2(PEAK) IOUT (VOUT VIN(MIN) ) IRIPPLE(BUCK-BOOST) Ku VIN(MIN) 2(1 LTOL ) where * * is efficiency LTOL is the inductor tolerance (16) Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 27 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Typical Application (continued) For this example, Equation 15 and Equation 16 yield: I1(PEAK) = 5.62 A I2(PEAK) = 13.4 A (17) (18) An acceptable current limit setting would be 6.7 A for buck mode because the LM5118 automatically doubles the current limit threshold in buck-boost mode. The selected inductor must have a saturation current rating at least as high as the buck-boost mode cycle-by-cycle current limit threshold, in this case at least 13.5 A. A 10-H, 15-A inductor was chosen for this application. 8.2.2.4 R13 = RSENSE To select the current sense resistor, begin by calculating the minimum K values for each mode using Equation 19 and Equation 20. K represents the slope compensation of the controller and is different for each mode, KBUCK and KBUCK-BOOST. 10 KBUCK t 1 VIN(MAX) VOUT (19) KBUCK * * BOOST t1 10 VIN(MIN) (20) KBUCK = 1.16 KBUCK-BOOST = 3 Use Equation 21 and Equation 22 to calculate RSENSE for each mode of operation. A design margin, M, should be selected between 10%-30% to allow for component tolerances. For this design M was selected to be 10%. R13(BUCK) I 10 OUT (c) K R13(BUCK-BOOST) * * 1.25(1 M) IRIPPLE(BUCK) 2 * K BUCK (21) 2.5 (1 M) VIN(MIN) VOUT IOUT 10 VIN(MIN) K (c) IRIPPLE(BUCK-BOOST) 2 * K BUCK-BOOST (22) R13(BUCK) = 19.75 m R13(BUCK-BOOST) = 15.5 m An RSENSE value of no more than 15.5 m must be used to ensure the required maximum output current in the buck-boost mode. A standard value of 15 m was selected for this design. 8.2.2.5 C15 = CRAMP With the inductor value selected, the value of C3 necessary for the emulation ramp circuit is: -6 C15 = CRAMP = L x 10 2 x RSENSE (23) With the inductance value (L1) selected as 10 H, the calculated value for CRAMP is 333 pF. A standard value of 330 pF was selected. 28 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Typical Application (continued) 8.2.2.6 Inductor Current Limit Calculation The current limit for each mode can be calculated using Equation 24 and Equation 26. If the peak current limit is less than the calculated inductor peak current the R13 and C15 need to be recalculated. This can be done by increasing the previous K values or M and reiterating the calculations. 50 u 10 6 u VOUT C15 u f u VIN(MAX) 1.25 ILIMIT(BUCK) (24) (25) 10 u R13 ILIMIT(BUCK) = 7.795 A 2.5 ILIMIT(BUCK-BOOST) 50 u 10 6 u VOUT C15 u f u (VIN(MIN) VOUT ) (26) (27) 10 u R13 ILIMIT(BUCK-BOOST) = 14.29 A 8.2.2.7 C9 - C12 = Output Capacitors In buck-boost mode, the output capacitors C9 - C12 must supply the entire output current during the switch ontime. For this reason, the output capacitors are chosen for operation in buck-boost mode, the demands being much less in buck operation. Both bulk capacitance and ESR must be considered to ensure a given output ripple voltage. Buck-boost mode capacitance can be estimated from: CMIN IOUT u DMAX with DMAX f u 'VOUT VOUT VIN(MIN) VOUT (28) ESR requirements can be estimated from: ESRMAX VOUT VIN(MIN) VIN(MIN) 'VOUT IRIPPLE(BUCK-BOOST) IOUT 2 (29) For this example, with a VOUT (output ripple) of 50 mV: CMIN = 141 F ESRMAX = 4.6 m (30) (31) If hold-up times are a consideration, the values of the input and output capacitors must be increased appropriately. Note that it is usually advantageous to use multiple capacitors in parallel to achieve the ESR value required. Also, it is good practice to put a .1-F to .47-F ceramic capacitor directly on the output pins of the supply to reduce high-frequency noise. Ceramic capacitors have good ESR characteristics, and are a good choice for input and output capacitors. It should be noted that the effective capacitance of ceramic capacitors decreases with dc bias. For larger bulk values of capacitance, a low ESR electrolytic is usually used. However, electrolytic capacitors have poor tolerance, especially over temperature, and the selected value should be selected larger than the calculated value to allow for temperature variation. Allowing for component tolerances, the following values of COUT were chosen for this design example: Two 180-F Oscon electrolytic capacitors for bulk capacitance Two 47-F ceramic capacitors to reduce ESR Two 0.47-F ceramic capacitors to reduce spikes at the output. Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 29 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Typical Application (continued) 8.2.2.8 D1 Reverse recovery currents degrade performance and decrease efficiency. For these reasons, a Schottky diode of appropriate ratings should be used for D1. The voltage rating of the boost diode should be equal to VOUT plus some margin. D1 conducts continually in buck mode and only when the buck switch is off in Buck-Boost mode. 8.2.2.9 D4 A Schottky type recirculating diode is required for all LM5118 applications. The near ideal reverse recovery characteristics and low forward voltage drop are particularly important diode characteristics for high input voltage and low output voltage applications. The reverse recovery characteristic determines how long the current surge lasts each cycle when the buck switch is turned on. The reverse recovery characteristics of Schottky diodes minimize the peak instantaneous power in the buck switch during the turnon transition. The reverse breakdown rating of the diode should be selected for the maximum VIN plus some safety margin. The forward voltage drop has a significant impact on the conversion efficiency, especially for applications with a low output voltage. Rated current for diodes vary widely from various manufacturers. For the LM5118 this current is user selectable through the current sense resistor value. Assuming a worst case 0.6-V drop across the diode, the maximum diode power dissipation can be high. The diode should have a voltage rating of VIN and a current rating of IOUT. A conservative design would at least double the advertised diode rating since specifications between manufacturers vary. For the reference design a 100-V, 10-A Schottky in a D2PAK package was selected. 8.2.2.10 C1 - C5 = Input Capacitor A typical regulator supply voltage has a large source impedance at the switching frequency. Good-quality input capacitors are necessary to limit the ripple voltage at the VIN pin while supplying most of the switch current during the buck switch on-time. When the buck switch turns on, the current into the buck switch steps from zero to the lower peak of the inductor current waveform, then ramps up to the peak value, and then drops to the zero at turnoff. The RMS current rating of the input capacitors depends on which mode of operation is most critical. IRMS(BUCK) = IOUT D(1 -D) (32) This value is a maximum at 50% duty cycle which corresponds to VIN = 24 V. IRMS(BUCK-BOOST) = IOUT D(1 -D) 1-D (33) Checking both modes of operation we find: * IRMS(BUCK) = 1.5 A * IRMS(BUCK-BOOST) = 4.7 A Therefore C1-C5 should be sized to handle 4.7 A of ripple current. Quality ceramic capacitors with a low ESR should be selected. To allow for capacitor tolerances, five 2.2-F, 100-V ceramic capacitors will be used. If step input voltage transients are expected near the maximum rating of the LM5118, a careful evaluation of the ringing and possible spikes at the device VIN pin should be completed. An additional damping network or input voltage clamp may be required in these cases. 8.2.2.11 C20 The capacitor at the VCC pin provides noise filtering and stability for the VCC regulator. The recommended value of C20 should be no smaller than 0.1 F, and should be a good-quality, low-ESR, ceramic capacitor. A value of 1 F was selected for this design. C20 should be 10 x C8. If operating without VCCX, then fOSC x (QCBuck + Boost) + ILOAD(INTERNAL) (34) must be less than the VCC current limit. 30 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Typical Application (continued) 8.2.2.12 C8 The bootstrap capacitor between the HB and HS pins supplies the gate current to charge the buck switch gate at turnon. The recommended value of C8 is 0.1 F to 0.47 F, and should be a good-quality, low-ESR, ceramic capacitor. A value of 0.1 F was chosen for this design. 8.2.2.13 C16 = CSS The capacitor at the SS pin determines the soft-start time, that is, the time for the reference voltage and the output voltage, to reach the final regulated value. The time is determined from: tSS = C16 x 1.23V 10 PA (35) and assumes a current limit > Iload + ICout For this application, a C16 value of 0.1 F was chosen which corresponds to a soft-start time of about 12 ms. 8.2.2.14 R8, R9 R8 and R9 set the output voltage level, the ratio of these resistors is calculated from: R8 VOUT - 1 = R9 1.23V (36) For a 12-V output, the R8/R9 ratio calculates to 8.76. The resistors should be chosen from standard value resistors and a good starting point is to select resistors within power ratings appropriate for the output voltage. Values of 309 for R9 and 2.67 k for R8 were selected. 8.2.2.15 R1, R3, C21 A voltage divider can be connected to the UVLO pin to set a minimum operating voltage VIN(UVLO) for the regulator. If this feature is required, the easiest approach to select the divider resistor values is to choose a value for R1 between 10 k and 100 k, while observing the minimum value of R1 necessary to allow the UVLO switch to pull the UVLO pin low. This value is: R1 1000 x VIN(MAX) R1 75 k in our example R3 is then calculated from: R3 = 1.23 x R1 VIN(MIN) + 5 PA x R1 - 1.23 (37) Because VIN(MIN) for our example is 5 V, set VIN(UVLO) to 4.0 V for some margin in component tolerances and input ripple. R1 = 75 k is chosen since it is a standard value. R3 = 29.332 k is calculated from Equation 37. The 29.4 k value was used since it is a standard value. Capacitor C21 provides filtering for the divider and the off-time of the hiccup duty cycle during current limit. The voltage at the UVLO pin should never exceed 15 V when using an external set-point divider. It may be necessary to clamp the UVLO pin at high input voltages. Knowing the desired off time during hiccup current limit, the value of C21 is given by: t OFF C21 R1 R3 R1 R3 Ln 1 .98 R1 R3 VIN R3 1/4 (38) Notice that tOFF varies with VIN In this example, C21 was chosen to be 0.1 F. This will set the tOFF time to 723 s with VIN = 12 V. Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 31 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Typical Application (continued) 8.2.2.16 R2 A 1-M pullup resistor connected from the EN pin to the VIN pin is sufficient to keep enable in a high state if on-off control is not used. 8.2.2.17 Snubber A snubber network across the buck recirculating diode reduces ringing and spikes at the switching node. Excessive ringing and spikes can cause erratic operation and increase noise at the regulator output. In the limit, spikes beyond the maximum voltage rating of the LM5118 or the recirculating diode can damage these devices. Selecting the values for the snubber is best accomplished through empirical methods. First, make sure the lead lengths for the snubber connections are very short. Start with a resistor value between 5 and 20 . Increasing the value of the snubber capacitor results in more damping, however the snubber losses increase. Select a minimum value of the capacitor that provides adequate clamping of the diode waveform at maximum load. A snubber may be required for the boost diode as well. The same empirical procedure applies. Snubbers were not necessary in this example. 8.2.2.18 Error Amplifier Configuration 8.2.2.18.1 R4, C18, C17 These components configure the error amplifier gain characteristics to accomplish a stable overall loop gain. One advantage of current mode control is the ability to close the loop with only three feedback components, R4, C18 and C17. The overall loop gain is the product of the modulator gain and the error amplifier gain. The DC modulator gain of the LM5118 is as follows: RLOAD x VIN DCGain(MOD) = 10RS(VIN + 2VOUT) (39) The dominant, low frequency pole of the modulator is determined by the load resistance (RLOAD) and output capacitance (COUT). The corner frequency of this pole is: 1 DMAX fP(MOD) 2S u RLOAD u COUT (40) For this example, RLOAD = 4 , DMAX = 0.705, and COUT = 454 F, therefore: fP(MOD) = 149 Hz DC Gain(MOD) = 4.598 = 13.25 dB (41) (42) Additionally, there is a right-half plane (RHP) zero associated with the modulator. The frequency of the RHP zero is: fRHPzero = RLOAD (1 - D)2 2S x L x D (43) (44) fRHPzero = 7.8 kHz The output capacitor ESR produces a zero given by: ESRzero = 1 2S x ESR x COUT (45) (46) ESRZERO = 76 kHz The RHP zero complicates compensation. The best design approach is to reduce the loop gain to cross zero at about 25% of the calculated RHP zero frequency. The Type ll error amplifier compensation provided by R4, C18, and C17 places one pole at the origin for high DC gain. The 2nd pole should be located close to the RHP zero. The error amplifier zero (Equation 47) should be placed near the dominate modulator pole. This is a good starting point for compensation. Refer to the on-line LM5118 Quick-Start calculator for ready to use equations and more details. 32 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Typical Application (continued) Components R4 and C18 configure the error amplifier as a type II configuration which has a DC pole and a zero at: 1 fz = 2 x S x R4 x C18 (47) C17 introduces an additional pole used to cancel high-frequency switching noise. The error amplifier zero cancels the modulator pole leaving a single pose response at the crossover frequency of the loop gain if the crossover frequency is much lower than the right half plane zero frequency. A single pole response at the crossover frequency yields a very stable loop with 90 degrees of phase margin. For the design example, a target loop bandwidth (crossover frequency) of 2.0 kHz was selected (about 25% of the right-half-plane zero frequency). The error amplifier zero (fz) should be selected at a frequency near that of the modulator pole and much less than the target crossover frequency. This constrains the product of R4 and C18 for a desired compensation network zero to be less than 2 kHz. Increasing R4, while proportionally decreasing C18 increases the error amp gain. Conversely, decreasing R4 while proportionally increasing C18 decreases the error amp gain. For the design example C18 was selected for 100 nF and R4 was selected to be 10 k. These values set the compensation network zero at 159 Hz. The overall loop gain can be predicted as the sum (in dB) of the modulator gain and the error amp gain. If a network analyzer is available, the modulator gain can be measured and the error amplifier gain can be configured for the desired loop transfer function. If a network analyzer is not available, the error amplifier compensation components can be designed with the guidelines given. Step load transient tests can be performed to verify acceptable performance. The step load goal is minimal overshoot with a damped response. Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 33 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com Typical Application (continued) 8.2.3 Application Curves The plots in Figure 21 through Figure 26 show the gain and phase diagrams of the design example. The overall bandwidth is lower in a buck-boost application due the compensation challenges associated with the right-halfplane zero. For a pure buck application, the bandwidth could be much higher. The LM5116 data sheet is a good reference for compensation design of a pure buck mode regulator. 34 Figure 21. Modulator Gain and Phase - Buck Mode Figure 22. Modulator Gain and Phase - Buck-Boost Mode Figure 23. Error Amplifier Gain and Phase - Buck Mode Figure 24. Error Amplifier Gain and Phase - Buck-Boost Mode Figure 25. Overall Loop Gain and Phase - Buck Mode Figure 26. Overall Loop Gain and Phase - Buck Boost Mode Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 9 Power Supply Recommendations 9.1 Thermal Considerations The highest power dissipating components are the two power MOSFETs, the recirculating diode, and the output diode. The easiest way to determine the power dissipated in the MOSFETs is to measure the total conversion losses (PIN - POUT), then subtract the power losses in the Schottky diodes, output inductor and any snubber resistors. An approximation for the recirculating Schottky diode loss is: P = (1 - D) x IOUT x VFWD. (48) The boost diode loss is P = IOUT x VFWD. (49) If a snubber is used, the power loss can be estimated with an oscilloscope by observation of the resistor voltage drop at both turnon and turnoff transitions. The LM5118 package has an exposed thermal pad to aid power dissipation. Selecting diodes with exposed pads will aid the power dissipation of the diodes as well. When selecting the MOSFETs, pay careful attention to RDS(ON) at high temperature. Also, selecting MOSFETs with low gate charge will result in lower switching losses. Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 35 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com 9.2 Bias Power Dissipation Reduction Buck or Buck-boost regulators operating with high input voltage can dissipate an appreciable amount of power while supplying the required bias current of the IC. The VCC regulator must step-down the input voltage VIN to a nominal VCC level of 7 V. The large voltage drop across the VCC regulator translates into high power dissipation in the VCC regulator. There are several techniques that can significantly reduce this bias regulator power dissipation. Figure 27 and Figure 28 depict two methods to bias the IC, one from the output voltage and one from a separate bias supply. In the first case, the internal VCC regulator is used to initially bias the VCC pin. After the output voltage is established, the VCC pin bias current is supplied through the VCCX pin, which effectively disables the internal VCC regulator. Any voltage greater than 4.0 V can supply VCC bias through the VCCX pin. However, the voltage applied to the VCCX pin should never exceed 15 V. The voltage supplied through VCCX must be large enough to drive the switching MOSFETs into full saturation. HB VIN HO Q1 D2 VOUT L1 HS D1 LM5118 Cout CS CSG LO Q2 GND VCC VCCX Figure 27. VCC Bias From VOUT 4 V < VOUT < 15 V HB VIN Q1 D2 HO VOUT L1 HS LM5118 D1 CS Cout Q2 LO VOUT CSG VCC VCCX VBIAS Figure 28. VCC Bias With Additional Bias Supply 36 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 Bias Power Dissipation Reduction (continued) In a buck-boost regulator, there are two loops where currents are switched very fast. The first loop starts from the input capacitors, and then to the buck switch, the inductor, the boost switch then back to the input capacitor. The second loop starts from the inductor, and then to the output diode, the output capacitor, the recirculating diode, and back to the inductor. Minimizing the PCB area of these two loops reduces the stray inductance and minimizes noise and the possibility of erratic operation. A ground plane in the PCB is recommended as a means to connect the input filter capacitors to the output filter capacitors and the PGND pins of the LM5118. Connect all of the low current ground connections (CSS, RT, CRAMP) directly to the regulator AGND pin. Connect the AGND and PGND pins together through topside copper area covering the entire underside of the device. Place several vias in this underside copper area to the ground plane of the input capacitors. Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 37 LM5118 SNVS566J - APRIL 2008 - REVISED JUNE 2017 www.ti.com 10 Layout 10.1 Layout Guidelines In a buck-boost regulator, there are two loops where currents are switched very fast. The first loop starts from the input capacitors, and then to the buck switch, the inductor, the boost switch then back to the input capacitor. The second loop starts from the inductor, and then to the output diode, the output capacitor, the re-circulating diode, and back to the inductor. Minimizing the PCB area of these two loops reduces the stray inductance and minimizes noise and the possibility of erratic operation. A ground plane in the PCB is recommended as a means to connect the input filter capacitors to the output filter capacitors and the PGND pins of the LM5118. Connect all of the low current ground connections (CSS, RT, CRAMP) directly to the regulator AGND pin. Connect the AGND and PGND pins together through topside copper area covering the entire underside of the device. Place several vias in this underside copper area to the ground plane of the input capacitors. 10.2 Layout Example Inductor DBUCK QBOOST DBOOST QBUCK RSENSE COUT CIN CIN VIN COUT Controller GND GND VOUT Figure 29. LM5118 Layout Example 38 Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 LM5118 www.ti.com SNVS566J - APRIL 2008 - REVISED JUNE 2017 11 Device and Documentation Support 11.1 Device Support 11.1.1 Development Support 11.1.1.1 Custom Design With WEBENCH(R) Tools Click here to create a custom design using the LM5118 device with the WEBENCH(R) Power Designer. 1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements. 2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial. 3. Compare the generated design with other possible solutions from Texas Instruments. The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time pricing and component availability. In most cases, these actions are available: * Run electrical simulations to see important waveforms and circuit performance * Run thermal simulations to understand board thermal performance * Export customized schematic and layout into popular CAD formats * Print PDF reports for the design, and share the design with colleagues Get more information about WEBENCH tools at www.ti.com/WEBENCH. 11.2 Trademarks WEBENCH is a registered trademark of Texas Instruments. All other trademarks are the property of their respective owners. 11.3 Electrostatic Discharge Caution These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 11.4 Glossary SLYZ022 -- TI Glossary. This glossary lists and explains terms, acronyms, and definitions. 12 Mechanical, Packaging, and Orderable Information The following pages include mechanical, packaging, and orderable information. This information is the most current data available for the designated devices. This data is subject to change without notice and revision of this document. For browser-based versions of this data sheet, refer to the left-hand navigation. Submit Documentation Feedback Copyright (c) 2008-2017, Texas Instruments Incorporated Product Folder Links: LM5118 39 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (C) Device Marking (3) (4/5) (6) LM5118MH/NOPB ACTIVE HTSSOP PWP 20 73 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM5118 MH LM5118MHX/NOPB ACTIVE HTSSOP PWP 20 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM5118 MH (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based flame retardants must also meet the <=1000ppm threshold requirement. (3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device. (5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation of the previous line and the two combined represent the entire Device Marking for that device. (6) Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two lines if the finish value exceeds the maximum column width. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. Addendum-Page 1 Samples PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 OTHER QUALIFIED VERSIONS OF LM5118 : * Automotive: LM5118-Q1 NOTE: Qualified Version Definitions: * Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects Addendum-Page 2 PACKAGE MATERIALS INFORMATION www.ti.com 14-Sep-2017 TAPE AND REEL INFORMATION *All dimensions are nominal Device LM5118MHX/NOPB Package Package Pins Type Drawing SPQ HTSSOP 2500 PWP 20 Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) 330.0 16.4 Pack Materials-Page 1 6.95 B0 (mm) K0 (mm) P1 (mm) 7.1 1.6 8.0 W Pin1 (mm) Quadrant 16.0 Q1 PACKAGE MATERIALS INFORMATION www.ti.com 14-Sep-2017 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LM5118MHX/NOPB HTSSOP PWP 20 2500 367.0 367.0 35.0 Pack Materials-Page 2 MECHANICAL DATA PWP0020A MXA20A (Rev C) www.ti.com IMPORTANT NOTICE AND DISCLAIMER TI PROVIDES TECHNICAL AND RELIABILITY DATA (INCLUDING DATASHEETS), DESIGN RESOURCES (INCLUDING REFERENCE DESIGNS), APPLICATION OR OTHER DESIGN ADVICE, WEB TOOLS, SAFETY INFORMATION, AND OTHER RESOURCES "AS IS" AND WITH ALL FAULTS, AND DISCLAIMS ALL WARRANTIES, EXPRESS AND IMPLIED, INCLUDING WITHOUT LIMITATION ANY IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE OR NON-INFRINGEMENT OF THIRD PARTY INTELLECTUAL PROPERTY RIGHTS. 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