LTC3890-1
1
38901fa
TYPICAL APPLICATION
FEATURES
APPLICATIONS
DESCRIPTION
60V Low IQ, Dual, 2-Phase
Synchronous Step-Down
DC/DC Controller
High Efficiency Dual 8.5V/3.3V Output Step-Down Converter
n Wide VIN Range: 4V to 60V (65V Abs Max)
n Low Operating IQ: 50μA (One Channel On)
n Wide Output Voltage Range: 0.8V ≤ VOUT ≤ 24V
n RSENSE or DCR Current Sensing
n Out-of-Phase Controllers Reduce Required Input
Capacitance and Power Supply Induced Noise
n Phase-Lockable Frequency (75kHz to 850kHz)
n Programmable Fixed Frequency (50kHz to 900kHz)
n Selectable Continuous, Pulse-Skipping or Low Ripple
Burst Mode
®
Operation at Light Loads
n Very Low Dropout Operation: 99% Duty Cycle
n Adjustable Output Voltage Soft-Start or Tracking
n Power Good Output Voltage Monitor
n Output Overvoltage Protection
n Low Shutdown IQ: <14µA
n Internal LDO Powers Gate Drive from VIN or EXTVCC
n No Current Foldback During Start-Up
n Narrow SSOP Package
n Automotive Always-On Systems
n Battery Operated Digital Devices
n Distributed DC Power Systems
Efficiency and Power Loss
vs Output Current
The LTC
®
3890-1 is a high performance dual step-down
switching regulator DC/DC controller that drives all
N-channel synchronous power MOSFET stages. A constant
frequency current mode architecture allows a phase-
lockable frequency of up to 850kHz. Power loss and supply
noise are minimized by operating the two controller output
stages out-of-phase.
The 50A no-load quiescent current extends operating life
in battery-powered systems. OPTI-LOOP
®
compensation
allows the transient response to be optimized over a wide
range of output capacitance and ESR values. A wide 4V
to 60V input supply range encompasses a wide range of
intermediate bus voltages and battery chemistries.
Independent TRACK/SS pins for each controller ramp the
output voltages during start-up. Current foldback limits
MOSFET heat dissipation during short-circuit conditions.
The PLLIN/MODE pin selects among Burst Mode operation,
pulse-skipping mode, or continuous conduction mode at
light loads.
For a leadless 32-pin QFN package with additional features
of adjustable current limit, clock out, phase modulation
and two PGOOD outputs, see the LTC3890 data sheet.
L, LT, LTC, LTM, Linear Technology, Burst Mode, OPTI-LOOP and the Linear logo are
registered trademarks of Linear Technology Corporation. All other trademarks are the property
of their respective owners. Protected by U.S. Patents including 5481178, 5705919, 5929620,
6100678, 6144194, 6177787, 6304066, 6580258, 7230497.
0.1µF
100k
4.7µH
1000pF
470µF
4.7µF 22µF
0.008Ω
31.6k 34.8k
VOUT1
3.3V
5A
330µF
0.1µF
100k
8µH
1000pF
0.01Ω
10.5k
34.8k
VOUT2
8.5V
3A
TG1 TG2
BOOST1 BOOST2
SW1 SW2
BG1 BG2
SGND
PGND
SENSE1+SENSE2+
SENSE1–SENSE2–
VFB1 VFB2
ITH1 ITH2
VIN INTVCC
TRACK/SS1 TRACK/SS2
VIN
9V TO 60V
38901 TA01a
0.1µF 0.1µF
LTC3890-1
OUTPUT CURRENT (A)
0
EFFICIENCY (%)
POWER LOSS (mW)
70
100
10.10.010.0010.0001 10
38901 TA01b
50
40
30
20 1
10
100
1000
10000
0.1
10
60
80
90
VIN = 12V
VOUT = 3.3V
LTC3890-1
2
38901fa
ABSOLUTE MAXIMUM RATINGS
(Note 1)
ORDER INFORMATION
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LTC3890EGN-1#PBF LTC3890EGN-1#TRPBF LTC3890GN-1 28-Lead Plastic SSOP –40°C to 125°C
LTC3890IGN-1#PBF LTC3890IGN-1#TRPBF LTC3890GN-1 28-Lead Plastic SSOP –40°C to 125°C
LTC3890HGN-1#PBF LTC3890HGN-1#TRPBF LTC3890GN-1 28-Lead Plastic SSOP –40°C to 150°C
LTC3890MPGN-1#PBF LTC3890MPGN-1#TRPBF LTC3890GN-1 28-Lead Plastic SSOP –55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
Input Supply Voltage (VIN) ......................... –0.3V to 65V
Topside Driver Voltages
BOOST1, BOOST2 ..................................–0.3V to 71V
Switch Voltage (SW1, SW2) ......................... –5V to 65V
(BOOST1-SW1), (BOOST2-SW2) ................ –0.3V to 6V
RUN1, RUN2 ............................................... –0.3V to 8V
Maximum Current Sourced into Pin
from Source >8V ...................................................100µA
SENSE1+, SENSE2+, SENSE1–
SENSE2– Voltages ..................................... –0.3V to 28V
PLLIN/MODE Voltage ................................... –0.3V to 6V
FREQ Voltage ........................................ –0.3V to INTVCC
EXTVCC ..................................................... –0.3V to 14V
ITH1, ITH2, VFB1, VFB2 Voltages ................... –0.3V to 6V
PGOOD1 Voltage ......................................... –0.3V to 6V
TRACK/SS1, TRACK/SS2 Voltages ............. –0.3V to 6V
Operating Junction Temperature Range (Notes 2, 3)
LTC3890E-1, LTC3890I-1 ................... –40°C to 125°C
LTC3890H-1 ....................................... –40°C to 150°C
LTC3890MP-1 .................................... –55°C to 150°C
Storage Temperature Range .................. –65°C to 150°C
1
2
3
4
5
6
7
8
9
10
11
12
13
14
TOP VIEW
GN PACKAGE
28-LEAD PLASTIC SSOP
28
27
26
25
24
23
22
21
20
19
18
17
16
15
ITH1
VFB1
SENSE1+
SENSE1–
FREQ
PLLIN/MODE
SGND
RUN1
RUN2
SENSE2–
SENSE2+
VFB2
ITH2
TRACK/SS2
TRACK/SS1
PGOOD1
TG1
SW1
BOOST1
BG1
VIN
PGND
EXTVCC
INTVCC
BG2
BOOST2
SW2
TG2
TJMAX = 150°C, θJA = 90°C/W
PIN CONFIGURATION
LTC3890-1
3
38901fa
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating junction
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted. (Note 2)
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
VIN Input Supply Operating Voltage Range 4 60 V
VFB1,2 Regulated Feedback Voltage ITH1,2 Voltage = 1.2V (Note 4)
–40°C to 85°C, All Grades
LTC3890E-1, LTC3890I-1,
LTC3890H-1, LTC3890MP-1
l
l
0.792
0.788
0.786
0.800
0.800
0.800
0.808
0.812
0.812
V
V
V
IFB1,2 Feedback Current (Note 4) ±5 ±50 nA
VREFLNREG Reference Voltage Line Regulation VIN = 4.5V to 60V (Note 4) 0.002 0.02 %/V
VLOADREG Output Voltage Load Regulation (Note 4)
Measured in Servo Loop,
ΔITH Voltage = 1.2V to 0.7V
l0.01 0.1 %
(Note 4)
Measured in Servo Loop,
ΔITH Voltage = 1.2V to 2V
l–0.01 –0.1 %
gm1,2 Transconductance Amplifier gmITH1,2 = 1.2V, Sink/Source = 5µA (Note 4) 2 mmho
IQInput DC Supply Current (Note 5)
Pulse-Skipping or Forced Continuous
Mode (One Channel On)
RUN1 = 5V and RUN2 = 0V, VFB1 = 0.83V or
RUN1 = 0V and RUN2 = 5V, VFB2 = 0.83V
2mA
Pulse-Skipping or Forced Continuous
Mode (Both Channels On)
RUN1,2 = 5V, VFB1,2 = 0.83V (No Load) 2 mA
Sleep Mode (One Channel On) RUN1 = 5V and RUN2 = 0V, VFB1 = 0.83V or
RUN1 = 0V and RUN2 = 5V, VFB2 = 0.83V
50 75 µA
Sleep Mode (Both Channels On) RUN1,2 = 5V, VFB1,2 = 0.83V (No Load) 60 100 µA
Shutdown RUN1,2 = 0V 14 25 µA
UVLO Undervoltage Lockout INTVCC Ramping Up
INTVCC Ramping Down
l
l3.6
3.92
3.80
4.2
4.0
V
V
VOVL Feedback Overvoltage Protection Measured at VFB1,2, Relative to Regulated VFB1,2 71013 %
ISENSE+SENSE+ Pin Current Each Channel ±1 µA
ISENSE–SENSE– Pins Current Each Channel
VSENSE– < INTVCC – 0.5V
VSENSE– > INTVCC + 0.5V 700
±1 µA
µA
DFMAX Maximum Duty Factor In Dropout 98 99 %
ITRACK/SS1,2 Soft-Start Charge Current VTRACK1,2 = 0V 0.7 1.0 1.4 µA
VRUN1 On
VRUN2 On
RUN1 Pin On Threshold
RUN2 Pin On Threshold
VRUN1 Rising
VRUN2 Rising
l
l
1.15
1.20
1.21
1.25
1.27
1.30
V
V
VRUN1,2 Hyst RUN Pin Hysteresis 50 mV
VSENSE(MAX) Maximum Current Sense Threshold VFB1,2 = 0.7V, VSENSE1–,2– = 3.3V, ILIM = 0 l64 75 85 mV
Gate Driver
TG1,2 Pull-Up On-Resistance
Pull-Down On-Resistance
2.5
1.5
BG1,2 Pull-Up On-Resistance
Pull-Down On-Resistance
2.4
1.1
TG1,2 tr
TG1,2 tf
TG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
BG1,2 tr
BG1,2 tf
BG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
LTC3890-1
4
38901fa
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating junction
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted. (Note 2)
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
TG/BG t1D Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
CLOAD = 3300pF Each Driver 30 ns
BG/TG t1D Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
CLOAD = 3300pF Each Driver 30 ns
tON(MIN) Minimum On-Time (Note 7) 95 ns
INTVCC Linear Regulator
VINTVCCVIN Internal VCC Voltage 6V < VIN < 60V, VEXTVCC = 0V 4.85 5.1 5.35 V
VLDOVIN INTVCC Load Regulation ICC = 0mA to 50mA, VEXTVCC = 0V 0.7 1.1 %
VINTVCCEXT Internal VCC Voltage 6V < VEXTVCC < 13V 4.85 5.1 5.35 V
VLDOEXT INTVCC Load Regulation ICC = 0mA to 50mA, VEXTVCC = 8.5V 0.6 1.1 %
VEXTVCC EXTVCC Switchover Voltage EXTVCC Ramping Positive 4.5 4.7 4.9 V
VLDOHYS EXTVCC Hysteresis 250 mV
Oscillator and Phase-Locked Loop
f25k Programmable Frequency RFREQ = 25k, PLLIN/MODE = DC Voltage 105 kHz
f65k Programmable Frequency RFREQ = 65k, PLLIN/MODE = DC Voltage 375 440 505 kHz
f105k Programmable Frequency RFREQ = 105k, PLLIN/MODE = DC Voltage 835 kHz
fLOW Low Fixed Frequency VFREQ = 0V, PLLIN/MODE = DC Voltage 320 350 380 kHz
fHIGH High Fixed Frequency VFREQ = INTVCC, PLLIN/MODE = DC Voltage 485 535 585 kHz
fSYNC Synchronizable Frequency PLLIN/MODE = External Clock l75 850 kHz
PGOOD1 Output
VPGL PGOOD1 Voltage Low IPGOOD = 2mA 0.2 0.4 V
IPGOOD PGOOD1 Leakage Current VPGOOD = 5V ±1 µA
VPG PGOOD1 Trip Level VFB with Respect to Set Regulated Voltage
V
FB Ramping Negative
Hysteresis
–13 –10
2.5
–7 %
%
VFB with Respect to Set Regulated Voltage
V
FB Ramping Positive
Hysteresis
710
2.5
13 %
%
tPG Delay for Reporting a Fault 25 µs
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Ratings for extended periods may affect device reliability and
lifetime.
Note 2: The LTC3890-1 is tested under pulsed load condition such that
TJ ≈ TA.The LTC3890E-1 is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 125°C operating
junction temperature range are assured by design, characterization and
correlation with statistical process controls. The LTC3890I-1 is guaranteed
over the –40°C to 125°C operating junction temperature range, the
LTC3890H-1 is guaranteed over the –40°C to 150°C operating junction
temperature range and the LTC3890MP-1 is tested and guaranteed over
the –55°C to 150°C operating junction temperature range.
High junction temperatures degrade operating lifetimes; operating lifetime
is derated for junction temperatures greater than 125°C. Note that the
maximum ambient temperature consistent with these specifications is
determined by specific operating conditions in conjunction with board layout,
the rated package thermal impedance and other environmental factors.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
T
J = TA + (PD • 90°C/W)
Note 4: The LTC3890-1 is tested in a feedback loop that servos VITH1,2 to
a specified voltage and measures the resultant VFB. The specification at
85°C is not tested in production and is assured by design, characterization
and correlation to production testing at other temperatures (125°C for the
LTC3890E-1/LTC3890I-1, 150°C for the LTC3890H-1/LTC3890MP-1). For
the LTC3890MP-1, the specification at –40°C is not tested in production
and is assured by design, characterization and correlation to production
testing at –55°C.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See the Applications information section.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 7: The minimum on-time condition is specified for an inductor peak-
to-peak ripple current ≥ of IMAX (See Minimum On-Time Considerations in
the Applications Information section).
LTC3890-1
5
38901fa
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency and Power Loss
vs Output Current Efficiency vs Output Current
Load Step
Burst Mode Operation
Load Step
Forced Continuous Mode
Load Step
Pulse-Skipping Mode
Inductor Current at Light Load Soft Start-Up Tracking Start-Up
Efficiency vs Input Voltage
VIN = 12V
VOUT = 3.3V
FIGURE 13 CIRCUIT
50µs/DIV 38901 G04
VOUT
100mV/DIV
AC-
COUPLED
IL
2A/DIV
VIN = 12V
VOUT = 3.3V
FIGURE 13 CIRCUIT
50µs/DIV 38901 G05
VOUT
100mV/DIV
AC-
COUPLED
IL
2A/DIV
VIN = 12V
VOUT = 3.3V
FIGURE 13 CIRCUIT
50µs/DIV 38901 G06
VOUT
100mV/DIV
AC-
COUPLED
IL
2A/DIV
VIN = 12V
VOUT = 3.3V
ILOAD = 200µA
5µs/DIV 38901 G07
FORCED
CONTINUOUS
MODE
PULSE-SKIPPING
MODE
Burst Mode
OPERATION
1A/DIV
FIGURE 13 CIRCUIT
2ms/DIV 38901 G08
VOUT2
2V/DIV
VOUT1
2V/DIV
FIGURE 13 CIRCUIT
2ms/DIV 38901 G09
VOUT2
2V/DIV
VOUT1
2V/DIV
OUTPUT CURRENT (A)
0
EFFICIENCY (%)
70
100
10.10.010.0010.0001 10
38901 G02
50
40
30
20
10
60
80
90 VOUT = 8.5V
VOUT = 3.3V
VIN = 12V
Burst Mode OPERATION
FIGURE 13 CIRCUIT
INPUT VOLTAGE (V)
80
EFFICIENCY (%)
94
100
20 30 35 40 45 50 55 6015105025
38901 G03
90
88
86
84
82
92
96
98
VOUT1 = 3.3V
VOUT2 = 8.5V
ILOAD = 2A
FIGURE 13 CIRCUIT
OUTPUT CURRENT (A)
FIGURE 13 CIRCUIT
0
EFFICIENCY (%)
POWER LOSS (mW)
70
100
10.10.010.0010.0001 10
38901 G01
50
40
30
20 1
10
100
1000
10000
0.1
10
60
80
90
VIN = 12V
VOUT = 3.3V BURST EFFICIENCY
CCM LOSS
PULSE-SKIPPING
LOSS
BURST LOSS
CCM EFFICIENCY
PULSE-SKIPPING
EFFICIENCY
LTC3890-1
6
38901fa
TYPICAL PERFORMANCE CHARACTERISTICS
Total Input Supply Current
vs Input Voltage
EXTVCC Switchover and INTVCC
Voltages vs Temperature INTVCC Line Regulation
Maximum Current Sense Voltage
vs ITH Voltage SENSE– Pin Input Bias Current
Maximum Current Sense
Threshold vs Duty Cycle
Foldback Current Limit Quiescent Current vs Temperature INTVCC vs Load Current
DUTY CYCLE (%)
0
MAXIMUM CURRENT SENSE VOLTAGE (mV)
75
85
80
80
38901 G15
65
70
60 20 40 50 100
90
60
10 30 90
70
INPUT VOLTAGE (V)
0
SUPPLY CURRENT (µA)
200
300
20 30 35 40 45 50 55 60 651510525
38901 G10
150
100
50
250
NO LOAD
300µA LOAD
VOUT = 3.3V
FIGURE 13 CIRCUIT
VSENSE COMMON MODE VOLTAGE (V)
0
SENSE– CURRENT (µA)
20
38901 G14
400
300
10 15525
800
700
600
500
200
100
0
–100
TEMPERATURE (°C)
–75
QUIESCENT CURRENT (µA)
75
50250
38901 G17
60
50
–50 –25 75
45
30
40
35
80
70
65
55
100 125 150
VIN = 12V
LOAD CURRENT (mA)
0
INTVCC VOLTAGE (V)
5.25
40
38901 G18
4.50
20 60
4.00
5.50
5.00
4.75
4.25
80 100
EXTVCC = 5V
EXTVCC = 8.5V
EXTVCC = 0V
VIN = 12V
VITH (V)
0
CURRENT SENSE THESHOLD (mV)
40
60
80
0.6 1.0
38901 G13
20
0
0.2 0.4 0.8 1.2 1.4
–20
–40
Burst Mode
OPERATION
PULSE-SKIPPING MODE
5% DUTY CYCLE
FORCED CONTINUOUS MODE
INPUT VOLTAGE (V)
3.0
INTVCC VOLTAGE (V)
4.0
4.5
5.5
20 30 35 40 45 50 55 60 65151005 25
38901 G12
3.5
5.0
ILOAD = 10mA
TEMPERATURE (°C)
–75
4.0
EXTVCC AND INTVCC VOLTAGE (V)
4.2
4.6
4.8
5.0
6.0
5.4
–25 25 50 75 100
38901 G11
4.4
5.6
5.8
5.2
–50 0125 150
INTVCC
EXTVCC RISING
EXTVCC FALLING
FEEDBACK VOLTAGE (MV)
0
MAXIMUM CURRENT SENSE VOLTAGE (mV)
40
60
800
38901 G16
20
0200 400 500
80
30
50
10
70
600
100 300 700
LTC3890-1
7
38901fa
TYPICAL PERFORMANCE CHARACTERISTICS
TRACK/SS Pull-Up Current
vs Temperature
Shutdown (RUN) Threshold
vs Temperature
Regulated Feedback Voltage
vs Temperature
SENSE– Pin Total Input Bias Current
vs Temperature
Shutdown Current
vs Input Voltage
Oscillator Frequency
vs Temperature
Undervoltage Lockout Threshold
vs Temperature
Oscillator Frequency
vs Input Voltage Shutdown Current vs Temperature
INPUT VOLTAGE (V)
344
OSCILLATOR FREQUENCY (kHz)
348
350
356
20 30 35 40 45 50 55 60 651510525
38901 G26
346
352
354
FREQ = GND
TEMPERATURE (°C)
–75
RUN PIN VOLTAGE (V)
1.30
1.35
1.40
0 100
38901 G20
1.25
1.20
–50 –25 5025 125 150
1.15
1.00
1.05
1.10
RUN1 FALLING
RUN2 RISING
RUN1 RISING
RUN2 FALLING
75
TEMPERATURE (°C)
–75
REGULATED FEEDBACK VOLTAGE (mV)
806
02550
38901 G21
800
796
–50 –25 75
794
792
808
804
802
798
100 125 150
INPUT VOLTAGE (V)
0
SHUTDOWN CURRENT (µA)
10
15
30
25
20 30 35 40 45 50 55 60 651510525
38901 G23
5
20
TEMPERATURE (°C)
8
SHUTDOWN CURRENT (µA)
12
14
22
20
25 50 75 100 125 150–25–75 –50 0
38901 G27
10
16
18
VIN = 12V
TEMPERATURE (°C)
–75
INTVCC VOLTAGE (V)
3.7
3.8
3.9
4.1
–25 75 100
38901 G25
3.6
4.2
4.0
–50 02550 125 150
FALLING
RISING
TEMPERATURE (°C)
–75
FREQUENCY (kHz)
500
550
600
0 25 50 100
38901 G24
450
400
–50 –25 75 125 150
350
300
FREQ = GND
FREQ = INTVCC
TEMPERATURE (°C)
–75
SENSE– CURRENT (µA)
50250
38901 G22
400
300
–25–50 75
800
700
600
500
200
100
0
–100 125100 150
VOUT > INTVCC + 0.5V
VOUT < INTVCC – 0.5V
TEMPERATURE (°C)
–75
0.90
TRACK/SS CURRENT (µA)
0.95
1.00
1.05
1.10
–50 –25 0 755025
38901 G19
100 125 150
LTC3890-1
8
38901fa
PIN FUNCTIONS
ITH1, ITH2 (Pin 1, Pin 13): Error Amplifier Outputs and
Switching Regulator Compensation Points. Each associ-
ated channel’s current comparator trip point increases
with this control voltage.
VFB1, VFB2 (Pin 2, Pin 12): Receives the remotely sensed
feedback voltage for each controller from an external
resistive divider across the output.
SENSE1+, SENSE2+ (Pin 3, Pin 11): The (+) input to the
differential current comparators are normally connected
to DCR sensing networks or current sensing resistors.
The ITH pin voltage and controlled offsets between the
SENSE– and SENSE+ pins in conjunction with RSENSE set
the current trip threshold.
SENSE1–, SENSE2– (Pin 4, Pin 10): The (–) Input to
the Differential Current Comparators. When greater than
INTVCC – 0.5V, the SENSE– pin supplies current to the
current comparator.
FREQ (Pin 5): The Frequency Control Pin for the Internal
VCO. Connecting the pin to GND forces the VCO to a fixed
low frequency of 350kHz. Connecting the pin to INTVCC
forces the VCO to a fixed high frequency of 535kHz.
Other frequencies between 50kHz and 900kHz can be
programmed using a resistor between FREQ and GND.
An internal 20µA pull-up current develops the voltage to
be used by the VCO to control the frequency.
PLLIN/MODE (Pin 6): External Synchronization Input to
Phase Detector and Forced Continuous Mode Input. When
an external clock is applied to this pin, the phase-locked
loop will force the rising TG1 signal to be synchronized
with the rising edge of the external clock. When not syn-
chronizing to an external clock, this input, which acts on
both controllers, determines how the LTC3890-1 operates
at light loads. Pulling this pin to ground selects Burst Mode
operation. An internal 100k resistor to ground also invokes
Burst Mode Operation when the pin is floated. Tying this
pin to INTVCC forces continuous inductor current operation.
Tying this pin to a voltage greater than 1.2V and less than
INTVCC – 1.3V selects pulse-skipping operation.
SGND (Pin 7): Small-signal ground common to both
controllers, must be routed separately from high current
grounds to the common (–) terminals of the CIN capacitors.
RUN1, RUN2 (Pin 8, Pin 9): Digital Run Control Inputs
for Each Controller. Forcing RUN1 below 1.16V or RUN2
below 1.20V shuts down that controller. Forcing both of
these pins below 0.7V shuts down the entire LTC3890-1,
reducing quiescent current to approximately 14µA.
INTVCC (Pin 19): Output of the Internal Linear Low Dropout
Regulator. The driver and control circuits are powered from
this voltage source. Must be decoupled to power ground
with a minimum of 4.7µF ceramic or other low ESR ca-
pacitor. Do not use the INTVCC pin for any other purpose.
EXTVCC (Pin 20): External Power Input to an Internal LDO
Connected to INTVCC. This LDO supplies INTVCC power,
bypassing the internal LDO powered from VIN whenever
EXTVCC is higher than 4.7V. See EXTVCC Connection in the
Applications Information section. Do not float or exceed
14V on this pin.
PGND (Pin 21): Driver Power Ground. Connects to the
sources of bottom (synchronous) N-channel MOSFETs
and the (–) terminal(s) of CIN.
VIN (Pin 22): Main Supply Pin. A bypass capacitor should
be tied between this pin and the signal ground pin.
LTC3890-1
9
38901fa
BG1, BG2 (Pin 23, Pin 18): High Current Gate Drives
for Bottom (Synchronous) N-Channel MOSFETs. Voltage
swing at these pins is from ground to INTVCC.
BOOST1, BOOST2 (Pin 24, Pin 17): Bootstrapped Supplies
to the Topside Floating Drivers. Capacitors are connected
between the BOOST and SW pins and Schottky diodes are
tied between the BOOST and INTVCC pins. Voltage swing
at the BOOST pins is from INTVCC to (VIN + INTVCC).
SW1, SW2 (Pin 25, Pin 16): Switch Node Connections
to Inductors.
TG1, TG2 (Pin 26, Pin 15): High Current Gate Drives for
Top N-Channel MOSFETs. These are the outputs of float-
ing drivers with a voltage swing equal to INTVCC – 0.5V
superimposed on the switch node voltage SW.
PIN FUNCTIONS
PGOOD1 (Pin 27): Open-Drain Logic Output. PGOOD1 is
pulled to ground when the voltage on the VFB1 pin is not
within ±10% of its set point.
TRACK/SS1, TRACK/SS2 (Pin 28, Pin 14): External Track-
ing and Soft-Start Input. The LTC3890-1 regulates the
VFB1,2 voltage to the smaller of 0.8V or the voltage on the
TRACK/SS1,2 pin. An internal 1µA pull-up current source
is connected to this pin. A capacitor to ground at this
pin sets the ramp time to final regulated output voltage.
Alternatively, a resistor divider on another voltage supply
connected to this pin allows the LTC3890-1 output to track
the other supply during start-up.
LTC3890-1
10
38901fa
FUNCTIONAL DIAGRAM
SW
TOP
BOOST
TG CB
CIN
D
DB
PGND
BOT BG
INTVCC
INTVCC
VIN
COUT
VOUT
38901 FD
RSENSE
DROP
OUT
DET BOT
TOP ON
S
R
Q
Q
SHDN
SLEEP
0.425V
ICMP
2.7V
0.65V
IR
3mV
SLOPE COMP
DUPLICATE FOR SECOND
CONTROLLER CHANNEL
SENSE+
SENSE–
PGOOD1
VFB1
0.88V
0.72V
L
+
–
+
–
FREQ
+
–
+
–
+
–
+
–
+
–
+
–
SWITCH
LOGIC
VFB
RA
CC
RC
CC2
RB
0.80V
TRACK/SS
0.88V
RUN
ITH
TRACK/SS
+
–
CSS
1µA
SHDN
FOLDBACK
SHDN
RST
2(VFB)
PLLIN/MODE
20µA
VCO
LDO
EN
INTVCC
5.1V
SYNC
DET
100k
CLK2
CLP
CLK1
VIN
EXTVCC
LDO
PFD
EN
4.7V
5.1V
+
–
SGND
EA
OV
7µA (RUN1)
0.5µA (RUN2)
11V
LTC3890-1
11
38901fa
OPERATION
(Refer to the Functional Diagram)
Main Control Loop
The LTC3890-1 uses a constant frequency, current mode
step-down architecture with the two controller channels
operating 180 degrees out-of-phase. During normal op-
eration, each external top MOSFET is turned on when the
clock for that channel sets the RS latch, and is turned off
when the main current comparator, ICMP, resets the RS
latch. The peak inductor current at which ICMP trips and
resets the latch is controlled by the voltage on the ITH pin,
which is the output of the error amplifier, EA. The error
amplifier compares the output voltage feedback signal at
the VFB pin, (which is generated with an external resistor
divider connected across the output voltage, VOUT
, to
ground) to the internal 0.800V reference voltage. When the
load current increases, it causes a slight decrease in VFB
relative to the reference, which causes the EA to increase
the ITH voltage until the average inductor current matches
the new load current.
After the top MOSFET is turned off each cycle, the bottom
MOSFET is turned on until either the inductor current starts
to reverse, as indicated by the current comparator IR, or
the beginning of the next clock cycle.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin.
When the EXTVCC pin is tied to a voltage less than 4.7V,
the VIN LDO (low dropout linear regulator) supplies 5.1V
from VIN to INTVCC. If EXTVCC is taken above 4.7V, the VIN
LDO is turned off and an EXTVCC LDO is turned on. Once
enabled, the EXTVCC LDO supplies 5.1V from EXTVCC to
INTVCC. Using the EXTVCC pin allows the INTVCC power
to be derived from a high efficiency external source such
as one of the LTC3890-1 switching regulator outputs.
Each top MOSFET driver is biased from the floating boot-
strap capacitor CB, which normally recharges during each
cycle through an external diode when the top MOSFET
turns off. If the input voltage, VIN, decreases to a voltage
close to VOUT
, the loop may enter dropout and attempt
to turn on the top MOSFET continuously. The dropout
detector detects this and forces the top MOSFET off for
about one-twelfth of the clock period every tenth cycle to
allow CB to recharge.
Shutdown and Start-Up (RUN1, RUN2 and
TRACK/ SS1, TRACK/SS2 Pins)
The two channels of the LTC3890-1 can be independently
shut down using the RUN1 and RUN2 pins. Pulling either of
these pins below 1.15V shuts down the main control loop
for that controller. Pulling both pins below 0.7V disables
both controllers and most internal circuits, including the
INTVCC LDOs. In this state, the LTC3890-1 draws only
14µA of quiescent current.
Releasing either RUN pin allows a small internal current to
pull up the pin to enable that controller. The RUN1 pin has a
7µA pull-up current while the RUN2 pin has a smaller 0.5µA.
The 7µA current on RUN1 is designed to be large enough
so that the RUN1 pin can be safely floated (to always en-
able the controller) without worry of condensation or other
small board leakage pulling the pin down. This is ideal for
always-on applications where one or both controllers are
enabled continuously and never shut down.
The RUN pin may be externally pulled up or driven directly
by logic. When driving the RUN pin with a low impedance
source, do not exceed the absolute maximum rating of
8V. The RUN pin has an internal 11V voltage clamp that
allows the RUN pin to be connected through a resistor to a
higher voltage (for example, VIN), so long as the maximum
current into the RUN pin does not exceed 100µA.
The start-up of each controller’s output voltage VOUT is
controlled by the voltage on the TRACK/SS pin for that
channel. When the voltage on the TRACK/SS pin is less
than the 0.8V internal reference, the LTC3890-1 regulates
the VFB voltage to the TRACK/SS pin voltage instead of the
0.8V reference. This allows the TRACK/SS pin to be used
to program a soft-start by connecting an external capacitor
from the TRACK/SS pin to SGND. An internal 1µA pull-up
current charges this capacitor creating a voltage ramp on
the TRACK/SS pin. As the TRACK/SS voltage rises linearly
from 0V to 0.8V (and beyond up to 5V), the output voltage
VOUT rises smoothly from zero to its final value.
Alternatively the TRACK/SS pin can be used to cause the
start-up of VOUT to track that of another supply. Typically,
this requires connecting to the TRACK/SS pin an external
resistor divider from the other supply to ground (see the
Applications Information section).
LTC3890-1
12
38901fa
OPERATION
(Refer to the Functional Diagram)
Light Load Current Operation (Burst Mode Operation,
Pulse-Skipping, or Forced Continuous Mode)
(PLLIN/MODE Pin)
The LTC3890-1 can be enabled to enter high efficiency
Burst Mode operation, constant frequency pulse-skipping
mode, or forced continuous conduction mode at low load
currents. To select Burst Mode operation, tie the PLLIN/
MODE pin to a DC voltage below 0.8V (e.g., SGND). To
select forced continuous operation, tie the PLLIN/MODE
pin to INTVCC. To select pulse-skipping mode, tie the
PLLIN/MODE pin to a DC voltage greater than 1.2V and
less than INTVCC – 1.3V.
When a controller is enabled for Burst Mode operation, the
minimum peak current in the inductor is set to approxi-
mately 25% of the maximum sense voltage even though
the voltage on the ITH pin indicates a lower value. If the
average inductor current is higher than the load current,
the error amplifier, EA, will decrease the voltage on the
ITH pin. When the ITH voltage drops below 0.425V, the
internal sleep signal goes high (enabling sleep mode)
and both external MOSFETs are turned off. The ITH pin is
then disconnected from the output of the EA and parked
at 0.450V.
In sleep mode, much of the internal circuitry is turned off,
reducing the quiescent current that the LTC3890-1 draws.
If one channel is shut down and the other channel is in
sleep mode, the LTC3890-1 draws only 50µA of quiescent
current. If both channels are in sleep mode, the LTC3890-1
draws only 60µA of quiescent current. In sleep mode,
the load current is supplied by the output capacitor. As
the output voltage decreases, the EA’s output begins to
rise. When the output voltage drops enough, the ITH pin
is reconnected to the output of the EA, the sleep signal
goes low, and the controller resumes normal operation
by turning on the top external MOSFET on the next cycle
of the internal oscillator.
When a controller is enabled for Burst Mode operation, the
inductor current is not allowed to reverse. The reverse cur-
rent comparator, IR, turns off the bottom external MOSFET
just before the inductor current reaches zero, preventing
it from reversing and going negative. Thus, the controller
operates in discontinuous operation.
In forced continuous operation or clocked by an external
clock source to use the phase-locked loop (see Frequency
Selection and Phase-Locked Loop section), the inductor
current is allowed to reverse at light loads or under large
transient conditions. The peak inductor current is deter-
mined by the voltage on the ITH pin, just as in normal
operation. In this mode, the efficiency at light loads is
lower than in Burst Mode operation. However, continuous
operation has the advantage of lower output voltage ripple
and less interference to audio circuitry. In forced continu-
ous mode, the output ripple is independent of load current.
When the PLLIN/MODE pin is connected for pulse-skipping
mode, the LTC3890-1 operates in PWM pulse-skipping
mode at light loads. In this mode, constant frequency
operation is maintained down to approximately 1% of
designed maximum output current. At very light loads, the
current comparator, ICMP, may remain tripped for several
cycles and force the external top MOSFET to stay off for
the same number of cycles (i.e., skipping pulses). The
inductor current is not allowed to reverse (discontinuous
operation). This mode, like forced continuous operation,
exhibits low output ripple as well as low audio noise and
reduced RF interference as compared to Burst Mode
operation. It provides higher low current efficiency than
forced continuous mode, but not nearly as high as Burst
Mode operation.
Frequency Selection and Phase-Locked Loop
(FREQ and PLLIN/MODE Pins)
The selection of switching frequency is a trade-off between
efficiency and component size. Low frequency opera-
tion increases efficiency by reducing MOSFET switching
losses, but requires larger inductance and/or capacitance
to maintain low output ripple voltage.
The switching frequency of the LTC3890-1’s controllers
can be selected using the FREQ pin.
LTC3890-1
13
38901fa
OPERATION
(Refer to the Functional Diagram)
If the PLLIN/MODE pin is not being driven by an external
clock source, the FREQ pin can be tied to SGND, tied to
INTVCC or programmed through an external resistor. Tying
FREQ to SGND selects 350kHz while tying FREQ to INTVCC
selects 535kHz. Placing a resistor between FREQ and
SGND allows the frequency to be programmed between
50kHz and 900kHz, as shown in Figure 10.
A phase-locked loop (PLL) is available on the LTC3890-1
to synchronize the internal oscillator to an external clock
source that is connected to the PLLIN/MODE pin. The
LTC3890-1’s phase detector adjusts the voltage (through
an internal lowpass filter) of the VCO input to align the
turn-on of controller 1’s external top MOSFET to the ris-
ing edge of the synchronizing signal. Thus, the turn-on
of controller 2’s external top MOSFET is 180 degrees out
of phase to the rising edge of the external clock source.
The VCO input voltage is prebiased to the operating
frequency set by the FREQ pin before the external clock
is applied. A resistor connected between the FREQ pin
and SGND can prebias VCO’s input voltage to the desired
frequency. If prebiased near the external clock frequency,
the PLL loop only needs to make slight changes to the
VCO input in order to synchronize the rising edge of the
external clock’s to the rising edge of TG1. The ability to
prebias the loop filter allows the PLL to lock-in rapidly
without deviating far from the desired frequency.
The typical capture range of the phase-locked loop is
from approximately 55kHz to 1MHz, with a guarantee
to be between 75kHz and 850kHz. In other words, the
LTC3890-1’s PLL is guaranteed to lock to an external clock
source whose frequency is between 75kHz and 850kHz.
The typical input clock thresholds on the PLLIN/MODE
pin are 1.6V (rising) and 1.2V (falling).
Output Overvoltage Protection
An overvoltage comparator guards against transient over-
shoots as well as other more serious conditions that may
overvoltage the output. When the VFB pin rises by more
than 10% above its regulation point of 0.800V, the top
MOSFET is turned off and the bottom MOSFET is turned
on until the overvoltage condition is cleared.
Power Good (PGOOD1 Pin)
The PGOOD1 pin is connected to an open drain of an internal
N-channel MOSFET. The MOSFET turns on and pulls the
PGOOD1 pin low when the corresponding VFB1 pin volt-
age is not within ±10% of the 0.8V reference voltage. The
PGOOD1 pin is also pulled low when the corresponding
RUN1 pin is low (shut down). When the VFB1 pin voltage
is within the ±10% requirement, the MOSFET is turned
off and the pin is allowed to be pulled up by an external
resistor to a source no greater than 6V.
Foldback Current
When the output voltage falls to less than 70% of its
nominal level, foldback current limiting is activated, pro-
gressively lowering the peak current limit in proportion to
the severity of the overcurrent or short-circuit condition.
Foldback current limiting is disabled during the soft-start
interval (as long as the VFB voltage is keeping up with the
TRACK/SS voltage).
Theory and Benefits of 2-Phase Operation
Why the need for 2-phase operation? Up until the 2-phase
family, constant-frequency dual switching regulators
operated both channels in phase (i.e., single phase
operation). This means that both switches turned on at
the same time, causing current pulses of up to twice the
amplitude of those for one regulator to be drawn from the
input capacitor and battery. These large amplitude current
pulses increased the total RMS current flowing from the
input capacitor, requiring the use of more expensive input
capacitors and increasing both EMI and losses in the input
capacitor and battery.
With 2-phase operation, the two channels of the dual
switching regulator are operated 180 degrees out-of-phase.
This effectively interleaves the current pulses drawn by the
switches, greatly reducing the overlap time where they add
LTC3890-1
14
38901fa
OPERATION
(Refer to the Functional Diagram)
Figure 1. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual Switching Regulators
Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple with the 2-Phase Regulator Allows
Less Expensive Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency
Figure 2. RMS Input Current Comparison
together. The result is a significant reduction in total RMS
input current, which in turn allows less expensive input
capacitors to be used, reduces shielding requirements for
EMI and improves real world operating efficiency.
Figure 1 compares the input waveforms for a representative
single-phase dual switching regulator to the LTC3890-1
2-phase dual switching regulator. An actual measure-
ment of the RMS input current under these conditions
shows that 2-phase operation dropped the input current
from 2.53ARMS to 1.55ARMS. While this is an impressive
reduction in itself, remember that the power losses are
proportional to IRMS2, meaning that the actual power wasted
is reduced by a factor of 2.66. The reduced input ripple
voltage also means less power is lost in the input power
path, which could include batteries, switches, trace/con-
nector resistances and protection circuitry. Improvements
in both conducted and radiated EMI also directly accrue
as a result of the reduced RMS input current and voltage.
Of course, the improvement afforded by 2-phase opera-
tion is a function of the dual switching regulator’s relative
duty cycles which, in turn, are dependent upon the input
voltage VIN (Duty Cycle = VOUT/VIN). Figure 2 shows how
the RMS input current varies for single phase and 2-phase
operation for 3.3V and 5V regulators over a wide input
voltage range.
It can readily be seen that the advantages of 2-phase op-
eration are not just limited to a narrow operating range,
for most applications is that 2-phase operation will reduce
the input capacitor requirement to that for just one chan-
nel operating at maximum current and 50% duty cycle.
IIN(MEAS) = 2.53ARMS IIN(MEAS) = 1.55ARMS
38901 F01
5V SWITCH
20V/DIV
3.3V SWITCH
20V/DIV
INPUT CURRENT
5A/DIV
INPUT VOLTAGE
500mV/DIV
INPUT VOLTAGE (V)
0
INPUT RMS CURRENT (A)
3.0
2.5
2.0
1.5
1.0
0.5
010 20 30 40
38901 F02
SINGLE PHASE
DUAL CONTROLLER
2-PHASE
DUAL CONTROLLER
VO1 = 5V/3A
VO2 = 3.3V/3A
LTC3890-1
15
38901fa
APPLICATIONS INFORMATION
Figure 3. Sense Lines Placement with Inductor or Sense Resistor
(4a) Using a Resistor to Sense Current
(4b) Using the Inductor DCR to Sense Current
Figure 4. Current Sensing Methods
The Typical Application on the first page is a basic
LTC3890-1 application circuit. LTC3890-1 can be configured
to use either DCR (inductor resistance) sensing or low
value resistor sensing. The choice between the two current
sensing schemes is largely a design trade-off between
cost, power consumption, and accuracy. DCR sensing
is becoming popular because it saves expensive current
sensing resistors and is more power efficient, especially
in high current applications. However, current sensing
resistors provide the most accurate current limits for the
controller. Other external component selection is driven
by the load requirement, and begins with the selection of
RSENSE (if RSENSE is used) and inductor value. Next, the
power MOSFETs and Schottky diodes are selected. Finally,
input and output capacitors are selected.
SENSE+ and SENSE– Pins
The SENSE+ and SENSE– pins are the inputs to the current
comparators. The common mode voltage range on these
pins is 0V to 28V (abs max), enabling the LTC3890-1 to
regulate output voltages up to a nominal 24V (allowing
margin for tolerances and transients).
The SENSE+ pin is high impedance over the full common
mode range, drawing at most ±1µA. This high impedance
allows the current comparators to be used in inductor
DCR sensing.
The impedance of the SENSE– pin changes depending on
the common mode voltage. When SENSE– is less than
INTVCC – 0.5V, a small current of less than 1µA flows out
of the pin. When SENSE– is above INTVCC + 0.5V, a higher
current (~700µA) flows into the pin. Between INTVCC –
0.5V and INTVCC + 0.5V, the current transitions from the
smaller current to the higher current.
Filter components mutual to the sense lines should be
placed close to the LTC3890-1, and the sense lines should
run close together to a Kelvin connection underneath the
current sense element (shown in Figure 3). Sensing cur-
rent elsewhere can effectively add parasitic inductance
and capacitance to the current sense element, degrading
the information at the sense terminals and making the
programmed current limit unpredictable. If inductor DCR
sensing is used (Figure 4b), sense resistor R1 should be
placed close to the switching node, to prevent noise from
coupling into sensitive small-signal nodes.
COUT
TO SENSE FILTER,
NEXT TO THE CONTROLLER
INDUCTOR OR RSENSE
38901 F03
VIN VIN
INTVCC
BOOST
TG
SW
BG
PLACE CAPACITOR NEAR
SENSE PINS
SENSE+
SENSE–
SGND
LTC3890-1
VOUT
RSENSE
38901 F04a
C1*
R1*
*R1 AND C1 ARE OPTIONAL.
VIN VIN
INTVCC
BOOST
TG
SW
BG
*PLACE C1 NEAR
SENSE PINS
INDUCTOR
DCRL
SENSE+
SENSE–
SGND
LTC3890-1
VOUT
38901 F04b
R1
R2C1*
(R1||R2) t C1 = L
DCR RSENSE(EQ) = DCR R2
R1 + R2
LTC3890-1
16
38901fa
APPLICATIONS INFORMATION
Low Value Resistor Current Sensing
A typical sensing circuit using a discrete resistor is shown
in Figure 4a. RSENSE is chosen based on the required
output current.
The current comparator has a maximum threshold
VSENSE(MAX). The current comparator threshold voltage
sets the peak of the inductor current, yielding a maximum
average output current, IMAX, equal to the peak value less
half the peak-to-peak ripple current, ΔIL. To calculate the
sense resistor value, use the equation:
RSENSE =VSENSE(MAX)
IMAX +ΔIL
2
To ensure that the application will deliver full load current
over the full operating temperature range, choose the
minimum value for the Maximum Current Sense Threshold
(VSENSE(MAX)).
When using the controller in very low dropout conditions,
the maximum output current level will be reduced due to
the internal compensation required to meet stability cri-
terion for buck regulators operating at greater than 50%
duty factor. A curve is provided in the Typical Performance
Characteristics section to estimate this reduction in peak
inductor current depending upon the operating duty factor.
Inductor DCR Sensing
For applications requiring the highest possible efficiency
at high load currents, the LTC3890-1 is capable of sensing
the voltage drop across the inductor DCR, as shown in
Figure 4b. The DCR of the inductor represents the small
amount of DC resistance of the copper wire, which can be
less than 1m for today’s low value, high current inductors.
In a high current application requiring such an inductor,
power loss through a sense resistor would cost several
points of efficiency compared to inductor DCR sensing.
If the external (R1||R2) • C1 time constant is chosen to be
exactly equal to the L/DCR time constant, the voltage drop
across the external capacitor is equal to the drop across
the inductor DCR multiplied by R2/(R1 + R2). R2 scales the
voltage across the sense terminals for applications where
the DCR is greater than the target sense resistor value.
To properly dimension the external filter components, the
DCR of the inductor must be known. It can be measured
using a good RLC meter, but the DCR tolerance is not
always the same and varies with temperature; consult
the manufacturers’ data sheets for detailed information.
Using the inductor ripple current value from the Inductor
Value Calculation section, the target sense resistor value is:
RSENSE(EQUIV) =VSENSE(MAX)
IMAX +ΔIL
2
To ensure that the application will deliver full load current
over the full operating temperature range, choose the
minimum value for the Maximum Current Sense Threshold
(VSENSE(MAX)).
Next, determine the DCR of the inductor. When provided,
use the manufacturer’s maximum value, usually given at
20°C. Increase this value to account for the temperature
coefficient of copper resistance, which is approximately
0.4%/°C. A conservative value for TL(MAX) is 100°C.
To scale the maximum inductor DCR to the desired sense
resistor value (RD), use the divider ratio:
RD=RSENSE(EQUIV)
DCRMAX atTL(MAX)
C1 is usually selected to be in the range of 0.1µF to 0.47µF.
This forces R1|| R2 to around 2k, reducing error that might
have been caused by the SENSE+ pin’s ±1µA current.
The equivalent resistance R1|| R2 is scaled to the room
temperature inductance and maximum DCR:
R1|| R2 =L
DCR at 20°C
()
•C1
The sense resistor values are:
R1=R1|| R2
RD
; R2 =R1• RD
1–RD
LTC3890-1
17
38901fa
APPLICATIONS INFORMATION
The maximum power loss in R1 is related to duty cycle,
and will occur in continuous mode at the maximum input
voltage:
PLOSS R1=VIN(MAX) –V
OUT
()
•V
OUT
R1
Ensure that R1 has a power rating higher than this value.
If high efficiency is necessary at light loads, consider this
power loss when deciding whether to use DCR sensing or
sense resistors. Light load power loss can be modestly
higher with a DCR network than with a sense resistor, due
to the extra switching losses incurred through R1. However,
DCR sensing eliminates a sense resistor, reduces conduc-
tion losses and provides higher efficiency at heavy loads.
Peak efficiency is about the same with either method.
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
MOSFET switching and gate charge losses. In addition to
this basic trade-off, the effect of inductor value on ripple
current and low current operation must also be considered.
The inductor value has a direct effect on ripple current. The
inductor ripple current, ΔIL, decreases with higher induc-
tance or higher frequency and increases with higher VIN:
ΔIL=1
f
()
L
()
VOUT 1– VOUT
VIN
⎛
⎝
⎜⎞
⎠
⎟
Accepting larger values of ΔIL allows the use of low in-
ductances, but results in higher output voltage ripple and
greater core losses. A reasonable starting point for setting
ripple current is ΔIL =0.3(IMAX). The maximum ΔIL occurs
at the maximum input voltage.
The inductor value also has secondary effects. The tran-
sition to Burst Mode operation begins when the average
inductor current required results in a peak current below
25% of the current limit determined by RSENSE. Lower
inductor values (higher ΔIL) will cause this to occur at
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease.
Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite or molypermalloy
cores. Actual core loss is independent of core size for a
fixed inductor value, but it is very dependent on inductance
value selected. As inductance increases, core losses go
down. Unfortunately, increased inductance requires more
turns of wire and therefore copper losses will increase.
Ferrite designs have very low core loss and are preferred
for high switching frequencies, so design goals can con-
centrate on copper loss and preventing saturation. Ferrite
core material saturates hard, which means that induc-
tance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Power MOSFET and Schottky Diode
(Optional) Selection
Two external power MOSFETs must be selected for each
controller in the LTC3890-1: one N-channel MOSFET for
the top (main) switch, and one N-channel MOSFET for the
bottom (synchronous) switch.
The peak-to-peak drive levels are set by the INTVCC voltage.
This voltage is typically 5.1V during start-up (see EXTVCC
Pin Connection). Consequently, logic-level threshold
MOSFETs must be used in most applications. Pay close
attention to the BVDSS specification for the MOSFETs as well.
Selection criteria for the power MOSFETs include the
on-resistance, RDS(ON), Miller capacitance, CMILLER, input
voltage and maximum output current. Miller capacitance,
CMILLER, can be approximated from the gate charge curve
LTC3890-1
18
38901fa
APPLICATIONS INFORMATION
usually provided on the MOSFET manufacturers’ data
sheet. CMILLER is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in VDS. This result is
then multiplied by the ratio of the application applied VDS
to the Gate charge curve specified VDS. When the IC is
operating in continuous mode the duty cycles for the top
and bottom MOSFETs are given by:
Main Switch Duty Cycle =VOUT
VIN
Synchronous Switch Duty Cycle =V
IN −VOUT
VIN
The MOSFET power dissipations at maximum output
current are given by:
P
MAIN =VOUT
VIN
IMAX
()
21+δ
()
RDS(ON) +
VIN
()
2IMAX
2
⎛
⎝
⎜⎞
⎠
⎟RDR
()
CMILLER
()
•
1
VINTVCC –V
THMIN
+1
VTHMIN
⎡
⎣
⎢⎤
⎦
⎥f
()
P
SYNC =VIN –V
OUT
VIN
IMAX
()
21+δ
()
RDS(ON)
where δ is the temperature dependency of RDS(ON) and
RDR (approximately 2) is the effective driver resistance
at the MOSFET’s Miller threshold voltage. VTHMIN is the
typical MOSFET minimum threshold voltage.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CMILLER actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
a short-circuit when the synchronous switch is on close
to 100% of the period.
The term (1+ δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
The optional Schottky diodes D3 and D4 shown in
Figure 11 conduct during the dead-time between the
conduction of the two power MOSFETs. This prevents
the body diode of the bottom MOSFET from turning on,
storing charge during the dead-time and requiring a
reverse recovery period that could cost as much as 3%
in efficiency at high VIN. A 1A to 3A Schottky is generally
a good compromise for both regions of operation due
to the relatively small average current. Larger diodes
result in additional transition losses due to their larger
junction capacitance.
CIN and COUT Selection
The selection of CIN is simplified by the 2-phase architec-
ture and its impact on the worst-case RMS current drawn
through the input network (battery/fuse/capacitor). It can be
shown that the worst-case capacitor RMS current occurs
when only one controller is operating. The controller with
the highest (VOUT)(IOUT) product needs to be used in the
formula shown in Equation 1 to determine the maximum
RMS capacitor current requirement. Increasing the out-
put current drawn from the other controller will actually
decrease the input RMS ripple current from its maximum
value. The out-of-phase technique typically reduces the
input capacitor’s RMS ripple current by a factor of 30%
to 70% when compared to a single phase power supply
solution.
In continuous mode, the source current of the top MOSFET
is a square wave of duty cycle (VOUT)/(VIN). To prevent
large voltage transients, a low ESR capacitor sized for the
maximum RMS current of one channel must be used. The
maximum RMS capacitor current is given by:
CIN Required IRMS ≈IMAX
VIN
VOUT
()
VIN –V
OUT
()
⎡
⎣⎤
⎦
1/ 2
(1)
This formula has a maximum at VIN = 2VOUT
, where IRMS
= IOUT/2. This simple worst-case condition is commonly
used for design because even significant deviations do not
LTC3890-1
19
38901fa
APPLICATIONS INFORMATION
Figure 5. Setting Output Voltage
offer much relief. Note that capacitor manufacturers’ ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet
size or height requirements in the design. Due to the high
operating frequency of the LTC3890-1, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.
The benefit of the LTC3890-1 2-phase operation can be
calculated by using Equation 1 for the higher power control-
ler and then calculating the loss that would have resulted
if both controller channels switched on at the same time.
The total RMS power lost is lower when both controllers
are operating due to the reduced overlap of current pulses
required through the input capacitor’s ESR. This is why
the input capacitor’s requirement calculated above for the
worst-case controller is adequate for the dual controller
design. Also, the input protection fuse resistance, battery
resistance, and PC board trace resistance losses are also
reduced due to the reduced peak currents in a 2-phase
system. The overall benefit of a multiphase design will
only be fully realized when the source impedance of the
power supply/battery is included in the efficiency testing.
The drains of the top MOSFETs should be placed within
1cm of each other and share a common CIN(s). Separating
the drains and CIN may produce undesirable voltage and
current resonances at VIN.
A small (0.1µF to 1µF) bypass capacitor between the chip
VIN pin and ground, placed close to the LTC3890-1, is
also suggested. A 10 resistor placed between CIN (C1)
and the VIN pin provides further isolation between the
two channels.
The selection of COUT is driven by the effective series
resistance (ESR). Typically, once the ESR requirement
is satisfied, the capacitance is adequate for filtering. The
output ripple (ΔVOUT) is approximated by:
ΔVOUT ≈ΔILESR +1
8•f•C
OUT
⎛
⎝
⎜⎞
⎠
⎟
where f is the operating frequency, COUT is the output
capacitance and ΔIL is the ripple current in the inductor.
The output ripple is highest at maximum input voltage
since ΔIL increases with input voltage.
Setting Output Voltage
The LTC3890-1 output voltages are each set by an exter-
nal feedback resistor divider carefully placed across the
output, as shown in Figure 5. The regulated output voltage
is determined by:
VOUT =0.8V 1+RB
RA
⎛
⎝
⎜⎞
⎠
⎟
To improve the frequency response, a feedforward ca-
pacitor, CFF
, may be used. Great care should be taken to
route the VFB line away from noise sources, such as the
inductor or the SW line.
1/2 LTC3890-1
VFB
VOUT
RBCFF
RA
38901 F05
Tracking and Soft-Start (TRACK/SS Pins)
The start-up of each VOUT is controlled by the voltage on
the respective TRACK/SS pin. When the voltage on the
TRACK/SS pin is less than the internal 0.8V reference, the
LTC3890-1 regulates the VFB pin voltage to the voltage on
the TRACK/SS pin instead of 0.8V. The TRACK/SS pin can
be used to program an external soft-start function or to
allow VOUT to track another supply during start-up.
Soft-start is enabled by simply connecting a capacitor
from the TRACK/SS pin to ground, as shown in Figure 6.
An internal 1µA current source charges the capacitor,
providing a linear ramping voltage at the TRACK/SS pin.
The LTC3890-1 will regulate the VFB pin (and hence VOUT)
according to the voltage on the TRACK/SS pin, allowing
VOUT to rise smoothly from 0V to its final regulated value.
The total soft-start time will be approximately:
tSS =CSS •0.8V
1µA
LTC3890-1
20
38901fa
APPLICATIONS INFORMATION
(7a) Coincident Tracking
(7b) Ratiometric Tracking
Figure 8. Using the TRACK/SS Pin for Tracking
Figure 7. Two Different Modes of Output Voltage Tracking
Alternatively, the TRACK/SS pin can be used to track two
(or more) supplies during start-up, as shown qualitatively
in Figures 7a and 7b. To do this, a resistor divider should
be connected from the master supply (VX) to the TRACK/
SS pin of the slave supply (VOUT), as shown in Figure 8.
During start-up VOUT will track VX according to the ratio
set by the resistor divider:
VX
VOUT
=RA
RTRACKA
•RTRACKA +RTRACKB
RA+RB
For coincident tracking (VOUT = VX during start-up):
R
A = RTRACKA
R
B = RTRACKB
INTVCC Regulators
The LTC3890-1 features two separate internal P-channel
low dropout linear regulators (LDO) that supply power at
the INTVCC pin from either the VIN supply pin or the EXTVCC
pin depending on the connection of the EXTVCC pin. INTVCC
powers the gate drivers and much of the LTC3890-1’s
internal circuitry. The VIN LDO and the EXTVCC LDO regulate
INTVCC to 5.1V. Each of these can supply a peak current of
50mA and must be bypassed to ground with a minimum
of 4.7µF ceramic capacitor. No matter what type of bulk
capacitor is used, an additional 1µF ceramic capacitor
placed directly adjacent to the INTVCC and PGND IC pins is
highly recommended. Good bypassing is needed to supply
the high transient currents required by the MOSFET gate
drivers and to prevent interaction between the channels.
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maxi-
mum junction temperature rating for the LTC3890-1 to be
exceeded. The INTVCC current, which is dominated by the
gate charge current, may be supplied by either the VIN
LDO or the EXTVCC LDO. When the voltage on the EXTVCC
TIME
VX(MASTER)
VOUT(SLAVE)
OUTPUT VOLTAGE
38901 F07a
TIME 38901 F07b
VX(MASTER)
VOUT(SLAVE)
OUTPUT VOLTAGE
1/2 LTC3890-1
VOUT
Vx
VFB
TRACK/SS
38901 F08
RB
RA
RTRACKA
RTRACKB
1/2 LTC3890-1
TRACK/SS
CSS
SGND
38901 F06
Figure 6. Using the TRACK/SS Pin to Program Soft-Start
LTC3890-1
21
38901fa
APPLICATIONS INFORMATION
pin is less than 4.7V, the VIN LDO is enabled. Power dis-
sipation for the IC in this case is highest and is equal to
VIN • IINTVCC. The gate charge current is dependent on
operating frequency as discussed in the Efficiency Con-
siderations section. The junction temperature can be
estimated by using the equations given in Note 3 of the
Electrical Characteristics. For example, the LTC3890-1
INTVCC current is limited to less than 15mA from a 40V
supply when not using the EXTVCC supply at 70°C ambi-
ent temperature:
T
J = 70°C + (15mA)(40V)(90°C/W) = 125°C
To prevent the maximum junction temperature from be-
ing exceeded, the input supply current must be checked
while operating in forced continuous mode (PLLIN/MODE
= INTVCC) at maximum VIN.
When the voltage applied to EXTVCC rises above 4.7V, the
VIN LDO is turned off and the EXTVCC LDO is enabled. The
EXTVCC LDO remains on as long as the voltage applied to
EXTVCC remains above 4.5V. The EXTVCC LDO attempts
to regulate the INTVCC voltage to 5.1V, so while EXTVCC
is less than 5.1V, the LDO is in dropout and the INTVCC
voltage is approximately equal to EXTVCC. When EXTVCC
is greater than 5.1V, up to an absolute maximum of 14V,
INTVCC is regulated to 5.1V.
Using the EXTVCC LDO allows the MOSFET driver and
control power to be derived from one of the LTC3890-1’s
switching regulator outputs (4.7V ≤ VOUT ≤ 14V) during
normal operation and from the VIN LDO when the out-
put is out of regulation (e.g., start-up, short-circuit). If
more current is required through the EXTVCC LDO than
is specified, an external Schottky diode can be added
between the EXTVCC and INTVCC pins. In this case, do
not apply more than 6V to the EXTVCC pin and make sure
that EXTVCC ≤ VIN.
Significant efficiency and thermal gains can be realized
by powering INTVCC from the output, since the VIN cur-
rent resulting from the driver and control currents will be
scaled by a factor of (Duty Cycle)/(Switcher Efficiency).
For 5V to 14V regulator outputs, this means connecting
the EXTVCC pin directly to VOUT
. Tying the EXTVCC pin to
an 8.5V supply reduces the junction temperature in the
previous example from 125°C to:
T
J = 70°C + (15mA)(8.5V)(90°C/W) = 82°C
However, for 3.3V and other low voltage outputs, additional
circuitry is required to derive INTVCC power from the output.
The following list summarizes the four possible connec-
tions for EXTVCC:
1. EXTVCC Grounded. This will cause INTVCC to be powered
from the internal 5.1V regulator resulting in an efficiency
penalty of up to 10% at high input voltages.
2. EXTVCC Connected directly to VOUT
. This is the normal
connection for a 5V to 14V regulator and provides the
highest efficiency.
3.
EXTVCC Connected to an External supply. If an external
supply is available in the 5V to 14V range, it may be
used to power EXTVCC providing it is compatible with
the MOSFET gate drive requirements. Ensure that
EXTVCC < VIN.
4. EXTVCC Connected to an Output-Derived Boost Network.
For 3.3V and other low voltage regulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage that has been boosted to greater
than 4.7V. This can be done with the capacitive charge
pump shown in Figure 9. Ensure that EXTVCC < VIN.
EXTVCC
VIN
TG1
SW
BG1
PGND
1/2 LTC3890-1 RSENSE
VOUT
NDS7002
COUT
38901 F09
MBOT
MTOP
CIN
L
D
BAT85 BAT85
BAT85
Figure 9. Capacitive Charge Pump for EXTVCC
LTC3890-1
22
38901fa
APPLICATIONS INFORMATION
Topside MOSFET Driver Supply (CB, DB)
External bootstrap capacitors, CB, connected to the BOOST
pins supply the gate drive voltages for the topside MOSFETs.
Capacitor CB in the Functional Diagram is charged though
external diode DB from INTVCC when the SW pin is low.
When one of the topside MOSFETs is to be turned on, the
driver places the CB voltage across the gate-source of the
desired MOSFET. This enhances the top MOSFET switch
and turns it on. The switch node voltage, SW, rises to VIN
and the BOOST pin follows. With the topside MOSFET
on, the boost voltage is above the input supply: VBOOST =
VIN + VINTVCC. The value of the boost capacitor, CB, needs
to be 100 times that of the total input capacitance of the
topside MOSFET(s). The reverse breakdown of the external
Schottky diode must be greater than VIN(MAX).
The external diode DB can be a Schottky diode or silicon
diode, but in either case it should have low-leakage and
fast recovery. Pay close attention to the reverse leakage
current specification for this diode, especially at high
temperatures where it generally increases substantially.
For applications with output voltages greater than ~5V
that are switching infrequently, a leaky diode DB can fully
discharge the bootstrap capacitor CB, creating a current
path from the output voltage to the BOOST pin to INTVCC.
Not only does this increase the quiescent current of the
converter, but it can cause INTVCC to rise to dangerous
levels if the leakage exceeds the current consumption on
INTVCC.
Particularly, this is a concern in Burst Mode operation at
no load or very light loads, where the part is switching
very infrequently and the current draw on INTVCC is very
low (typically about 35µA). Generally, pulse-skipping and
forced continuous modes are less sensitive to leakage,
since the more frequent switching keeps the bootstrap
capacitor CB charged, preventing a current path from the
output voltage to INTVCC.
However, in cases where the converter has been operat-
ing (in any mode) and then is shut down, if the leakage
of diode DB fully discharges the bootstrap capacitor CB
before the output voltage discharges to below ~5V, then
the leakage current path can be created from the output
voltage to INTVCC. In shutdown, the INTVCC pin is able to
sink about 30µA. To accommodate diode leakage greater
than this amount in shutdown, INTVCC can be loaded
with an external resistor or clamped with a Zener diode.
Alternatively, the PGOOD resistor can be used to sink the
current (assuming the resistor pulls up to INTVCC) since
PGOOD is pulled low when the converter is shut down.
Nonetheless, using a low-leakage diode is the best choice
to maintain low quiescent current under all conditions.
Fault Conditions: Current Limit and Current Foldback
The LTC3890-1 includes current foldback to help limit
load current when the output is shorted to ground. If
the output voltage falls below 70% of its nominal output
level, then the maximum sense voltage is progressively
lowered from 100% to 45% of its maximum selected
value. Under short-circuit conditions with very low duty
cycles, the LTC3890-1 will begin cycle skipping in order to
limit the short-circuit current. In this situation the bottom
MOSFET will be dissipating most of the power but less
than in normal operation. The short-circuit ripple current
is determined by the minimum on-time. tON(MIN), of the
LTC3890-1 (≈90ns), the input voltage and inductor value:
ΔIL(SC) =tON(MIN)
VIN
L
⎛
⎝
⎜⎞
⎠
⎟
The resulting average short-circuit current is:
ISC =45% •ILIM(MAX) –1
2ΔIL(SC)
Fault Conditions: Overvoltage Protection (Crowbar)
The overvoltage crowbar is designed to blow a system
input fuse when the output voltage of the regulator rises
much higher than nominal levels. The crowbar causes huge
currents to flow, that blow the fuse to protect against a
shorted top MOSFET if the short occurs while the control-
ler is operating.
A comparator monitors the output for overvoltage condi-
tions. The comparator detects faults greater than 10%
above the nominal output voltage. When this condition
is sensed, the top MOSFET is turned off and the bottom
MOSFET is turned on until the overvoltage condition is
cleared. The bottom MOSFET remains on continuously for
as long as the overvoltage condition persists; if VOUT returns
to a safe level, normal operation automatically resumes.
LTC3890-1
23
38901fa
APPLICATIONS INFORMATION
Figure 10. Relationship Between Oscillator Frequency
and Resistor Value at the FREQ Pin
A shorted top MOSFET will result in a high current condition
which will open the system fuse. The switching regulator
will regulate properly with a leaky top MOSFET by altering
the duty cycle to accommodate the leakage.
Phase-Locked Loop and Frequency Synchronization
The LTC3890-1 has an internal phase-locked loop (PLL)
comprised of a phase frequency detector, a lowpass filter,
and a voltage-controlled oscillator (VCO). This allows the
turn-on of the top MOSFET of controller 1 to be locked to
the rising edge of an external clock signal applied to the
PLLIN/MODE pin. The turn-on of controller 2’s top MOSFET
is thus 180 degrees out of phase with the external clock.
The phase detector is an edge sensitive digital type that
provides zero degrees phase shift between the external
and internal oscillators. This type of phase detector does
not exhibit false lock to harmonics of the external clock.
If the external clock frequency is greater than the internal
oscillator’s frequency, fOSC, then current is sourced continu-
ously from the phase detector output, pulling up the VCO
input. When the external clock frequency is less than fOSC,
current is sunk continuously, pulling down the VCO input.
If the external and internal frequencies are the same but
exhibit a phase difference, the current sources turn on for
an amount of time corresponding to the phase difference.
The voltage at the VCO input is adjusted until the phase
and frequency of the internal and external oscillators are
identical. At the stable operating point, the phase detector
output is high impedance and the internal filter capacitor,
CLP, holds the voltage at the VCO input.
Note that the LTC3890-1 can only be synchronized to an
external clock whose frequency is within range of the
LTC3890-1’s internal VCO, which is nominally 55kHz to
1MHz. This is guaranteed to be between 75kHz and 850kHz.
Typically, the external clock (on the PLLIN/MODE pin) input
high threshold is 1.6V, while the input low threshold is 1.1V.
Rapid phase locking can be achieved by using the FREQ pin
to set a free-running frequency near the desired synchro-
nization frequency. The VCO’s input voltage is prebiased
at a frequency corresponding to the frequency set by the
FREQ pin. Once prebiased, the PLL only needs to adjust
the frequency slightly to achieve phase lock and synchro-
nization. Although it is not required that the free-running
frequency be near external clock frequency, doing so will
prevent the operating frequency from passing through a
large range of frequencies as the PLL locks.
Table 2 summarizes the different states in which the FREQ
pin can be used.
Table 2
FREQ PIN PLLIN/MODE PIN FREQUENCY
0V DC Voltage 350kHz
INTVCC DC Voltage 535kHz
Resistor DC Voltage 50kHz to 900kHz
Any of the Above External Clock Phase-Locked to
External Clock
Minimum On-Time Considerations
Minimum on-time, tON(MIN), is the smallest time dura-
tion that the LTC3890-1 is capable of turning on the top
MOSFET. It is determined by internal timing delays and the
gate charge required to turn on the top MOSFET. Low duty
cycle applications may approach this minimum on-time
limit and care should be taken to ensure that:
tON(MIN) <VOUT
VIN f
()
If the duty cycle falls below what can be accommodated
by the minimum on-time, the controller will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
FREQ PIN RESISTOR (k)
15
FREQUENCY (kHz)
600
800
1000
35 45 5525
38901 F10
400
200
500
700
900
300
100
065 75 85 95 105 115 125
LTC3890-1
24
38901fa
APPLICATIONS INFORMATION
The minimum on-time for the LTC3890-1 is approximately
90ns. However, as the peak sense voltage decreases the
minimum on-time gradually increases up to about TBDns.
This is of particular concern in forced continuous applica-
tions with low ripple current at light loads. If the duty cycle
drops below the minimum on-time limit in this situation,
a significant amount of cycle skipping can occur with cor-
respondingly larger current and voltage ripple.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percent-
age of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3890-1 circuits: 1) IC VIN current, 2) INTVCC
regulator current, 3) I2R losses, 4) topside MOSFET
transition losses
.
1. The VIN current is the DC supply current given in the
Electrical Characteristics table, which excludes MOSFET
driver and control currents. VIN current typically results
in a small (<0.1%) loss.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge, dQ, moves
from INTVCC to ground. The resulting dQ/dt is a current
out of INTVCC that is typically much larger than the
control circuit current. In continuous mode, IGATECHG
= f(QT + QB), where QT and QB are the gate charges of
the topside and bottom side MOSFETs.
Supplying INTVCC from an output-derived source power
through EXTVCC will scale the VIN current required for
the driver and control circuits by a factor of (Duty Cycle)/
(Efficiency). For example, in a 20V to 5V application,
10mA of INTVCC current results in approximately 2.5mA
of VIN current. This reduces the midcurrent loss from
10% or more (if the driver was powered directly from
VIN) to only a few percent.
3. I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resis-
tor, and input and output capacitor ESR. In continuous
mode the average output current flows through L and
RSENSE, but is chopped between the topside MOSFET
and the synchronous MOSFET. If the two MOSFETs have
approximately the same RDS(ON), then the resistance
of one MOSFET can simply be summed with the resis-
tances of L, RSENSE and ESR to obtain I2R losses. For
example, if each RDS(ON) = 30m, RL = 50m, RSENSE
= 10m and RESR = 40m (sum of both input and
output capacitance losses), then the total resistance
is 130m. This results in losses ranging from 3% to
13% as the output current increases from 1A to 5A for
a 5V output, or a 4% to 20% loss for a 3.3V output.
Efficiency varies as the inverse square of VOUT for the
same external components and output power level. The
combined effects of increasingly lower output voltages
and higher currents required by high performance digital
systems is not doubling but quadrupling the importance
of loss terms in the switching regulator system!
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high
input voltages (t
y
pically 15V or greater). Transition
losses can be estimated from:
Transition Loss = (1.7) • VIN • 2 • IO(MAX) • CRSS • f
Other hidden losses such as copper trace and internal
battery resistances can account for an additional 5%
to 10% efficiency degradation in portable systems. It
is very important to include these system level losses
during the design phase. The internal battery and fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and very low ESR at
the switching frequency. A 25W supply will typically
require a minimum of 20µF to 40µF of capacitance
having a maximum of 20m to 50m of ESR. The
LTC3890-1 2-phase architecture typically halves this
LTC3890-1
25
38901fa
APPLICATIONS INFORMATION
input capacitance requirement over competing solu-
tions. Other losses including Schottky conduction losses
during dead-time and inductor core losses generally
account for less than 2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by an
amount equal to ΔILOAD (ESR), where ESR is the effective
series resistance of COUT
. ΔILOAD also begins to charge or
discharge COUT generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. OPTI-
LOOP compensation allows the transient response to be
optimized over a wide range of output capacitance and
ESR values. The availability of the ITH pin not only allows
optimization of control loop behavior, but it also provides
a DC coupled and AC filtered closed-loop response test
point. The DC step, rise time and settling at this test
point truly reflects the closed-loop response. Assuming
a predominantly second order system, phase margin and/
or damping factor can be estimated using the percentage
of overshoot seen at this pin. The bandwidth can also
be estimated by examining the rise time at the pin. The
ITH external components shown in Figure 13 circuit will
provide an adequate starting point for most applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1µs to 10µs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking
the feedback loop.
Placing a power MOSFET directly across the output ca-
pacitor and driving the gate with an appropriate signal
generator is a practical way to produce a realistic load step
condition. The initial output voltage step resulting from
the step change in output current may not be within the
bandwidth of the feedback loop, so this signal cannot be
used to determine phase margin. This is why it is better
to look at the ITH pin signal which is in the feedback loop
and is the filtered and compensated control loop response.
The gain of the loop will be increased by increasing RC
and the bandwidth of the loop will be increased by de-
creasing CC. If RC is increased by the same factor that CC
is decreased, the zero frequency will be kept the same,
thereby keeping the phase shift the same in the most
critical frequency range of the feedback loop. The output
voltage settling behavior is related to the stability of the
closed-loop system and will demonstrate the actual overall
supply performance.
A second, more severe transient is caused by switching
in loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT
, causing a rapid drop in VOUT
. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than 1:50, the switch rise time
should be controlled so that the load rise time is limited
to approximately 25 • CLOAD. Thus a 10µF capacitor would
require a 250µs rise time, limiting the charging current
to about 200mA.
Design Example
As a design example for one channel, assume VIN =
12V(nominal), VIN = 22V (max), VOUT = 3.3V, IMAX = 5A,
VSENSE(MAX) = 75mV and f = 350kHz.
The inductance value is chosen first based on a 30% ripple
current assumption. The highest value of ripple current
occurs at the maximum input voltage. Tie the FREQ pin
to GND, generating 350kHz operation. The minimum
inductance for 30% ripple current is:
ΔIL=VOUT
f
()
L
()
1– VOUT
VIN(NOM)
⎛
⎝
⎜
⎜
⎞
⎠
⎟
⎟
LTC3890-1
26
38901fa
APPLICATIONS INFORMATION
A 4.7µH inductor will produce 29% ripple current. The
peak inductor current will be the maximum DC value plus
one half the ripple current, or 5.73A. Increasing the ripple
current will also help ensure that the minimum on-time
of 95ns is not violated. The minimum on-time occurs at
maximum VIN:
tON(MIN) =VOUT
VIN(MAX) f
()
=3.3V
22V 350kHz
()
=429ns
The equivalent RSENSE resistor value can be calculated by
using the minimum value for the maximum current sense
threshold (43mV):
RSENSE ≤64mV
5.73A ≈0.01Ω
Choosing 1% resistors: RA = 25k and RB = 78.7k yields
an output voltage of 3.32V.
The power dissipation on the topside MOSFET can be easily
estimated. Choosing a Fairchild FDS6982S dual MOSFET
results in: RDS(ON) = 0.035/0.022, CMILLER = 215pF. At
maximum input voltage with T(estimated) = 50°C:
P
MAIN =3.3V
22V 5A
()
21+0.005
()
50°C–25°C
()
⎡
⎣⎤
⎦
0.035Ω
()
+22V
()
25A
22.5Ω
()
215pF
()
•
1
5V – 2.3V +1
2.3V
⎡
⎣
⎢⎤
⎦
⎥350kHz
()
=331mW
A short-circuit to ground will result in a folded back cur-
rent of:
ISC =34mV
0.01Ω–1
2
95ns 22V
()
4.7µH
⎛
⎝
⎜⎞
⎠
⎟=3.18A
with a typical value of RDS(ON) and δ = (0.005/°C)(25°C)
= 0.125. The resulting power dissipated in the bottom
MOSFET is:
P
SYNC =3.18A
()
21.125
()
0.022Ω
()
=250mW
which is less than under full-load conditions.
CIN is chosen for an RMS current rating of at least 3A at
temperature assuming only this channel is on. COUT is
chosen with an ESR of 0.02 for low output ripple. The
output ripple in continuous mode will be highest at the
maximum input voltage. The output voltage ripple due to
ESR is approximately:
V
ORIPPLE = RESR (ΔIL) = 0.02(1.45A) = 29mVP-P
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the IC. These items are also illustrated graphically in the
layout diagram of Figure 11. Figure 12 illustrates the current
waveforms present in the various branches of the 2-phase
synchronous regulators operating in the continuous mode.
Check the following in your layout:
1. Are the top N-channel MOSFETs MTOP1 and MTOP2
located within 1cm of each other with a common drain
connection at CIN? Do not attempt to split the input
decoupling for the two channels as it can cause a large
resonant loop.
2. Are the signal and power grounds kept separate? The
combined IC signal ground pin and the ground return
of CINTVCC must return to the combined COUT (–) ter-
minals. The path formed by the top N-channel MOSFET,
Schottky diode and the CIN capacitor should have short
leads and PC trace lengths. The output capacitor (–)
terminals should be connected as close as possible
to the (–) terminals of the input capacitor by placing
the capacitors next to each other and away from the
Schottky loop described above.
3. Do the LTC3890-1 VFB pins’ resistive dividers connect to
the (+) terminals of COUT? The resistive divider must be
connected between the (+) terminal of COUT and signal
ground. The feedback resistor connections should not
be along the high current input feeds from the input
capacitor(s).
LTC3890-1
27
38901fa
APPLICATIONS INFORMATION
4. Are the SENSE– and SENSE+ leads routed together with
minimum PC trace spacing? The filter capacitor between
SENSE+ and SENSE– should be as close as possible
to the IC. Ensure accurate current sensing with Kelvin
connections at the SENSE resistor.
5. Is the INTVCC decoupling capacitor connected close
to the IC, between the INTVCC and the power ground
pins? This capacitor carries the MOSFET drivers’ cur-
rent peaks. An additional 1µF ceramic capacitor placed
immediately next to the INTVCC and PGND pins can help
improve noise performance substantially.
6. Keep the switching nodes (SW1, SW2), top gate nodes
(TG1, TG2), and boost nodes (BOOST1, BOOST2) away
from sensitive small-signal nodes, especially from
the opposites channel’s voltage and current sensing
feedback pins. All of these nodes have very large and
fast moving signals and therefore should be kept on
the output side of the LTC3890-1 and occupy minimum
PC trace area.
7. Use a modified star ground technique: a low impedance,
large copper area central grounding point on the same
side of the PC board as the input and output capacitors
with tie-ins for the bottom of the INTVCC decoupling
capacitor, the bottom of the voltage feedback resistive
divider and the SGND pin of the IC.
PC Board Layout Debugging
Start with one controller on at a time. It is helpful to use
a DC-50MHz current probe to monitor the current in the
inductor while testing the circuit. Monitor the output switch-
ing node (SW pin) to synchronize the oscilloscope to the
internal oscillator and probe the actual output voltage as
well. Check for proper performance over the operating
voltage and current range expected in the application.
The frequency of operation should be maintained over the
input voltage range down to dropout and until the output
load drops below the low current operation threshold—
typically 25% of the maximum designed current level in
Burst Mode operation.
The duty cycle percentage should be maintained from cycle
to cycle in a well-designed, low noise PCB implementa-
tion. Variation in the duty cycle at a subharmonic rate can
suggest noise pickup at the current or voltage sensing
inputs or inadequate loop compensation. Overcompensa-
tion of the loop can be used to tame a poor PC layout if
regulator bandwidth optimization is not required. Only after
each controller is checked for its individual performance
should both controllers be turned on at the same time.
A particularly difficult region of operation is when one
controller channel is nearing its current comparator trip
point when the other channel is turning on its top MOSFET.
This occurs around 50% duty cycle on either channel due
to the phasing of the internal clocks and may cause minor
duty cycle jitter.
Reduce VIN from its nominal level to verify operation of
the regulator in dropout. Check the operation of the un-
dervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher out-
put currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling. If problems are encountered with
high current output loading at lower input voltages, look
for inductive coupling between CIN, Schottky and the top
MOSFET components to the sensitive current and voltage
sensing traces. In addition, investigate common ground
path voltage pickup between these components and the
SGND pin of the IC.
An embarrassing problem, which can be missed in an
otherwise properly working switching regulator, results
when the current sensing leads are hooked up backwards.
The output voltage under this improper hookup will still
be maintained but the advantages of current mode control
will not be realized. Compensation of the voltage loop will
be much more sensitive to component selection. This
behavior can be investigated by temporarily shorting out
the current sensing resistor—don’t worry, the regulator
will still maintain control of the output voltage.
LTC3890-1
28
38901fa
Figure 11. Recommended Printed Circuit Layout Diagram
CB2
CB1
RPU1
PGOOD1
VPULL-UP
CINTVCC CIN
D1*
1µF
CERAMIC
M1 M2
M3 M4 D2*
CVIN
VIN
RIN
L1
L2
COUT1
VOUT1
GND
VOUT2
38901 F11
COUT2
RSENSE
RSENSE
fIN
1µF
CERAMIC
ITH1
VFB1
SENSE1+
SENSE1–
SENSE2–
SENSE2+
VFB2
ITH2
TRACK/SS2
PGOOD1
TG1
SW1
BOOST1
BG1
VIN
PGND
EXTVCC
INTVCC
BG2
BOOST2
SW2
TG2
FREQ
PLLIN/MODE
RUN1
RUN2
SGND
+
+
+
LTC3890-1
C1*
R1*
TRACK/SS1
C2*
R2*
*R1, R2, C1, C2, D1, D2 ARE OPTIONAL.
APPLICATIONS INFORMATION
LTC3890-1
29
38901fa
Figure 12. Branch Current Waveforms
RL1
D1
L1
SW1 RSENSE1 VOUT1
COUT1
VIN
CIN
RIN
RL2
D2
BOLD LINES INDICATE
HIGH SWITCHING
CURRENT. KEEP LINES
TO A MINIMUM LENGTH.
L2
SW2
38901 F12
RSENSE2 VOUT2
COUT2
APPLICATIONS INFORMATION
LTC3890-1
30
38901fa
TYPICAL APPLICATIONS
Efficiency and Power Loss
vs Output Current Efficiency vs Load Current Efficiency vs Input Voltage
Figure 13. High Efficiency Dual 8.5V/3.3V Step-Down Converter
OUTPUT CURRENT (A)
0
EEFICIENCY (%)
POWER LOSS (mW)
70
100
10.10.010.0010.0001 10
38901 TA02b
50
40
30
20 1
10
100
1000
10000
0.1
10
60
80
90
VIN = 12V
VOUT = 3.3V BURST EFFICIENCY
CCM LOSS
PULSE-SKIPPING
LOSS
BURST LOSS
CCM EFFICIENCY
PULSE-SKIPPING
EFFICIENCY
OUTPUT CURRENT (A)
0
EEFICIENCY (%)
70
100
10.10.010.0010.0001 10
38901 TA02c
50
40
30
20
10
60
80
90 VOUT = 8.5V
VOUT = 3.3V
VIN = 12V
INPUT VOLTAGE (V)
80
EEFICIENCY (%)
94
100
20 30 35 40 45 50 55 6015105025
38901 TA02d
90
88
86
84
82
92
96
98
VOUT1 = 3.3V
VOUT2 = 8.5V
ILOAD = 2A
MTOP2
SENSE1+
SENSE1–
VFB1
TRACK/SS2
ITH2
VFB2
SENSE2–
SENSE2+
ITH1
TRACK/SS1
PLLIN/MODE
SGND
EXTVCC
RUN1
RUN2
FREQ
BG1
CITH1A 100pF
CITH1 1000pF
CSS2 0.01µF
CSS1 0.01µF
CITH2 470pF
C1
1nF
COUT1
470µF
RA1
31.6k
RITH1
34.8k
RB1
100k
RB2
100k
RFREQ
41.2k
RITH2
34.8k
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: COILCRAFT SER1360-472KL
L2: COILCRAFT SER1360-802KL
COUT1: SANYO 6TPE470M
COUT2: SANYO 10TPE330M
D1, D2: DFLS1100
D1
D2
LTC3890-1
RSENSE1
8m
RSENSE2
10m
L1
4.7µH
L2
8µH
MBOT1
TG1 MTOP1
VOUT1
3.3V
5A
VIN
9V TO 60V
38901 TA02a
PGOOD1
INTVCC
100k
CB1
0.1µF
RA2
10.5k
SW1
VOUT2
8.5V
3A
SW2
BOOST1
CB2 0.1µF
BOOST2
VIN
INTVCC
PGND
TG2
BG2
COUT2
330µF
CINT
4.7µF
CIN
220µF
MBOT2
C2
1nF
VOUT2
LTC3890-1
31
38901fa
TYPICAL APPLICATIONS
MTOP2
SENSE1+
SENSE1–
VFB1
TRACK/SS2
ITH2
VFB2
SENSE2–
SENSE2+
ITH1
TRACK/SS1
PLLIN/MODE
EXTVCC
RUN1
RUN2
FREQ
PGOOD1
BG1
CITH1A 100pF
CITH1
470pF CSS1 0.01µF
C1
1nF
COUT1
330µF
RA1
10.5k
RITH1 34.8k
RB1
100k
RRUN
1000k
RMODE
100k
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1, L2: COILCRAFT SER1360-802KL
COUT1, COUT2: SANYO 10TPE330M
D1, D2: DFLS1100
D1
D2
LTC3890-1
RSENSE1
10m
RSENSE2
10m
L1
8µH
L2
8µH
MBOT1
TG1 MTOP1
VOUT1
8.5V
6A
VIN
9V TO 60V
38901 TA03
INTVCC
100k
CB1
0.1µF
SW1
SW2
BOOST1
CB2 0.1µF
BOOST2
VIN
INTVCC
PGND
TG2
BG2 COUT2
330µF
CINT
4.7µF
CIN
220µF
MBOT2
C2
1nF
CITH2
100pF
VOUT
INTVCC
VIN
SGND
RFREQ 41.2k
High Efficiency 8.5V Dual-Phase Step-Down Converter
LTC3890-1
32
38901fa
TYPICAL APPLICATIONS
High Efficiency Dual 12V/5V Step-Down Converter
MTOP2
SENSE1+
SENSE1–
VFB1
TRACK/SS2
ITH2
VFB2
SENSE2–
SENSE2+
ITH1
TRACK/SS1
PLLIN/MODE
SGND
EXTVCC
RUN1
RUN2
FREQ
PGOOD1
BG1
CITH1A 100pF
CITH1 470pF
CSS2 0.01µF
CSS1 0.01µF
CITH2 470pF
C1
1nF
COUT1
180µF
RA1
6.98k
RITH1
34.8k
RB1
100k
RB2
100k
RFREQ
41.2k
RITH2
20k
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: COILCRAFT SER1360-802KL
L2: COILCRAFT SER1360-472KL
COUT1: 16SVP180MX
COUT2: SANYO 6TPE470M
D1, D2: DFLS1100
D1
D2
LTC3890-1
RSENSE1
9m
RSENSE2
10m
L1
8µH
L2
4.7µH
MBOT1
TG1 MTOP1
VOUT1
12V
3A
VIN
12.5V TO 60V
38901 TA04
INTVCC
100k
CB1
0.47µF
RA2
18.7k
SW1
VOUT2
5V
5A
SW2
BOOST1
CB2 0.47µF
BOOST2
VIN
INTVCC
PGND
TG2
BG2 COUT2
470µF
CINT
4.7µF
CIN
220µF
MBOT2
C2
1nF
LTC3890-1
33
38901fa
TYPICAL APPLICATIONS
High Efficiency Dual 24V/5V Step-Down Converter
MTOP2
SENSE1+
SENSE1–
VFB1
TRACK/SS2
ITH2
VFB2
SENSE2–
SENSE2+
ITH1
TRACK/SS1
PLLIN/MODE
SGND
EXTVCC
RUN1
RUN2
FREQ
BG1
CITH1A 100pF
CITH1 680pF
CSS2 0.01µF
CSS1 0.01µF
CITH2 470pF
C1
1nF
COUT1
22µF
×2 CERAMIC
RB1
487k
RA1
16.9k
CF1 33pF
RITH1
46k
RFREQ
60k
RITH2
20k
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: SUMIDA CDR7D43MN
L2: COILCRAFT SER1360-472KL
COUT1: KEMET T525D476MO16E035
COUT2: SANYO 6TPE470M
D1, D2: DFLS1100
D1
D2
LTC3890-1
L1
22µH
L2
4.7µH
MBOT1
TG1 MTOP1
VOUT1
24V
1A
VIN
28V TO 60V
38901 TA05
INTVCC
PGOOD1
100k
CB1
0.47µF
RA2
18.7k RB2 100k
C2
1nF
SW1
VOUT2
5V
5A
SW2
BOOST1
CB2 0.47µF
BOOST2
VIN
INTVCC
PGND
TG2
BG2 COUT2
470µF
CINT
4.7µF
CIN
220µF
MBOT2
RSENSE1
25m
RSENSE2
10m
LTC3890-1
34
38901fa
PACKAGE DESCRIPTION
GN Package
28-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.386 – .393*
(9.804 – 9.982)
GN28 (SSOP) 0204
12
345678 9 10 11 12
.229 – .244
(5.817 – 6.198)
.150 – .157**
(3.810 – 3.988)
202122232425262728 19 18 17
13 14
1615
.016 – .050
(0.406 – 1.270)
.015 p .004
(0.38 p 0.10) s 45o
0o – 8o TYP
.0075 – .0098
(0.19 – 0.25)
.0532 – .0688
(1.35 – 1.75)
.008 – .012
(0.203 – 0.305)
TYP
.004 – .0098
(0.102 – 0.249)
.0250
(0.635)
BSC
.033
(0.838)
REF
.254 MIN
RECOMMENDED SOLDER PAD LAYOUT
.150 – .165
.0250 BSC.0165 p.0015
.045 p.005
* DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
INCHES
(MILLIMETERS)
NOTE:
1. CONTROLLING DIMENSION: INCHES
2. DIMENSIONS ARE IN
3. DRAWING NOT TO SCALE
LTC3890-1
35
38901fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
REVISION HISTORY
REV DATE DESCRIPTION PAGE NUMBER
A 01/11 Added MP-grade and H-grade. Changes reflected throughout the data sheet. 1-38
LTC3890-1
36
38901fa
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2010
LT 0111 REV A • PRINTED IN USA
TYPICAL APPLICATION
RELATED PARTS
High Efficiency Dual 12V/3.3V Step-Down Converter
PART NUMBER DESCRIPTION COMMENTS
LT3891 60V, Low IQ, Synchronous Step-Down DC/DC Controller PLL Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 60V,
0.8V ≤ VOUT ≤ 24V, TSSOP-20E, 3mm × 4mm QFN-20
LTC3857/LTC3857-1/
LTC3858/LTC3858-1
Low IQ, Dual Output 2-Phase Synchronous Step-Down
DC/DC Controllers with 99% Duty Cycle
PLL Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 38V,
0.8V ≤ VOUT ≤ 24V, IQ = 50µA/170µA
LTC3834/LTC3834-1/
LTC3835/LTC3835-1
Low IQ, Single Output Synchronous Step-Down
DC/DC Controllers with 99% Duty Cycle
PLL Fixed Frequency 140kHz to 650kHz, 4V ≤ VIN ≤ 36V,
0.8V ≤ VOUT ≤ 10V, IQ = 30µA/80µA
LTC3810 100V Synchronous Step-Down DC/DC Controller Constant On-Time Valley Current Mode, 4V ≤ VIN ≤ 100V,
0.8V ≤ VOUT ≤ 0.93VIN, SSOP-28
LTC3859 Low IQ, Triple Output Buck/Buck/Boost Synchronous
DC/DC Controller
Outputs (≥5V) Remain in Regulation Through Cold Crank,
2.5V ≤ VIN ≤ 38V, VOUT(BUCK) Up to 24V, VOUT(BOOST) Up to 60V
MTOP2
SENSE1+
SENSE1–
VFB1
TRACK/SS2
ITH2
VFB2
SENSE2–
SENSE2+
ITH1
TRACK/SS1
PLLIN/MODE
SGND
EXTVCC
RUN1
RUN2
FREQ
PGOOD1
BG1
CITH1A 100pF
CITH1 470pF
CSS2 0.01µF
CSS1 0.01µF
CITH2 1000pF
CITH2A
100pF
C1
1nF
COUT1
180µF
RA1
6.98k
RITH1
34.8k
RB1
100k
RB2
100k
RFREQ
41.2k
RITH2
34.8k
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: COILCRAFT SER1360-802KL
L2: COILCRAFT SER1360-472KL
COUT1: 16SVP180MX
COUT2: SANYO 6TPE470M
D1, D2: DFLS1100
D1
D2
LTC3890-1
RSENSE1
9m
RSENSE2
10m
L1
8µH
L2
4.7µH
MBOT1
TG1 MTOP1
VOUT1
12V
3A
VIN
12.5V TO 60V
38901 TA06
INTVCC
100k
CB1
0.47µF
RA2
31.6k
SW1
VOUT2
3.3V
5A
SW2
BOOST1
CB2 0.47µF
BOOST2
VIN
INTVCC
PGND
TG2
BG2 COUT2
470µF
CINT
4.7µF
CIN
220µF
MBOT2
C2
1nF
VOUT1
