LM4781
LM4781 Overture Audio Power Amplifier Series 3 Channel 35W Audio Power
Amplifier with Mute
Literature Number: SNAS208
LM4781 Overture™Audio Power Amplifier Series
3 Channel 35W Audio Power Amplifier with Mute
General Description
The LM4781 is a three channel audio amplifier capable of
typically delivering 35W per channel of continuous average
output power into an 8Ωload with less than 0.5% THD+N
from 20Hz - 20kHz.
The LM4781 is fully protected utilizing National’s Self Peak
Instantaneous Temperature (˚Ke) (SPiKe
TM
) protection cir-
cuitry. SPiKe provides a dynamically optimized Safe Operat-
ing Area (SOA). SPiKe protection completely safeguards the
LM4781’s outputs against over-voltage, under-voltage, over-
loads, shorts to the supplies or GND, thermal runaway and
instantaneous temperature peaks. The advanced protection
features of the LM4781 places it in a class above discrete
and hybrid amplifiers.
Each amplifier of the LM4781 has an independent smooth
transition fade-in/out mute.
The LM4781 can easily be configured for bridge or parallel
operation for higher power and bi-amp solutions.
Key Specifications
jOutput Power/Channel with 0.5% THD+N,
1kHz into 8Ω35W (typ)
jTHD+N at 3 x 20W into 8Ω(20Hz - 20kHz) 0.2% (typ)
jTHD+N at 3 x 20W into 6Ω(20Hz - 20kHz) 0.3% (typ)
jTHD+N at 3 x 20W into 4Ω(20Hz - 20kHz) 0.5% (typ)
jMute Attenuation 110dB (typ)
jPSRR 85dB (min)
jSlew Rate 9V/µs (typ)
Features
nSPiKe Protection
nLow external component count
nQuiet fade-in/out mute mode
nWide supply range: 20V - 70V
nSignal-to-Noise Ratio ≥93dB (ref. to P
O
= 1W)
Applications
nAudio amplifier for component stereo
nAudio amplifier for compact stereo
nAudio amplifier for self-powered speakers
nAudio amplifier for high-end and HD TVs
Typical Application
SPiKe™Protection and Overture™are trademarks of National Semiconductor Corporation.
200673B5
FIGURE 1. Typical Audio Amplifier Application Circuit
August 2003
LM4781 Overture
™
Audio Power Amplifier Series
3 Channel 35W Audio Power Amplifier with Mute
© 2003 National Semiconductor Corporation DS200673 www.national.com
Connection Diagrams
Plastic Package (Note 14)
200673B2
Top View
Order Number LM4781TA
See NS Package Number TA27A
TO-220 Top Marking
200673A6
Top View
U - Wafer Fab Code
Z - Assembly Plant Code
XY - Date Code
TT - Die Run Traceability
L4781TA - LM4781TA
LM4781
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Absolute Maximum Ratings (Notes 1,
2)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Supply Voltage |V
+
|+|V
-
|
(No Signal) 72V
Supply Voltage |V
+
|+|V
-
|
(Input Signal) 70V
Common Mode Input Voltage (V
+
or V
-
) and
|V
+
|+|V
-
|≤60V
Differential Input Voltage (Note 13) 60V
Output Current Internally Limited
Power Dissipation (Note 3) 125W
ESD Susceptability (Note 4) 2.0kV
ESD Susceptability (Note 5) 200V
Junction Temperature (T
JMAX
) (Note 9) 150˚C
Soldering Information
TA Package (10 seconds) 260˚C
Storage Temperature -40˚C to +150˚C
Thermal Resistance
θ
JA
30˚C/W
θ
JC
0.9˚C/W
Operating Ratings (Notes 1, 2)
Temperature Range
T
MIN
≤T
A
≤T
MAX
−20˚C ≤T
A
≤+85˚C
Supply Voltage |V
+
|+|V
-
| 20V ≤V
TOTAL
≤70V
Note: Operation is guaranteed up to 70V; however, distor-
tion may be introduced from SPiKe protection circuitry if
proper thermal considerations are not taken into account.
Refer to the Thermal Considerations section for more
information.
Electrical Characteristics (Notes 1, 2)
The following specifications apply for V
+
= +28V, V
-
= −28V, I
MUTE
= -1mA/channel and R
L
=8Ωunless otherwise specified.
Limits apply for T
A
= 25˚C.
Symbol Parameter Conditions LM4781 Units
(Limits)
Typical Limit
(Note 6) (Notes 7, 8)
|V
+
|+|V
-
| Power Supply Voltage (Note
10)
GND−V
-
≥9V 18 20
70
V (min)
V (max)
A
M
Mute Attenuation I
MUTE
= 0mA 110 80 dB (min)
P
O
Output Power (RMS)
THD+N = 0.5% (max)
f = 1kHz
|V
+
|=|V
-
| = 20V, R
L
=4Ω
|V
+
|=|V
-
| = 24V, R
L
=6Ω
|V
+
|=|V
-
| = 28V, R
L
=8Ω
25
30
35
20
25
30
W (min)
W (min)
W (min)
THD+N Total Harmonic Distortion +
Noise
P
O
= 20W, f = 20Hz - 20kHz
A
V
= 26dB
|V
+
|=|V
-
| = 20V, R
L
=4Ω
|V
+
|=|V
-
| = 24V, R
L
=6Ω
|V
+
|=|V
-
| = 28V, R
L
=8Ω
0.5
0.3
0.2
%
%
%
X
talk
Channel Separation (Note 11) P
O
= 10W, f = 1kHz 70 dB
P
O
= 10W, f = 10kHz 66 dB
SR Slew Rate V
IN
= 1.2V
RMS
, f = 10kHz square
Wave.R
L
=2kΩ
9 5 V/µs (min)
I
DD
Total Quiescent Power V
CM
= 0V 75 150 mA (max)
Supply Current V
O
= 0V, I
O
=0A
V
OS
Input Offset Voltage V
CM
= 0V, I
O
= 0mA 1 10 mV (max)
I
B
Input Bias Current V
CM
= 0V, I
O
= 0mA 0.2 1 µA (max)
I
O
Output Current Limit |V
+
|=|V
-
| = 20V, t
ON
= 10ms 4 3 A (min)
V
OD
Output Dropout Voltage (Note
12)
|V
+
-V
O
|, V
+
= 20V, I
O
= +100mA
|V
O
-V
-
|, V
-
= -20V, I
O
= -100mA
1.5
2.5
4
4
V (max)
V (max)
PSRR Power Supply Rejection Ratio
(Note 15)
V
+
= 30V to 10V, V
-
= -30V
V
CM
= 0V, I
O
= 0mA 125 85 dB (min)
V
+
= 30V, V
-
= -30V to -10V
V
CM
= 0V, I
O
= 0mA 110 85 dB (min)
LM4781
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Electrical Characteristics (Notes 1, 2) (Continued)
The following specifications apply for V
+
= +28V, V
-
= −28V, I
MUTE
= -1mA/channel and R
L
=8Ωunless otherwise specified.
Limits apply for T
A
= 25˚C.
Symbol Parameter Conditions LM4781 Units
(Limits)
Typical Limit
(Note 6) (Notes 7, 8)
CMRR Common Mode Rejection Ratio
(Note 15)
V
+
= 30V to 10V, V
-
= -10V to -30V
V
CM
= 20V to -20V, I
O
= 0mA
110 75 dB (min)
A
VOL
Open Loop Voltage Gain |V
+
|=|V
-
| = 28V, R
L
=2kΩ
∆V
O
= 20V 115 80 dB (min)
GBWP Gain Bandwidth Product |V
+
|=|V
-
| = 28V, f
IN
= 100kHz
V
IN
= 50mV
RMS
8 2 MHz (min)
e
IN
Input Noise IHF-A-Weighting Filter,
R
IN
= 600Ω(Input Referred)
3.0 8 µV (max)
SNR Signal-to-Noise Ratio
P
O
=1W
RMS
; A-Weighted Filter
f
IN
= 1kHz, R
S
=25Ω93 dB
P
O
= 25W
RMS
; A-Weighted Filter
f
IN
= 1kHz, R
S
=25Ω107 dB
IMD Intermodulation Distortion 60Hz, 7kHz, 4:1 (SMPTE)
60Hz, 7kHz, 1:1 (SMPTE)
0.004
0.006
%
%
Note 1: All voltages are measured with respect to the ground pins, unless otherwise specified.
Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which
guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit
is given; however, the typical value is a good indication of device performance.
Note 3: The maximum power dissipation must be de-rated at elevated temperatures and is dictated by TJMAX,θJC, and the ambient temperature TA. The maximum
allowable power dissipation is PDMAX =(T
JMAX -TA)/θJC or the number given in the Absolute Maximum Ratings, whichever is lower. For the LM4781, TJMAX = 150˚C
and the typical θJC is 0.9˚C/W. Refer to the DETERMINING THE CORRECT HEAT SINK section for more information.
Note 4: Human body model, 100pF discharged through a 1.5kΩresistor.
Note 5: Machine Model: a 220pF - 240pF discharged through all pins.
Note 6: Typical specifications are measured at 25˚C and represent the parametric norm.
Note 7: Tested limits are guaranteed to National’s AOQL (Average Outgoing Quality Level).
Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.
Note 9: The maximum operating junction temperature is 150˚C. However, the instantaneous Safe Operating Area temperature is 250˚C.
Note 10: V-must have at least - 9V at its pin with reference to GND in order for the under-voltage protection circuitry to be disabled. In addition, the voltage
differential between V+and V-must be greater than 14V.
Note 11: Cross talk performance was measured using the demo board shown in the datasheet. PCB layout will affect cross talk. It is recommended that the input
and output traces be separated by as much distance as possible. Return ground traces from outputs should also be independent back to sinlge ground point and
use as wide of traces as possible.
Note 12: The output dropout voltage is the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs. Supply Voltage graph in the Typical
Performance Characteristics section.
Note 13: The Differential Input Voltage Absolute Maximum Rating is based on supply voltages V+= 30V and V-= - 30V.
Note 14: The TA27A is a non-isolated package. The package’s metal back and any heat sink to which it is mounted are connected to the V-potential when using
only thermal compound. If a mica washer is used in addition to thermal compound, θCS (case to sink) is increased, but the heat sink will be electrically isolated from
V-.
Note 15: DC electrical test.
LM4781
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Bridged Amplifier Application Circuit
200673B4
FIGURE 2. Bridged Amplifier Application Circuit
LM4781
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Parallel Amplifier Application Circuit
200673B3
FIGURE 3. Parallel Amplifier Application Circuit
LM4781
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Single Supply Application Circuit
Note: *Optional components dependent upon specific design requirements.
Auxiliary Amplifier Application Circuit
20067306
FIGURE 4. Single Supply Amplifier Application Circuit
200673B6
FIGURE 5. Special Audio Amplifier Application Circuit
LM4781
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External Components Description
(Figures 1-5)
Components Functional Description
1R
B
Prevents current from entering the amplifier’s non-inverting input. This current may pass through to the load
during system power down, because of the amplifier’s low input impedance when the undervoltage circuitry
is off. This phenomenon occurs when the V
+
and V
-
supply voltages are below 1.5V.
2R
i
Inverting input resistance. Along with R
f
, sets AC gain.
3R
f
Feedback resistance. Along with R
i
, sets AC gain.
4R
f2
(Note 16)
Feedback resistance. Works with Cf and Rf creating a lowpass filter that lowers AC gain at high
frequencies. The -3dB point of the pole occurs when: (R
f
-R
i
)/2=R
f
// [1/(2πf
c
C
f
)+R
f2
] for the
Non-Inverting configuration shown in Figure 5.
5C
f
(Note 16)
Compensation capacitor. Works with R
f
and R
f2
to reduce AC gain at higher frequencies.
6C
C
(Note 16)
Compensation capacitor. Reduces the gain at higher frequencies to avoid quasi-saturation oscillations of the
output transistor. Also suppresses external electromagnetic switching noise created from fluorescent lamps.
7C
i
(Note 16)
Feedback capacitor which ensures unity gain at DC. Along with R
i
also creates a highpass filter at f
c
=
1/(2πR
i
C
i
).
8C
S
Provides power supply filtering and bypassing. Refer to the Supply Bypassing application section for proper
placement and selection of bypass capacitors.
9R
V
(Note 16)
Acts as a volume control by setting the input voltage level.
10 R
IN
(Note 16)
Sets the amplifier’s input terminals DC bias point when C
IN
is present in the circuit. Also works with C
IN
to
create a highpass filter at f
C
= 1/(2πR
IN
C
IN
). If the value of R
IN
is too large, oscillations may be observed on
the outputs when the inputs are floating. Recommended values are 10kΩto 47kΩ. Refer to Figure 5.
11 C
IN
(Note 16)
Input capacitor. Prevents the input signal’s DC offsets from being passed onto the amplifier’s inputs.
12 R
SN
(Note 16)
Works with C
SN
to stabilize the output stage by creating a pole that reduces high frequency instabilities.
13 C
SN
(Note 16)
Works with R
SN
to stabilize the output stage by creating a pole that reduces high frequency instabilities. The
pole is set at f
C
= 1/(2πR
SN
C
SN
). Refer to Figure 5.
14 L (Note 16) Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce
the Q of the series resonant circuit. Also provides a low impedance at low frequencies to short out R and
pass audio signals to the load. Refer to Figure 5.
15 R (Note 16)
16 R
A
Provides DC voltage biasing for the transistor Q1 in single supply operation.
17 C
A
Provides bias filtering for single supply operation.
18 R
INP
(Note 16)
Limits the voltage difference between the amplifier’s inputs for single supply operation. Refer to the Clicks
and Pops application section for a more detailed explanation of the function of R
INP
.
19 R
BI
Provides input bias current for single supply operation. Refer to the Clicks and Pops application section for
a more detailed explanation of the function of R
BI
.
20 R
E
Establishes a fixed DC current for the transistor Q1 in single supply operation. This resistor stabilizes the
half-supply point along with C
A
.
21 R
M
Mute resistance set up to allow 0.5mA to be drawn from each MUTE pin to turn the muting function off.
→R
M
is calculated using: R
M
≤(|V
EE
| − 2.6V)/l where l ≥0.5mA. Refer to the Mute Attenuation vs Mute
Current curves in the Typical Performance Characteristics section.
22 C
M
Mute capacitance set up to create a large time constant for turn-on and turn-off muting.
23 S
1
Mute switch. When open or switched to GND, the amplifier will be in mute mode.
24 R
OUT
Reduces current flow between outputs that are caused by Gain or DC offset differences between the
amplifiers.
Note 16: Optional components dependent upon specific design requirements.
LM4781
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Optional External Component
Interaction
Although the optional external components have specific
desired functions that are designed to reduce the bandwidth
and eliminate unwanted high frequency oscillations they may
cause certain undesirable effects when they interact. Inter-
action may occur for components whose reactances are in
close proximity to one another. One example would be the
coupling capacitor, C
C
, and the compensation capacitor, C
f
.
These two components act as low impedances to certain
frequencies which will couple signals from the input to the
output. Please take careful note of basic amplifier compo-
nent functionality when designing in these components.
The optional external components shown in Figure 4 and
Figure 5 and described above are applicable in both single
and split voltage supply configurations.
Typical Performance Characteristics
THD+N vs Frequency
±20V, P
OUT
= 1W & 20W/Channel
R
L
=4Ω, 80kHz BW
THD+N vs Frequency
±24V, P
OUT
= 1W & 20W/Channel
R
L
=6Ω, 80kHz BW
20067325 20067343
THD+N vs Frequency
±28V, P
OUT
= 1W & 20W/Channel
R
L
=8Ω, 80kHz BW
THD+N vs Output Power/Channel
±20V, R
L
=4Ω, 80kHz BW
20067326 20067344
LM4781
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Typical Performance Characteristics (Continued)
THD+N vs Output Power/Channel
±24V, R
L
=6Ω, 80kHz BW
THD+N vs Output Power/Channel
±28V, R
L
=8Ω, 80kHz BW
20067327 20067336
Output Power/Channel
vs Supply Voltage
f = 1kHz, R
L
=4Ω, 80kHz BW
Output Power/Channel
vs Supply Voltage
f = 1kHz, R
L
=6Ω, 80kHz BW
20067331 20067340
Output Power/Channel
vs Supply Voltage
f = 1kHz, R
L
=8Ω, 80kHz BW
Total Power Dissipation
vs Output Power/Channel
1% THD (max), R
L
=4Ω, 80kHz BW
20067323 20067330
LM4781
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Typical Performance Characteristics (Continued)
Total Power Dissipation
vs Output Power/Channel
1% THD (max), R
L
=6Ω, 80kHz BW
Total Power Dissipation
vs Output Power/Channel
1% THD (max), R
L
=8Ω, 80kHz BW
20067339 20067322
Crosstalk vs Frequency
±20V, P
OUT
= 10W,
R
L
=4Ω, 80kHz BW
Crosstalk vs Frequency
±28V, P
OUT
= 10W,
R
L
=8Ω, 80kHz BW
20067338 200673C0
Two Channels On Crosstalk vs Frequency
±20V, P
OUT
= 10W,
R
L
=4Ω, 80kHz BW
Two Channels On Crosstalk vs Frequency
±28V, P
OUT
= 10W,
R
L
=8Ω, 80kHz BW
20067348 20067329
LM4781
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Typical Performance Characteristics (Continued)
Mute Attenuation vs
Mute Pin Current
P
OUT
= 10W/Channel
Supply Current vs
Supply Voltage
20067321 20067333
Large Signal Response
Power Supply
Rejection Ratio
200673C1 200673C2
Common-Mode
Rejection Ratio
Open Loop
Frequency Response
200673C3 200673C4
LM4781
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Typical Performance Characteristics (Continued)
Clipping Voltage
vs Supply Voltage
THD+N vs Frequency
±20V, P
OUT
= 1W & 25W,
Chs 1&3 in Bridge Mode (Note17),
R
L
=8Ω, 80kHz BW
200673C5 20067335
THD+N vs Frequency
±28V, P
OUT
= 1W & 35W,
Chs 1&3 in Parallel Mode (Note18),
R
L
=4Ω, 80kHz BW
THD+N vs Frequency
±24V, P
OUT
= 1W & 45W,
All Chs in Parallel Mode (Note18),
R
L
=2Ω, 80kHz BW
20067334 20067342
THD+N vs Output Power
±20V, Chs 1&3 in Bridge Mode (Note17),
R
L
=8Ω, 80kHz BW
THD+N vs Output Power
±28V, Chs 1&3 in Parallel Mode (Note18),
R
L
=4Ω, 80kHz BW
20067345 20067337
LM4781
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Typical Performance Characteristics (Continued)
THD+N vs Output Power
±24V, All Chs in Parallel Mode (Note18),
R
L
=2Ω, 80kHz BW
Output Power vs Supply Voltage
Chs 1&3 in Bridge Mode (Note17),
f = 1kHz, R
L
=8Ω, 80kHz BW
20067328 20067332
Output Power vs Supply Voltage
Chs 1&3 in Parallel Mode (Note18),
f = 1kHz, R
L
=4Ω, 80kHz BW
Output Power vs Supply Voltage
All Chs in Parallel Mode (Note18),
f = 1kHz, R
L
=2Ω, 80kHz BW
20067324 20067341
Safe Area SpiKe Protection Response
200673C6 200673C7
LM4781
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Typical Performance Characteristics (Continued)
Frequency Response of Demo Board
P
OUT
= 10W/Channel = 0dB, R
IN
= 47kΩ
R
L
=8Ω, No Filters
200673C9
Note 17: Bridge mode graphs were taken using the demo board and inverting the signal to the channel B input.
Note 18: Parallel mode graphs were taken using the demo board connecting each output through a 0.1Ω/3W resistor to the load.
Application Information
MUTE MODE
The muting function allows the user to mute the amplifier.
This can be accomplished as shown in the Typical Applica-
tion Circuit. The resistor R
M
is chosen with reference to the
negative supply voltage and is used in conjunction with a
switch. The switch, when opened or switched to GND, cuts
off the current flow from the MUTE pins to −V
EE
, thus placing
the LM4781 into mute mode. Refer to the Mute Attenuation
vs Mute Current curves in the Typical Performance Char-
acteristics section for values of attenuation per current out
of each MUTE pin. The resistance R
M
is calculated by the
following equation:
R
M
≤(|−V
EE
| − 2.6V) / I
MUTE
Where I
MUTE
≥0.5mA for each MUTE pin.
The MUTE pins can be tied together so that only one resistor
is required for the mute function. The mute resistor value
must be chosen so that a minimum of 1.5mA is pulled
through the resistor R
M
. This ensures that each amplifier is
fully operational. Taking into account supply line fluctuations,
it is a good idea to pull out 1mA per MUTE pin or 3mA total
if all pins are tied together.
A turn-on MUTE or soft start circuit may also be used during
power up. A simple circuit like the one shown below may be
used.
200673B7
The RC combination of C
M
and R
M1
may cause the voltage
at point A to change more slowly than the -V
EE
supply
voltage. Until the voltage at point A is low enough to have
0.5mA of current per MUTE pin flow through R
M2
, the IC will
be in mute mode. The series combination of R
M1
and R
M2
needs to satisfy the mute equation above for all operating
voltages or mute mode may be activated during normal
operation. For a longer turn-on mute time, a larger time
constant, τ=RC=R
M1
C
M
(sec), is needed. For the values
show above and with the MUTE pins tied together, the
LM4781 will enter play mode when the voltage at point A is
-25.1V. The voltage at point A is found with Equation (1)
below.
V
A
(t)=(V
f
-V
O
)e
-t/τ
(Volts) (1)
where:
t = time (sec)
τ= RC (sec)
V
o
= Voltage on C att=0(Volts)
V
f
= Final voltage, -V
EE
in this circuit (Volts)
UNDER-VOLTAGE PROTECTION
Upon system power-up, the under-voltage protection cir-
cuitry allows the power supplies and their corresponding
capacitors to come up close to their full values before turning
on the LM4781. Since the supplies have essentially settled
to their final value, no DC output spikes occur. At power
down, the outputs of the LM4781 are forced to ground before
the power supply voltages fully decay preventing transients
on the output.
OVER-VOLTAGE PROTECTION
The LM4781 contains over-voltage protection circuitry that
limits the output current while also providing voltage clamp-
ing. The clamp does not, however, use internal clamping
diodes. The clamping effect is quite the same because the
output transistors are designed to work alternately by sinking
large current spikes.
SPiKe PROTECTION
The LM4781 is protected from instantaneous peak-
temperature stressing of the power transistor array. The Safe
Operating graph in the Typical Performance Characteris-
LM4781
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Application Information (Continued)
tics section shows the area of device operation where
SPiKe Protection Circuitry is not enabled. The SPiKe Pro-
tection Response waveform graph shows the waveform dis-
tortion when SPiKe is enabled. Please refer to AN-898 for
more detailed information.
THERMAL PROTECTION
The LM4781 has a sophisticated thermal protection scheme
to prevent long-term thermal stress of the device. When the
temperature on the die exceeds 150˚C, the LM4781 shuts
down. It starts operating again when the die temperature
drops to about 145˚C, but if the temperature again begins to
rise, shutdown will occur again above 150˚C. Therefore, the
device is allowed to heat up to a relatively high temperature
if the fault condition is temporary, but a sustained fault will
cause the device to cycle in a Schmitt Trigger fashion be-
tween the thermal shutdown temperature limits of 150˚C and
145˚C. This greatly reduces the stress imposed on the IC by
thermal cycling, which in turn improves its reliability under
sustained fault conditions.
Since the die temperature is directly dependent upon the
heat sink used, the heat sink should be chosen so that
thermal shutdown is not activated during normal operation.
Using the best heat sink possible within the cost and space
constraints of the system will improve the long-term reliability
of any power semiconductor device, as discussed in the
Determining the Correct Heat Sink section.
DETERMlNlNG MAXIMUM POWER DISSIPATION
Power dissipation within the integrated circuit package is a
very important parameter requiring a thorough understand-
ing if optimum power output is to be obtained. An incorrect
maximum power dissipation calculation may result in inad-
equate heat sinking causing thermal shutdown and thus
limiting the output power.
Equation (2) shows the theoretical maximum power dissipa-
tion point for each amplifier in a single-ended configuration
where V
CC
is the total supply voltage.
P
DMAX
=(V
CC
)
2
/2π
2
R
L
(2)
Thus by knowing the total supply voltage and rated output
load, the maximum power dissipation point can be calcu-
lated. The package dissipation is three times the number
which results from Equation (2) since there are three ampli-
fiers in each LM4781. Refer to the graphs of Power Dissipa-
tion versus Output Power in the Typical Performance Char-
acteristics section which show the actual full range of power
dissipation not just the maximum theoretical point that re-
sults from Equation (2).
DETERMINING THE CORRECT HEAT SINK
The choice of a heat sink for a high-power audio amplifier is
made entirely to keep the die temperature at a level such
that the thermal protection circuitry is not activated under
normal circumstances.
The thermal resistance from the die to the outside air, θ
JA
(junction to ambient), is a combination of three thermal re-
sistances, θ
JC
(junction to case), θ
CS
(case to sink), and θ
SA
(sink to ambient). The thermal resistance, θ
JC
(junction to
case), of the LM4781T is 0.9˚C/W. Using Thermalloy Ther-
macote thermal compound, the thermal resistance, θ
CS
(case to sink), is about 0.2˚C/W. Since convection heat flow
(power dissipation) is analogous to current flow, thermal
resistance is analogous to electrical resistance, and tem-
perature drops are analogous to voltage drops, the power
dissipation out of the LM4781 is equal to the following:
P
DMAX
=(T
JMAX
−T
AMB
)/θ
JA
(3)
where T
JMAX
= 150˚C, T
AMB
is the system ambient tempera-
ture and θ
JA
=θ
JC
+θ
CS
+θ
SA
.
200673B8
Once the maximum package power dissipation has been
calculated using Equation 2, the maximum thermal resis-
tance, θ
SA
, (heat sink to ambient) in ˚C/W for a heat sink can
be calculated. This calculation is made using Equation 4
which is derived by solving for θ
SA
in Equation 3.
θ
SA
= [(T
JMAX
−T
AMB
)−P
DMAX
(θ
JC
+θ
CS
)]/P
DMAX
(4)
Again it must be noted that the value of θ
SA
is dependent
upon the system designer’s amplifier requirements. If the
ambient temperature that the audio amplifier is to be working
under is higher than 25˚C, then the thermal resistance for the
heat sink, given all other things are equal, will need to be
smaller.
SUPPLY BYPASSING
The LM4781 has excellent power supply rejection and does
not require a regulated supply. However, to improve system
performance as well as eliminate possible oscillations, the
LM4781 should have its supply leads bypassed with low-
inductance capacitors having short leads that are located
close to the package terminals. Inadequate power supply
bypassing will manifest itself by a low frequency oscillation
known as “motorboating” or by high frequency instabilities.
These instabilities can be eliminated through multiple by-
passing utilizing a large tantalum or electrolytic capacitor
(10µF or larger) which is used to absorb low frequency
variations and a small ceramic capacitor (0.1µF) to prevent
any high frequency feedback through the power supply lines.
If adequate bypassing is not provided, the current in the
supply leads which is a rectified component of the load
current may be fed back into internal circuitry. This signal
causes distortion at high frequencies requiring that the sup-
plies be bypassed at the package terminals with an electro-
lytic capacitor of 470µF or more.
BRIDGED AMPLIFIER APPLICATION
The LM4781 has three operational amplifiers internally, al-
lowing for a few different amplifier configurations. One of
these configurations is referred to as “bridged mode” and
involves driving the load differentially through two of the
LM4781’s outputs. This configuration is shown in Figure 2.
Bridged mode operation is different from the classical single-
ended amplifier configuration where one side of its load is
connected to ground.
A bridge amplifier design has a distinct advantage over the
single-ended configuration, as it provides differential drive to
the load, thus doubling output swing for a specified supply
voltage. Theoretically, four times the output power is pos-
LM4781
www.national.com 16
Application Information (Continued)
sible as compared to a single-ended amplifier under the
same conditions. This increase in attainable output power
assumes that the amplifier is not current limited or clipped.
A direct consequence of the increased power delivered to
the load by a bridge amplifier is an increase in internal power
dissipation. For each operational amplifier in a bridge con-
figuration, the internal power dissipation will increase by a
factor of two over the single ended dissipation. Using Equa-
tion (2) the load impedance should be divided by a factor of
two to find the maximum power dissipation point for each
amplifier in a bridge configuration. In the case of an 8Ωload
in a bridge configuration, the value used for R
L
in Equation
(2) would be 4Ωfor each amplifier in the bridge. When using
two of the amplifiers of the LM4781 in bridge mode, the third
amplifier should have a load impedance equal to or higher
than the equivalent impedance seen by each of the bridged
amplifiers. In the example above where the bridge load is 8Ω
and each amplifier in the bridge sees a load value of 4Ωthen
the third amplifier should also have a 4Ωload impedance or
higher. Using a lower load impedance on the third amplifier
will result in higher power dissipation in the third amplifier
than the other two amplifiers and may result in unwanted
activation of thermal shut down on the third amplifier. Once
the impedance seen by each amplifier is known then Equa-
tion (2) can be used to calculated the value of P
DMAX
for
each amplifier. The P
DMAX
of the IC package is found by
adding up the power dissipation for each amplifier within the
IC package.
This value of P
DMAX
can be used to calculate the correct size
heat sink for a bridged amplifier application. Since the inter-
nal dissipation for a given power supply and load is in-
creased by using bridged-mode, the heatsink’s θ
SA
will have
to decrease accordingly as shown by Equation 4. Refer to
the section, Determining the Correct Heat Sink, for a more
detailed discussion of proper heat sinking for a given appli-
cation.
PARALLEL AMPLIFIER APPLICATION
Parallel configuration is normally used when higher output
current is needed for driving lower impedance loads (i.e. 4Ω
or lower) to obtain higher output power levels. As shown in
Figure 3 , the parallel amplifier configuration consist of de-
signing the amplifiers in the IC to have identical gain, con-
necting the inputs in parallel and then connecting the outputs
in parallel through a small external output resistor. Any num-
ber of amplifiers can be connected in parallel to obtain the
needed output current or to divide the power dissipation
across multiple IC packages. Ideally, each amplifier shares
the output current equally. Due to slight differences in gain
the current sharing will not be equal among all channels. If
current is not shared equally among all channels then the
power dissipation will also not be equal among all channels.
It is recommended that 0.1% tolerance resistors be used to
set the gain (R
i
and R
f
) for a minimal amount of difference in
current sharing.
When operating two or more amplifiers in parallel mode the
impedance seen by each amplifier is equal to the total load
impedance multiplied by the number of amplifiers driving the
load in parallel as shown by Equation (5) below:
R
L(parallel)
=R
L(total)
* Number of amplifiers (5)
Once the impedance seen by each amplifier in the parallel
configuration is known then Equation (2) can be used with
this calculated impedance to find the amount of power dis-
sipation for each amplifier. Total power dissipation (P
DMAX
)
within an IC package is found by adding up the power
dissipation for each amplifier in the IC package. Using the
calculated P
DMAX
the correct heat sink size can be deter-
mined. Refer to the section, Determining the Correct Heat
Sink, for more information and detailed discussion of proper
heat sinking.
If only two amplifiers of the LM4781 are used in parallel
mode then the third amplifier should have a load impedance
equal to or higher than the equivalent impedance seen by
each of the amplifiers in parallel mode. Having the same
load impedance on all amplifiers means that the power
dissipation in each amplifier will be equal. Using a lower load
impedance on the third amplifier will result in higher power
dissipation in the third amplifier than the other two amplifiers
and may result in unwanted activation of thermal shut down
on the third amplifier. Having a higher impedance on the third
amplifier than the equivalent impedance on the two amplifi-
ers in parallel will reduce total IC package power dissipation
reducing the heat sink size requirement.
BI-AMP AND TRI-AMP APPLICATIONS
Bi-amping is the practice of using two different amplifiers to
power the individual drivers in a speaker enclosure. For
example, a two-way speaker enclosure might have a tweeter
and a subwoofer. One amplifier would drive the tweeter and
another would drive the subwoofer. One advantage is that
the gain of each amplifier can be adjusted for the different
driver sensitivities. Another advantage is the crossover can
be designed before the amplifier stages with low cost op
amps instead of large passive components. With the cross-
over before the amplifier stages no power is wasted in the
passive crossover as each individual amplifier provides the
correct frequencies for the driver. Tri-Amping is using three
different amplifier stages in the same way bi-amping is done.
Bi-amping can also be done on a three-way speaker design
by using one amplifier for the subwoofer and another for the
midrange and tweeter.
The LM4781 is perfectly suited for bi-amp or tri-amp appli-
cations with it’s three amplifiers. Two of the amplifiers can be
configured for bridge or parallel mode to drive a subwoofer
with the third amplifier driving the tweeter or tweeter and
midrange. An example would be to use a 4Ωsubwoofer and
8Ωtweeter/midrange with the LM4781 in parallel and single-
ended modes. Each amplifier would see an 8Ωload but the
subwoofer would have twice the output power as the
tweeter/midrange. The gain of each amplifier may also be
adjusted for the desired response. Using the LM4781 in a
tri-amp configuration would allow the gain of each amplifier
to be adjusted to achieve the desired speaker response.
SINGLE-SUPPLY AMPLIFIER APPLICATION
The typical application of the LM4781 is a split supply am-
plifier. But as shown in Figure 4, the LM4781 can also be
used in a single power supply configuration. This involves
using some external components to create a half-supply bias
which is used as the reference for the inputs and outputs.
Thus, the signal will swing around half-supply much like it
swings around ground in a split-supply application. Along
with proper circuit biasing, a few other considerations must
be accounted for to take advantage of all of the LM4781
functions, like the mute function.
CLICKS AND POPS
In the typical application of the LM4781 as a split-supply
audio power amplifier, the IC exhibits excellent “click” and
“pop” performance when utilizing the mute mode. In addition,
LM4781
www.national.com17
Application Information (Continued)
the device employs Under-Voltage Protection, which elimi-
nates unwanted power-up and power-down transients. The
basis for these functions are a stable and constant half-
supply potential. In a split-supply application, ground is the
stable half-supply potential. But in a single-supply applica-
tion, the half-supply needs to charge up at the same rate as
the supply rail, V
CC
. This makes the task of attaining a
clickless and popless turn-on more challenging. Any uneven
charging of the amplifier inputs will result in output clicks and
pops due to the differential input topology of the LM4781.
To achieve a transient free power-up and power-down, the
voltage seen at the input terminals should be ideally the
same. Such a signal will be common-mode in nature, and
will be rejected by the LM4781. In Figure 4, the resistor R
INP
serves to keep the inputs at the same potential by limiting the
voltage difference possible between the two nodes. This
should significantly reduce any type of turn-on pop, due to an
uneven charging of the amplifier inputs. This charging is
based on a specific application loading and thus, the system
designer may need to adjust these values for optimal perfor-
mance.
As shown in Figure 4, the resistors labeled R
BI
help bias up
the LM4781 off the half-supply node at the emitter of the
2N3904. But due to the input and output coupling capacitors
in the circuit, along with the negative feedback, there are two
different values of R
BI
, namely 10kΩand 200kΩ. These
resistors bring up the inputs at the same rate resulting in a
popless turn-on. Adjusting these resistors values slightly
may reduce pops resulting from power supplies that ramp
extremely quick or exhibit overshoot during system turn-on.
PROPER SELECTION OF EXTERNAL COMPONENTS
Proper selection of external components is required to meet
the design targets of an application. The choice of external
component values that will affect gain and low frequency
response are discussed below.
The gain of each amplifier is set by resistors R
f
and R
i
for the
non-inverting configuration shown in Figure 1. The gain is
found by Equation (6) below:
A
V
=1+R
f
/R
i
(V/V) (6)
For best noise performance, lower values of resistors are
used. A value of 1kΩis commonly used for R
i
and then
setting the value of R
f
for the desired gain. For the LM4781
the gain should be set no lower than 10V/V and no higher
than 50V/V. Gain settings below 10V/V may experience
instability and using the LM4781 for gains higher than 50V/V
will see an increase in noise and THD.
The combination of R
i
with C
i
(see Figure 1) creates a high
pass filter. The low frequency response is determined by
these two components. The -3dB point can be found from
Equation (7) shown below:
f
i
=1/(2πR
i
C
i
) (Hz) (7)
If an input coupling capacitor is used to block DC from the
inputs as shown in Figure 5, there will be another high pass
filter created with the combination of C
IN
and R
IN
. When
using a input coupling capacitor R
IN
is needed to set the DC
bias point on the amplifier’s input terminal. The resulting
-3dB frequency response due to the combination of C
IN
and
R
IN
can be found from Equation (8) shown below:
f
IN
=1/(2πR
IN
C
IN
) (Hz) (8)
With large values of R
IN
oscillations may be observed on the
outputs when the inputs are left floating. Decreasing the
value of R
IN
or not letting the inputs float will remove the
oscillations. If the value of R
IN
is decreased then the value of
C
IN
will need to increase in order to maintain the same -3dB
frequency response.
HIGH PERFORMANCE CONSIDERATIONS
Using low cost electrolytic capacitors in the signal path such
as C
IN
and C
i
(see Figures1-5)will result in very good
performance. However, electrolytic capacitors are less linear
than other premium capacitors. Higher THD+N performance
may be obtained by using high quality polypropylene capaci-
tors in the signal path. A more cost effective solution may be
the use of smaller value premium capacitors in parallel with
the larger electrolytic capacitors. This will maintain signal
quality in the upper audio band where any degradation is
most noticeable while also coupling in the signals in the
lower audio band for good bass response.
Distortion is introduced as the audio signal approaches the
lower -3dB point, determined as discussed in the section
above. By using larger values of capacitors such that the
-3dB point is well outside of the audio band will reduce this
distortion and improve THD+N performance.
Increasing the value of the large supply bypass capacitors
will improve burst power output. The larger the supply by-
pass capacitors the higher the output pulse current without
supply droop increasing the peak output power. This will also
increase the headroom of the amplifier and reduce THD.
SIGNAL-TO-NOISE RATIO
In the measurement of the signal-to-noise ratio, misinterpre-
tations of the numbers actually measured are common. One
amplifier may sound much quieter than another, but due to
improper testing techniques, they appear equal in measure-
ments. This is often the case when comparing integrated
circuit designs to discrete amplifier designs. Discrete transis-
tor amps often “run out of gain” at high frequencies and
therefore have small bandwidths to noise as indicated below.
20067399
Integrated circuits have additional open loop gain allowing
additional feedback loop gain in order to lower harmonic
distortion and improve frequency response. It is this addi-
tional bandwidth that can lead to erroneous signal-to-noise
measurements if not considered during the measurement
process. In the typical example above, the difference in
bandwidth appears small on a log scale but the factor of 10in
bandwidth, (200kHz to 2MHz) can result in a 10dB theoreti-
cal difference in the signal-to-noise ratio (white noise is
proportional to the square root of the bandwidth in a system).
In comparing audio amplifiers it is necessary to measure the
magnitude of noise in the audible bandwidth by using a
“weighting” filter (Note 19). A “weighting” filter alters the
frequency response in order to compensate for the average
human ear’s sensitivity to the frequency spectra. The weight-
ing filters at the same time provide the bandwidth limiting as
discussed in the previous paragraph.
LM4781
www.national.com 18
Application Information (Continued)
Note 19: CCIR/ARM: A Practical Noise Measurement Method; by Ray
Dolby, David Robinson and Kenneth Gundry, AES Preprint No. 1353 (F-3).
In addition to noise filtering, differing meter types give differ-
ent noise readings. Meter responses include:
1. RMS reading,
2. average responding,
3. peak reading, and
4. quasi peak reading.
Although theoretical noise analysis is derived using true
RMS based calculations, most actual measurements are
taken with ARM (Average Responding Meter) test equip-
ment.
Typical signal-to-noise figures are listed for an A-weighted
filter which is commonly used in the measurement of noise.
The shape of all weighting filters is similar, with the peak of
the curve usually occurring in the 3kHz–7kHz region.
LEAD INDUCTANCE
Power op amps are sensitive to inductance in the output
leads, particularly with heavy capacitive loading. Feedback
to the input should be taken directly from the output terminal,
minimizing common inductance with the load.
Lead inductance can also cause voltage surges on the sup-
plies. With long leads to the power supply, energy is stored in
the lead inductance when the output is shorted. This energy
can be dumped back into the supply bypass capacitors when
the short is removed. The magnitude of this transient is
reduced by increasing the size of the bypass capacitor near
the IC. With at least a 20µF local bypass, these voltage
surges are important only if the lead length exceeds a couple
feet (>1µH lead inductance). Twisting together the supply
and ground leads minimizes the effect.
PHYSICAL IC MOUNTING CONSIDERATIONS
Mounting of the package to a heat sink must be done such
that there is sufficient pressure from the mounting screws to
insure good contact with the heat sink for efficient heat flow.
Over tightening the mounting screws will cause the package
to warp reducing contact area with the heat sink. Less
contact with the heat sink will increase the thermal resis-
tance from the package case to the heat sink (θ
CS
) resulting
in higher operating die temperatures and possible unwanted
thermal shut down activation. Extreme over tightening of the
mounting screws will cause severe physical stress resulting
in cracked die and catastrophic IC failure. The recom-
mended mounting screw size is M3 with a maximum torque
of 50 N-cm. Additionally, it is best to use washers under the
screws to distribute the force over a wider area or a screw
with a wide flat head. To further distribute the mounting force
a solid mounting bar in front of the package and secured in
place with the two mounting screws may be used. Other
mounting options include a spring clip. If the package is
secured with pressure on the front of the package the maxi-
mum pressure on the molded plastic should not exceed
150N/mm
2
.
Additionally, if the mounting screws are used to force the
package into correct alignment with the heat sink, package
stress will be increased. This increase in package stress will
result in reduced contact area with the heat sink increasing
die operating temperature and possible catastrophic IC fail-
ure.
LM4781
www.national.com19
Application Information (Continued)
LAYOUT, GROUND LOOPS AND STABILITY
The LM4781 is designed to be stable when operated at a
closed-loop gain of 10 or greater, but as with any other
high-current amplifier, the LM4781 can be made to oscillate
under certain conditions. These oscillations usually involve
printed circuit board layout or output/input coupling issues.
When designing a layout, it is important to return the load
ground, the output compensation ground, and the low level
(feedback and input) grounds to the circuit board common
ground point through separate paths. Otherwise, large cur-
rents flowing along a ground conductor will generate volt-
ages on the conductor which can effectively act as signals at
the input, resulting in high frequency oscillation or excessive
distortion. It is advisable to keep the output compensation
components and the 0.1µF supply decoupling capacitors as
close as possible to the LM4781 to reduce the effects of PCB
trace resistance and inductance. For the same reason, the
ground return paths should be as short as possible.
In general, with fast, high-current circuitry, all sorts of prob-
lems can arise from improper grounding which again can be
avoided by returning all grounds separately to a common
point. Without isolating the ground signals and returning the
grounds to a common point, ground loops may occur.
“Ground Loop” is the term used to describe situations occur-
ring in ground systems where a difference in potential exists
between two ground points. Ideally a ground is a ground, but
unfortunately, in order for this to be true, ground conductors
with zero resistance are necessary. Since real world ground
leads possess finite resistance, currents running through
them will cause finite voltage drops to exist. If two ground
return lines tie into the same path at different points there will
be a voltage drop between them. The first figure below
shows a common ground example where the positive input
ground and the load ground are returned to the supply
ground point via the same wire. The addition of the finite wire
resistance, R
2
, results in a voltage difference between the
two points as shown below.
20067398
The load current I
L
will be much larger than input bias current
I
I
, thus V
1
will follow the output voltage directly, i.e. in phase.
Therefore the voltage appearing at the non-inverting input is
effectively positive feedback and the circuit may oscillate. If
there was only one device to worry about then the values of
R
1
and R
2
would probably be small enough to be ignored;
however, several devices normally comprise a total system.
Any ground return of a separate device, whose output is in
phase, can feedback in a similar manner and cause insta-
bilities. Out of phase ground loops also are troublesome,
causing unexpected gain and phase errors.
The solution to most ground loop problems is to always use
a single-point ground system, although this is sometimes
impractical. The third figure above is an example of a single-
point ground system.
The single-point ground concept should be applied rigor-
ously to all components and all circuits when possible. Vio-
lations of single-point grounding are most common among
printed circuit board designs, since the circuit is surrounded
by large ground areas which invite the temptation to run a
device to the closest ground spot. As a final rule, make all
ground returns low resistance and low inductance by using
large wire and wide traces.
Occasionally, current in the output leads (which function as
antennas) can be coupled through the air to the amplifier
input, resulting in high-frequency oscillation. This normally
happens when the source impedance is high or the input
leads are long. The problem can be eliminated by placing a
small capacitor, C
C
, (on the order of 50pF to 500pF) across
the LM4781 input terminals. Refer to the External Compo-
nents Description section relating to component interaction
with C
f
.
REACTIVE LOADING
It is hard for most power amplifiers to drive highly capacitive
loads very effectively and normally results in oscillations or
ringing on the square wave response. If the output of the
LM4781 is connected directly to a capacitor with no series
resistance, the square wave response will exhibit ringing if
the capacitance is greater than about 0.2µF. If highly capaci-
tive loads are expected due to long speaker cables, a
method commonly employed to protect amplifiers from low
impedances at high frequencies is to couple to the load
through a 10Ωresistor in parallel with a 0.7µH inductor. The
inductor-resistor combination as shown in the Figure 5 iso-
lates the feedback amplifier from the load by providing high
output impedance at high frequencies thus allowing the 10Ω
resistor to decouple the capacitive load and reduce the Q of
the series resonant circuit. The LR combination also pro-
vides low output impedance at low frequencies thus shorting
out the 10Ωresistor and allowing the amplifier to drive the
series RC load (large capacitive load due to long speaker
cables) directly.
INVERTING AMPLIFIER APPLICATION
The inverting amplifier configuration may be used instead of
the more common non-inverting amplifier configuration
shown in Figure 1. The inverting amplifier can have better
THD+N performance and eliminates the need for a large
capacitor (Ci) reducing cost and space requirements. The
values show in Figure 6 are only one example of an amplifier
with a gain of 20V/V (Gain = -R
f
/R
i
). For different resistor
values, the value of R
B
should be eqaul to the parallel
combination of R
f
and Ri.
LM4781
www.national.com 20
Application Information (Continued)
If the DC blocking input capacitor (C
IN
) is used as shown, the
lower -3dB point is found using Equation (8) as discussed in
the Proper Selection of External Components section.
200673B9
FIGURE 6. Inverting Amplifier Application Circuit
LM4781
www.national.com21
Application Information (Continued)
200673C8
FIGURE 7. Reference PCB Schematic
LM4781
www.national.com 22
Application Information (Continued)
LM4781 REFERENCE BOARD ARTWORK
200673A9
Composite Layer
200673B0
Silk Layer
200673A8
Top Layer
200673B1
Bottom Layer
LM4781
www.national.com23
Application Information (Continued)
Bill Of Materials for Reference PCB
Symbol Value Tolerance Type/Description Comment
R
IN1
,R
IN2
,R
IN3
33kΩ5% 1/4 Watt
R
B1
,R
B2
,R
B3
1kΩ1% 1/4 Watt
R
F1
,R
F2
,R
F3
20kΩ1% 1/4 Watt
R
i1
,R
i2
,R
i3
1kΩ1% 1/4 Watt
R
SN1
,R
SN2
,R
SN3
4.7Ω5% 1/4 Watt
R
G
2.7Ω5% 1/4 Watt
R
M
5.6kΩ5% 1/4 Watt
C
IN1
,C
IN2
,C
IN3
1µF 10% Metallized Polyester Film
C
i1
,C
i2
,C
i3
47µF 20% Electrolytic Radial / 50V
C
SN1
,C
SN2
,C
SN3
0.1µF 20% Monolithic Ceramic
C
S1
,C
S2
0.1µF 20% Monolithic Ceramic
C
S3
,C
S4
10µF 20% Electrolytic Radial / 50V
C
S5
,C
S6
2,200µF 20% Electrolytic Radial / 50V
S
1
SPDT (on-on) Switch
J
1
,J
2
,J
3
Non-Switched PC Mount RCA
Jack
J
5
,J
7
,J
9
,J
11
PCB Banana Jack - BLACK
J
4
,J
6
,J
8
,J
10
,J
12
PCB Banana Jack - RED
U
1
27 lead TO-220 Power Socket
with push release lever or
LM4781 IC
LM4781
www.national.com 24
Physical Dimensions inches (millimeters)
unless otherwise noted
Non-Isolated TO-220 27-Lead Package
Order Number LM4781TA
NS Package Number TA27A
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whose failure to perform when properly used in
accordance with instructions for use provided in the
labeling, can be reasonably expected to result in a
significant injury to the user.
2. A critical component is any component of a life
support device or system whose failure to perform
can be reasonably expected to cause the failure of
the life support device or system, or to affect its
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LM4781 Overture
™
Audio Power Amplifier Series
3 Channel 35W Audio Power Amplifier with Mute
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.
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