LM4962 www.ti.com SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 LM4962 Ceramic Speaker Driver Check for Samples: LM4962 FEATURES DESCRIPTION * The LM4962 is an audio power amplifier primarily designed for driving Ceramic Speaker for applications in Cell Phones, Smart Phones, PDA's and other portable applications. It is capable of driving 15Vpp (typ) BTL with less than 1% THD+N from a 3.2VDC power supply. The LM4962 features and low power consumption shutdown mode, an internal thermal shutdown protection mechanism, along with over current protection (OCP) and over voltage protection (OVP). 1 2 * * * * * * * * * * * Click and Pop Circuitry Eliminates Noise During Turn-On and Turn-Off Transitions Low Current Shutdown Mode Low Quiescent Current Mono 15Vp-p BTL Output, RL = 2F+9.4, f = 1kHz, 1% THD+N Over-Current Protection Over-Voltage Protection Unity-Gain Stable External Gain Configuration Capability Including Band Switch Function Leakage Cut Switch (SW-LEAK) Soft-Start Function Space-Saving DSBGA Package (2mm x 2.5mm) APPLICATIONS * * * * * Smart Phones Mobile Phones and Multimedia Terminals PDA's, Internet Appliances, and Portable Gaming Portable DVD Digital Still Cameras/Camcorders Boomer audio power amplifiers were designed specifically to provide high quality output power with a minimal number of external components. The LM4962 does not require bootstrap capacitors, or snubber circuits. The LM4962 also features a Band-Switch function which allows the user to use one amplifier device for both receiver (earpiece) mode and ringer/loudspeaker mode. The LM4962 contains advanced click and pop circuitry that eliminates noises which would otherwise occur during turn-on and turn-off transitions. Additionally, the internal boost converter features a soft-start function. The LM4962 is unity-gain stable and can be configured by external gain-setting resistors. KEY SPECIFICATIONS * * * * Quiescent Power Supply Current (Boost Converter + Amplifier): 9 mA (typ) Voltage Swing in BTL at 1% THD, f = 1kHz: 15 Vp-p (typ) Shutdown Current: 0.1 A (typ) OVP: 8.5V < VAMP < 9.5 V 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright (c) 2005-2013, Texas Instruments Incorporated LM4962 SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 www.ti.com Connection Diagram Top View 1 2 3 4 A SDAmp SDBoost GND (SW) SW B SoftStart Flagout VDD BootStrap C Bypass CCHG VIN OC/OV Detect GND BW2 BW1 FB VO1 VAMP VO2 SWLeak D E Figure 1. DSBGA Package See Package Number YZR002011A 2 Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 LM4962 www.ti.com SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 Typical Application L1 10 PH D2 V1 = 1.2(1+R2/R5) Vdd R3 1.6k Cs1 4.7 PF Vdd SW R2 25k C3 100p C2 4.7 PF FB GND (sw) R5 4.9k Soft-Start Css 10 nF SW-LEAK Bootstrap Flagout Flagout OC/OV Detect Rs 100m Vamp Cs2 4.7 PF Shutdown 1 SD Boost Shutdown 2 SD Amp GND To LM4951 for stereo solution Bypass Cb1 1.0 PF Ro1 4.7 Vo2 Rchg1 1k Ceramic Speaker 2.0 PF Cchg Ro2 4.7 CinA 0.1 PF Vo1 Vin Load RinA 20k BW1 BW2 Rf1 200k Rf2 20k Cf1 82 pF Figure 2. Typical Audio Amplifier Application Circuit Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 3 LM4962 SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 www.ti.com Block Diagram VDD L1 D2 Cs1 C2 R3 R2 C3 SW VDD FB Driver + R5 V BG PWM SW-LEAK GND(SW) GND LM4962 Rs Vamp Cs2 CinA RinA VIN Rchg1 BW1 Rf1 BW2 Rf2 Cchg Cf1 Flagout flagout Cf2 - Vo1 Soft-Start Css Bypass BIAS, SHUTDOWN, and PROTECTION CIRCUITRY + Ro1 Cb1 V IH SD Amp V IL Shutdown control V IH V IL Shutdown control Ro2 + Vo2 - SD Boost OC/OV Detect GND Figure 3. LM4962 Block Diagram These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 4 Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 LM4962 www.ti.com SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 Absolute Maximum Ratings (1) (2) (3) Supply Voltage (VDD) 9.5V Amplifier Supply Voltage (VAMP) 9.5V -65C to +150C Storage Temperature -0.3V to VDD + 0.3V Input Voltage (4) Internally limited ESD Susceptibility (5) 2000V ESD Susceptibility (6) 200V Power Dissipation Junction Temperature Thermal Resistance (1) (2) (3) (4) (5) (6) (7) 150C JA (DSBGA) (7) 73C/W All voltages are measured with respect to the GND pin, unless otherwise specified. Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which ensure specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not ensured for parameters where no limit is given, however, the typical value is a good indication of device performance. If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and specifications. The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, JA, and the ambient temperature, TA. The maximum allowable power dissipation is PDMAX = (TJMAX - TA) / JA or the given in Absolute Maximum Ratings, whichever is lower. Human body model, 100pF discharged through a 1.5k resistor. Machine Model, 220pF-240pF discharged through all pins. The value for a JA is measured with the LM4962 mounted on a 3" x 1.5" 4 layer board. The copper thickness for all 4 layers is 0.5oz (roughly 0.18mm). Operating Ratings Temperature Range (TMIN TA TMAX) (1) -40C TA +85C Supply Voltage (VDD) 3.0V < VDD < 5.0V Amplifier Supply Voltage (V1) (2) (1) (2) 2.7V < VAMP < 9.0V Temperature range is tentative, pending characterization. An amplifier supply voltage of 9.0V can only be obtained when the over current and over voltage protection circuitry is disabled (OV/OC Detect pin is disabled). Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 5 LM4962 SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 www.ti.com Electrical Characteristics The following specifications apply for VDD = 3.2V, AV-BTL = 26dB, ZL = 2F+9.4, Cb = 1.0F, R2 = 25K, R5 = 4.9K unless otherwise specified. Limits apply for TA = 25C. Parameter LM4962 Test Conditions Typ (1) Limit (2) (3) Units (Limits) IDD Quiescent Power Supply Current in Boosted Ringer Mode VIN = 0V, 9 12 mA (max) Iddrcv Quiescent Power Supply Current in Receiver Mode SD Boost = GND SD Amp = VDD 3 5 mA (max) ISD Shutdown Current (4) SD Boost = SD Amp = GND 0.1 2.0 A (max) VLH Logic High Threshold Voltage For SD Boost, SD Amp 1.2 V (min) VLL Logic Low Threshold Voltage For SD Boost, SD Amp 0.4 V (max) RPULLDOWN Pulldown Resistor For SD Amp, SD Boost 80 60 k (min) TWUBC Boost Converter Wake-up Time CSS = 10nF 2 5 ms (max) TWUA Audio Amplifier Wake-up Time (For Vdd = 2.7V to 8.5V) 20 40 msec VOUT Output Voltage Swing THD = 1% (max), f = 1kHz 15 14 Vpp (min) THD+N Total Harmonic Distortion + Noise Vout = 14Vpp, f = 1kHz 0.4 1.0 OS Output Noise A-Weighted Filter, VIN = 0V 125 PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 100Hz, Input Referred 86 71 dB (min) Ron-sw-leak On Resistance on SW-Leak SD Boost = GND Isink = 100A 30 50 (max) Ron Flagout On resistance Isink = 1mA 50 100 (max) Vovp Sensitivity of Over Voltage Protection Flagout = GND on VAMP 9.0 9.5 8.5 V (max) V (min) Vocp Sensitivity of Over Current Protection Flagout = GND (Voltage Across RS) 185 275 75 mV (max) mV (min) Ileak Leak Current on Flagout pin 2 A (max) ISW SW Current Limit 2 2.7 A (max) 1.2 A (min) 150 C (min) TSD Thermal Shutdown Temperature Vos Output Offset Voltage VFB Feedback Voltage (1) (2) (3) (4) 6 Vflagout = VDD SD Boost = VDD SD Amp = VDD % V 5 25 mV 1.23 1.15 1.31 V (min) V (max) Typicals are measured at 25C and represent the parametric norm. Limits are specified to AOQL (Average Outgoing Quality Level). Datasheet min/max specification limits are specified by design, test, or statistical analysis. Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to Vin for minimum shutdown current. Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 LM4962 www.ti.com SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 Typical Performance Characteristics THD+N vs Frequency VDD = 3.2V, VO = 4.95VRMS, ZL = 2F+9.4 THD+N vs Frequency VDD = 4.2V, VO = 4.95VRMS, ZL = 2F+9.4 10 10 5 5 2 2 1 THD+N (%) THD+N (%) 1 0.5 0.2 0.1 0.5 0.2 0.1 0.05 0.05 0.02 0.02 0.01 100 300 1k 10k 4k 2k 0.01 100 300 FREQUENCY (Hz) 4k 2k 1k 10k FREQUENCY (Hz) Figure 4. Figure 5. THD+N vs Frequency VDD = 5V, VO = 4.95VRMS, ZL = 2F+9.4 THD+N vs Output Voltage Swing = 3.2V, ZL = 2F+9.4, f = 1kHz VDD 10 10 5 5 f = 1 kHz f = 10 kHz 2 2 1 THD+N (%) THD+N (%) 1 0.5 0.2 0.5 0.2 0.1 f = 100 Hz 0.05 0.1 0.02 0.05 0.01 20m 50m 0.02 0.01 100 200m 500m 1 2 5 10 OUTPUT VOLTAGE SWING (Vrms) 300 1k 4k 2k 10k FREQUENCY (Hz) 10 5 Figure 6. Figure 7. THD+N vs Output Voltage Swing VDD = 4.2V, ZL = 2F+9.4, f = 1kHz THD+N vs Output Voltage Swing VDD = 5V, ZL = 2F+9.4, f = 1kHz 10 5 f = 1 kHz 2 2 1 1 f = 1 kHz THD+N (%) THD+N (%) f = 10 kHz 0.5 0.2 f = 10 kHz 0.1 0.05 0.5 0.2 0.1 0.05 f = 100 Hz 0.02 f = 100 Hz 0.02 0.01 20m 50m 200m 500m 1 2 5 10 0.01 20m 50m OUTPUT VOLTAGE SWING (Vrms) 200m 500m 1 2 5 10 OUTPUT VOLTAGE SWING (Vrms) Figure 8. Figure 9. Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 7 LM4962 SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 www.ti.com Typical Performance Characteristics (continued) PSRR vs Frequency VDD = 3.2, ZL = 2F+9.4, VRIPPLE = 200mVP-P PSRR vs Frequency VDD = 4.2, ZL = 2F+9.4, VRIPPLE = 200mVP-P 0 -10 -10 -20 -20 -30 -30 -40 -40 PSRR (dB) PSRR (dB) 0 -50 -60 -50 -60 -70 -70 -80 -80 -90 -90 -100 -100 20 1k 100 5k 20k 20 5k 20k FREQUENCY (Hz) Figure 10. Figure 11. PSRR vs Frequency VDD = 5, ZL = 2F+9.4, VRIPPLE = 200mVP-P Frequency Response vs Input Capacitor Size 0 20 16 -20 12 OUTPUT LEVEL (dB) -10 -30 PSRR (dB) 1k 100 FREQUENCY (Hz) -40 -50 -60 -70 -80 Ci = 1.0 PF 8 4 0 -4 -8 -12 Ci = 0.039 PF -16 Ci = 0.39 PF -20 -90 -24 -28 -100 20 100 1k 5k 20k 20 50 100 200 500 1k 2k 5k 10k 20k FREQUENCY (Hz) FREQUENCY (Hz) Figure 12. Figure 13. Boost Efficiency vs Output Voltage Swing f = 1kHz, ZL = 2F+9.4 Inductor Current vs Output Voltage Swing f = 1kHz, ZL = 2F+9.4 100 200 VDD = 5V INDUCTOR CURRENT (mA) BOOST EFFICIENCY (%) 95 90 VDD = 3V 85 80 VDD = 4.2V 75 70 0 1 2 3 150 VDD = 4.2V 100 50 VDD = 5V 0 4 5 6 0 1 2 3 4 5 6 OUTPUT VOLTAGE SWING (Vrms) OUTPUT VOLTAGE SWING (Vrms) Figure 14. 8 VDD = 3V Figure 15. Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 LM4962 www.ti.com SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 Typical Performance Characteristics (continued) 12 Supply Current vs Supply Voltage Feedback Voltage vs Temperature 1.26 1.25 FEEDBACK VOLTAGE (V) SUPPLY CURRENT (mA) 10 8 6 4 2 0 2.5 3.5 3.0 4.0 5.0 4.5 1.24 1.23 1.22 1.21 1.20 1.19 -60 -40 5.5 -20 SUPPLY VOLTAGE (V) 20 40 80 100 60 TEMPERATURE (C) Figure 16. Figure 17. VOCP vs Vamp 250 0 1200 Rds(on) vs VBOOTSTRAP 1000 200 Rds(on) (m:) VOCP (V) 800 150 100 50 600 400 200 0 2 4 6 8 10 0 2.5 3.5 4.5 5.5 6.5 7.5 8.5 BOOTSTRAP VOLTAGE (V) VAMP (V) Figure 18. Figure 19. Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 9 LM4962 SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 www.ti.com APPLICATION INFORMATION BRIDGE CONFIGURATION EXPLANATION The Audio Amplifier portion of the LM4962 has two internal amplifiers allowing different amplifier configurations. The first amplifier's gain is externally configurable, whereas the second amplifier is internally fixed in a unity-gain, inverting configuration. The closed-loop gain of the first amplifier is set by selecting the ratio of Rf to Ri while the second amplifier's gain is fixed by the two internal 20k resistors. Figure 2 shows that the output of amplifier one serves as the input to amplifier two. This results in both amplifiers producing signals identical in magnitude, but out of phase by 180. Consequently, the differential gain for the Audio Amplifier is AVD = 2 *(Rf/Ri) (1) By driving the load differentially through outputs Vo1 and Vo2, an amplifier configuration commonly referred to as "bridged mode" is established. Bridged mode operation is different from the classic single-ended amplifier configuration where one side of the load is connected to ground. A bridge amplifier design has a few distinct advantages over the single-ended configuration. It provides differential drive to the load, thus doubling the output swing for a specified supply voltage. BOOST CONVERTER POWER DISSIPATION At higher duty cycles, the increased ON-time of the switch FET means the maximum output current will be determined by power dissipation within the LM4962 FET switch. The switch power dissipation from ON-time conduction is calculated by Equation (2). PD(SWITCH) = DC x IIND(AVE)2 x RDS(ON) (2) where: DC is the duty cycle. There will be some switching losses as well, so some derating needs to be applied when calculating IC power dissipation. MAXIMUM AMPLIFIER POWER DISSIPATION Power dissipation is a major concern when designing a successful amplifier, whether the amplifier is bridged or single-ended. A direct consequence of the increased power delivered to the load by a bridge amplifier is an increase in internal power dissipation. Since the amplifier portion of the LM4962 has two operational amplifiers, the maximum internal power dissipation is 4 times that of a single-ended amplifier. The maximum power dissipation for a given BTL application can be derived from Equation (3). PDMAX(AMP) = (2VDD 2) / (2RL) (3) where: RL = Ro1 + Ro2 MAXIMUM TOTAL POWER DISSIPATION The total power dissipation for the LM4962 can be calculated by adding Equation (2) and Equation (3) together to establish Equation (4): PDMAX(TOTAL) = (2VDD 2) / (2EFF2RL) (4) where: EFF = Efficiency of boost converter RL = Ro1 + Ro2 The result from Equation (4) must not be greater than the power dissipation that results from Equation (5): PDMAX = (TJMAX - TA) / JA 10 (5) Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 LM4962 www.ti.com SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 For the LQA28A, JA = 73C/W. TJMAX = 125C for the LM4962. Depending on the ambient temperature, TA, of the system surroundings, Equation (5) can be used to find the maximum internal power dissipation supported by the IC packaging. If the result of Equation (4) is greater than that of Equation (5), then either the supply voltage must be increased, the load impedance increased or TA reduced. For typical applications, power dissipation is not an issue. Power dissipation is a function of output power and thus, if typical operation is not around the maximum power dissipation point, the ambient temperature may be increased accordingly. START-UP SEQUENCE For the LM4962 correct start-up sequencing is important for optimal device performance. Using the correct start up sequence will improve click/pop performance as well as avoid transients that could reduce battery life. For ringer/loudspeaker mode, the supply voltage should be applied first and both the boost converter and the amplifier should be in shutdown. The boost converter can then be activated followed by the amplifier (see timing diagram, Figure 20). If the boost converter shutdown is toggled while the amplifier is active a very audible pop will be heard. SHUTDOWN FUNCTION In many applications, a microcontroller or microprocessor output is used to control the shutdown circuitry to provide a quick, smooth transition into shutdown. Another solution is to use a single-pole, single-throw switch connected between VDD and Shutdown pins. BAND SWITCH FUNCTION The LM4962 features a Band Switch function which allows the user to use one amplifier for both receiver (earpiece) mode and ringer/loudspeaker mode. When the boost converter and the amplifier are both active the device is is in ringer mode. This enables the boost converter and sets the externally configurable closed loop gain selection to BW1. If the boost converter is in the shutdown and the amplifier is active the device is in receiver mode. In this mode the gain selection is switched to BW2. This allows the amplifier to be powered directly from the battery minus the voltage drop across the Schottky diode. SD Boost SD Amp Receiver Mode (BW2) Low High Boosted Ringer Mode (BW1) High High Shutdown Low Low BOOTSTRAP PIN The bootstrap pin, featured in the LM4962, provides a voltage supply for the internal switch driver. Connecting the bootstrap pin to V1 (See Figure 2) allows for a higher voltage to drive the gate of the switch thereby reducing the Ron. This configuration is necessary in applications with heavier loads. The bootstrap pin can be connected to VDD when driving lighter loads to improve device performance (Iddq, THD+N, Noise, etc.). Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 11 LM4962 SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 www.ti.com Vdd Battery Voltage 0V Vdd Boost Converter Shutdown 0V Vdd Amplifier Shutdown 0V V1 V1 (output of Boost Converter) Vdd-Vd* Vamp/2 Vo1/Vo2 (Amplifier DC Bias) (Vdd-Vd)/2* 0V Ringer Mode Receiver Mode *Vd = Voltage drop across diode D2 Figure 20. Power on Sequence Timing Diagram OVER-CURRENT AND OVER-VOLTAGE PROTECTION FUNCTION Flagout Pin The Flagout pin indicates a fault when an over current or over voltage condition has been detected. The Flagout pin is high impedance when inactive. When active, the Flagout pin is pulled down to a 50 short to GND. Over-Voltage Protection (OVP) Operation When a voltage (Vamp) greater than 8.5V (min) is detected at the OC/OV Detect pin, the LM4962 indicates a fault by activating the Flagout pin. The boost converter momentarily shutdown and reinitialize the soft-start sequence. The Flagout pin will remain active until both shutdowns pins are pulled low. Over-Current Protection (OCP) Operation The OCP circuitry monitors the voltage across Rocd to detect the output current of the boost converter. If a voltage greater than 185mV (typ) is detected the device will shutdown and the Flagout pin will be activated. For the device to return to normal operation both shutdown pins need to be pulled low to reset the Flagout pin. Disable OCP The Over-Current Protection Circuitry can be disabled by shorting out RS. In this configuration the OVP circuitry is still active. Disable both OVP and OCP Both features can be disabled by grounding the OC/OV Detect pin. In this configuration the Flagout pin will be inactive. 12 Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 LM4962 www.ti.com SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 Timing Diagrams Vdd Amplifier Shutdown 0V Vdd Boost Shutdown 0V 185 mV Vscd * 0V Vdd Flagout 0V On Internal Boost Operation Off On Internal Amplifier Operation Off t1 < 3 Ps t1 * Vscd refers to the voltage differential across Rs Figure 21. OCP Timing Diagram Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 13 LM4962 SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 www.ti.com Vdd Amplifier Shutdown 0V Vdd Boost Shutdown 0V 8.5V 7.5V Vdd Vamp 0V Vdd Flagout 0V On Internal Boost Operation Off On Internal Amplifier Operation Off t3 t2 t1 t3 t2 t3 t1 t1 <3 Ps t2 = 16 ms t3 = 3 ms Figure 22. OVP Timing Diagram PROPER SELECTION OF EXTERNAL COMPONENTS Proper selection of external components in applications using integrated power amplifiers, and switching DC-DC converters, is critical for optimizing device and system performance. Consideration to component values must be used to maximize overall system quality. The best capacitors for use with the switching converter portion of the LM4962 are multi-layer ceramic capacitors. They have the lowest ESR (equivalent series resistance) and highest resonance frequency, which makes them optimum for high frequency switching converters. When selecting a ceramic capacitor, only X5R and X7R dielectric types should be used. Other types such as Z5U and Y5F have such severe loss of capacitance due to effects of temperature variation and applied voltage, they may provide as little as 20% of rated capacitance in many typical applications. Always consult capacitor manufacturer's data curves before selecting a capacitor. High-quality ceramic capacitors can be obtained from Taiyo-Yuden. 14 Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 LM4962 www.ti.com SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 POWER SUPPLY BYPASSING As with any amplifier, proper supply bypassing is critical for low noise performance and high power supply rejection. The capacitor location on both V1 and VDD pins should be as close to the device as possible. SELECTING INPUT CAPACITOR FOR AUDIO AMPLIFIER One of the major considerations is the closedloop bandwidth of the amplifier. To a large extent, the bandwidth is dictated by the choice of external components shown in Figure 2. The input coupling capacitor, Ci, forms a first order high pass filter which limits low frequency response. This value should be chosen based on needed frequency response for a few distinct reasons. High value input capacitors are both expensive and space hungry in portable designs. Clearly, a certain value capacitor is needed to couple in low frequencies without severe attenuation. But ceramic speakers used in portable systems, whether internal or external, have little ability to reproduce signals below 100Hz to 150Hz. Thus, using a high value input capacitor may not increase actual system performance. In addition to system cost and size, click and pop performance is affected by the value of the input coupling capacitor, Ci. A high value input coupling capacitor requires more charge to reach its quiescent DC voltage (nominally 1/2 VDD). This charge comes from the output via the feedback and is apt to create pops upon device enable. Thus, by minimizing the capacitor value based on desired low frequency response, turn-on pops can be minimized. SELECTING FEEDBACK CAPACITOR FOR AUDIO AMPLIFIER The LM4962 is unity-gain stable which gives the designer maximum system flexability. However, to drive ceramic speakers, a typical application requires a closed-loop differential gain of 10. In this case a feedback capacitor (Cf2) will be needed as shown in Figure 2 to bandwidth limit the amplifier. This feedback capacitor creates a low pass filter that eliminates possible high frequency noise. Care should be taken when calculating the -3dB frequency because an incorrect combination of Rf and Cf2 will cause rolloff before the desired frequency SELECTING OUTPUT CAPACITOR (C2) FOR BOOST CONVERTER A single 4.7F to 10F ceramic capacitor will provide sufficient output capacitance for most applications. If larger amounts of capacitance are desired for improved line support and transient response, tantalum capacitors can be used. Aluminum electrolytics with ultra low ESR such as Sanyo Oscon can be used, but are usually prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies above 500 kHz because of significant ringing and temperature rise due to self-heating from ripple current. An output capacitor with excessive ESR can also reduce phase margin and cause instability. In general, if electrolytics are used, we recommended that they be paralleled with ceramic capacitors to reduce ringing, switching losses, and output voltage ripple. SELECTING INPUT CAPACITOR (Cs1) FOR BOOST CONVERTER An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each time the switch turns ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We recommend a nominal value of 4.7F, but larger values can be used. Since this capacitor reduces the amount of voltage ripple seen at the input pin, it also reduces the amount of EMI passed back along that line to other circuitry. SETTING THE OUTPUT VOLTAGE (V1) OF BOOST CONVERTER The output voltage is set using the external resistors R2 and R5 (see Figure 2). A value of approximately 25k is recommended for R2 to establish the open loop gain of the boost converter. V1 = VFB [1 + (R2 / R5)] (6) Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 15 LM4962 SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 www.ti.com FEED-FORWARD COMPENSATION FOR BOOST CONVERTER Although the LM4962's internal Boost converter is internally compensated, the external feed-forward capacitor Cf is required for stability (see Figure 2). Adding this capacitor puts a zero in the loop response of the converter. The recommended frequency for the zero fz should be approximately 60kHz. C3 can be calculated using the formula: C3 = 1 / (2 x R2 x fz) (7) SELECTING A SOFT-START CAPACITOR (Css) The soft-start function charges the boost converter reference voltage slowly, which allows the output of the boost converter to ramp up slowly thus limiting the transient current at startup. Selecting a soft-start capacitor (Css) value presents a trade off between the wake-up time of the boost converter (TWUBC) and the startup transient current. Using a larger capacitor value will increase wake-up time and decrease startup transient current; on the flip side, using a smaller capacitor value will decrease wake-up time and increase the transient current seen at startup. A standard rule of thumb is to use a capacitor 1000 times smaller than the output capacitance of the boost converter (C2+Cs2). A 10nF soft-start capacitor is recommended for a typical application. SELECTING A VALUE FOR Rchg The audio power amplifier integrated in the LM4962 is designed for very fast turn on time. The Cchg pin allows the input capacitor (CInA) to charge quickly to improve click/pop performance. Resistor, Rchg, protects the Cchg pin from any over/under voltage conditions caused by excessive input signal, or an active input signal when the device is in shutdown. The recommended value for Rchg is 1k. If the input signal is less than VDD+0.3V and greater than -0.3V, and if the input signal is disabled when in shutdown mode, Rchg may be shorted. SELECTING DIODES The external diode used in Figure 2 should be a Schottky diode. A 20V diode such as the MBR0520 from Fairchild Semiconductor is recommended. The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications exceeding 0.5A average but less than 1A, a Microsemi UPS5817 can be used. DUTY CYCLE The maximum duty cycle of the boost converter determines the maximum boost ratio of output-to-input voltage that the converter can attain in continuous mode of operation. The duty cycle for a given boost application is defined as: Duty Cycle = (VOUT + VDIODE - VDD) / (VAMP + VDIODE - VSW) This applies for continuous mode operation. INDUCTANCE VALUE The first question we are usually asked is: "How small can I make the inductor." (because they are the largest sized component and usually the most costly). The answer is not simple and involves trade-offs in performance. Larger inductors mean less inductor ripple current, which typically means less output voltage ripple (for a given size of output capacitor). Larger inductors also mean more load power can be delivered because the energy stored during each switching cycle is: E = L/2 x (lp)2 (8) where: "lp" is the peak inductor current 16 Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 LM4962 www.ti.com SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 An important point to observe is that the LM4962 will limit its switch current based on peak current. This means that since lp(max) is fixed, increasing L will increase the maximum amount of power available to the load. Conversely, using too little inductance may limit the amount of load current which can be drawn from the output. Best performance is usually obtained when the converter is operated in "continuous" mode at the load current range of interest, typically giving better load regulation and less output ripple. Continuous operation is defined as not allowing the inductor current to drop to zero during the cycle. It should be noted that all boost converters shift over to discontinuous operation as the output load is reduced far enough, but a larger inductor stays "continuous" over a wider load current range. Taiyo-Yudens NR4012 inductor series is recommended. MAXIMUM SWITCH CURRENT The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the application. This is illustrated in a graph in the Typical Performance Characteristics section which shows typical values of switch current as a function of effective (actual) duty cycle. CALCULATING OUTPUT CURRENT OF BOOST CONVERTER (IAMP) The load current of the Boost Converter is related to the average inductor current by the relation: IAMP = IIND(AVG) x (1 - DC) (9) Where "DC" is the duty cycle of the application. The switch current can be found by: ISW = IIND(AVG) + 1/2 (IRIPPLE) (10) Inductor ripple current is dependent on inductance, duty cycle, supply voltage and frequency: IRIPPLE = DC x (VDD-VSW) / (f x L) (11) combining all terms, we can develop an expression which allows the maximum available load current to be calculated: IAMP(max) = (1-DC)x(ISW(max)-DC(VDD-VSW))/2fL (12) The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF switching losses of the FET and diode. DESIGN PARAMETERS VSW AND ISW The value of the FET "ON" voltage (referred to as VSW in Equations (9) thru (12) is dependent on load current. A good approximation can be obtained by multiplying the "ON Resistance" of the FET times the average inductor current. The maximum peak switch current the device can deliver is dependent on duty cycle. INDUCTOR SUPPLIERS The recommended inductors for the LM4962 is the Taiyo-Yuden NR4012. When selecting an inductor, make certain that the continuous current rating is high enough to avoid saturation at peak currents. A suitable core type must be used to minimize core (switching) losses, and wire power losses must be considered when selecting the current rating. Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 17 LM4962 SNAS300D - NOVEMBER 2005 - REVISED APRIL 2013 www.ti.com REVISION HISTORY Rev Date Description 1.0 03/31/06 Edited 20142203 and 06, then re-released D/S to the WEB. Changes from Revision C (April 2013) to Revision D * 18 Page Changed layout of National Data Sheet to TI format .......................................................................................................... 17 Submit Documentation Feedback Copyright (c) 2005-2013, Texas Instruments Incorporated Product Folder Links: LM4962 PACKAGE OPTION ADDENDUM www.ti.com 5-Apr-2013 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish (2) MSL Peak Temp Op Temp (C) Top-Side Markings (3) (4) LM4962TL/NOPB ACTIVE DSBGA YZR 20 250 Green (RoHS & no Sb/Br) SNAGCU Level-1-260C-UNLIM GF7 LM4962TLX/NOPB ACTIVE DSBGA YZR 20 3000 Green (RoHS & no Sb/Br) SNAGCU Level-1-260C-UNLIM GF7 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) Multiple Top-Side Markings will be inside parentheses. Only one Top-Side Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation of the previous line and the two combined represent the entire Top-Side Marking for that device. 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Addendum-Page 1 Samples PACKAGE MATERIALS INFORMATION www.ti.com 8-Apr-2013 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) LM4962TL/NOPB DSBGA YZR 20 250 178.0 8.4 LM4962TLX/NOPB DSBGA YZR 20 3000 178.0 8.4 Pack Materials-Page 1 B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant 2.18 2.69 0.76 4.0 8.0 Q1 2.18 2.69 0.76 4.0 8.0 Q1 PACKAGE MATERIALS INFORMATION www.ti.com 8-Apr-2013 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LM4962TL/NOPB DSBGA YZR LM4962TLX/NOPB DSBGA YZR 20 250 210.0 185.0 35.0 20 3000 210.0 185.0 35.0 Pack Materials-Page 2 MECHANICAL DATA YZR0020xxx 0.6000.075 D E TLA20XXX (Rev D) D: Max = 2.49 mm, Min = 2.43 mm E: Max = 1.99 mm, Min = 1.93 mm 4215053/A NOTES: A. All linear dimensions are in millimeters. Dimensioning and tolerancing per ASME Y14.5M-1994. B. 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