LT3844IFE#PBF - ANALOG DEVICES

LT3844

3844fc

For more information www.linear.com/LT3844

Typical applicaTion

FeaTures

applicaTions

DescripTion

High Voltage, Current Mode

Switching Regulator Controller with

Programmable Operating Frequency

n Industrial Power Distribution

n 12V and 42V Automotive and Heavy Equipment

n High Voltage Single Board Systems

n Distributed Power Systems

n Avionics

n Telecom Power

High Voltage Step-Down Regulator 48V to 12V, 50W Efficiency and Power Loss

vs Load Current

n High Voltage Operation: Up to 60V

n Output Voltages Up to 36V (Step Down)

n Programmable Constant Frequency: 100kHz to

500kHz

n Synchronizable up to 600kHz

n Burst Mode

Operation: 120µA Supply Current

n 10µA Shutdown Supply Current

n ±1.3% Reference Accuracy

n Drives N-Channel MOSFET

n Programmable Soft-Start

n Programmable Undervoltage Lockout

n Internal High Voltage Regulator for Gate Drive

n Thermal Shutdown

n Current Limit Unaffected by Duty Cycle

n 16-Pin Thermally Enhanced TSSOP Package

The LT

3844 is a DC/DC controller used for medium power,

low part count, high efficiency supplies. It offers a wide

4V to 60V input range (7.5V minimum start-up voltage)

and can implement step-down, step-up, inverting and

SEPIC topologies.

The LT3844 includes Burst Mode operation, which reduces

quiescent current below 120µA and maintains high effi-

ciency at light loads. An internal high voltage bias regulator

allows for simple biasing.

Additional features include current mode control for fast

line and load transient response; programmable fixed

operating frequency that can be synchronized to an ex-

ternal clock for noise sensitive applications; a gate driver

capable of driving large N-channel MOSFETs; a precision

undervoltage lockout function; 10µA shutdown current;

short-circuit protection and a programmable soft-start

function.

The LT3844 is available in a 16-lead thermally enhanced

TSSOP package.

L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.

Burst Mode is a registered trademark of Linear Technology Corporation.

All other trademarks are the property of their respective owners.

Protected by U.S. Patents, including 5731694, 6498466, 6611131

VIN

SHDN

CSS

BURST_EN

VFB

SYNC

fSET

BOOST

VCC

PGND

SENSE+

SENSE–

SGND

LT3844

3844 TA01

49.9k

OUT

12V

50W

68µF

680p

22µF

120pF

68µF

10k

10Ω

1000pF

0.22µF

10µH

33µF

0.015Ω

14.7k

130k

82.5k

1µF

PDS5100H

Si7850DP

36V TO 60V

LOAD CURRENT (A)

3844 TA01b

EFFICIENCY (%)

POWER LOSS (W)

0.1 10

LOSS

EFFICIENCY

VIN = 48V

LT3844

3844fc

For more information www.linear.com/LT3844

pin conFiguraTionabsoluTe MaxiMuM raTings

(Note 1)

FE PACKAGE

16-LEAD PLASTIC TSSOP

TOP VIEW

VIN

SHDN

CSS

BURST_EN

VFB

SYNC

fSET

BOOST

VCC

PGND

SENSE

–

SGND

TJMAX = 125°C, θJA = 40°C/W, θJC = 10°C/W

EXPOSED PAD (PIN 17) IS SGND, MUST BE SOLDERED TO PCB

orDer inForMaTion

LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE

LT3844EFE#PBF LT3844EFE#TRPBF 3844EFE 16-Lead Plastic TSSOP –40°C to 125°C

LT3844IFE#PBF LT3844IFE#TRPBF 3844IFE 16-Lead Plastic TSSOP –40°C to 125°C

Consult LTC Marketing for parts specified with wider operating temperature ranges.

Consult LTC Marketing for information on non-standard lead based finish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through

designated sales channels with #TRMPBF suffix.

Input Supply Voltage (VIN) ......................... 65V to –0.3V

Boosted Supply Voltage (BOOST) .............. 80V to –0.3V

Switch Voltage (SW) (Note 8) ........................65V to –1V

Differential Boost Voltage

BOOST to SW ........................................ 24V to –0.3V

Bias Supply Voltage (VCC) .......................... 24V to –0.3V

SENSE+ and SENSE– Voltages ................... 40V to –0.3V

Differential Sense Voltage

SENSE+ to SENSE– ......................................1V to –1V

BURST_EN Voltage .................................... 24V to –0.3V

SYNC, VC, VFB, CSS and SHDN Voltages ...... 5V to –0.3V

SHDN Pin Currents ..................................................1mA

Operating Junction Temperature Range (Note 2)

LT3844E (Note 3) .............................. –40°C to 125°C

LT3844I ............................................. –40°C to 125°C

Storage Temperature.............................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec) ................... 300°C

elecTrical characTerisTics

The l denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. VIN = 20V, VCC = BOOST = BURST_EN = 10V, SHDN = 2V, RSET = 49.9k,

SENSE– = SENSE+ = 10V, SGND = PGND = SW = SYNC = 0V, unless otherwise noted.

PARAMETER CONDITIONS MIN TYP MAX UNITS

VIN Operating Voltage Range (Note 4)

VIN Minimum Start Voltage

VIN UVLO Threshold (Falling)

VIN UVLO Threshold Hysteresis

3.6

3.8

670

7.5

VIN Supply Current

VIN Burst Mode Current

VIN Shutdown Current

VCC > 9V

VBURST_EN = 0V, VFB = 1.35V

VSHDN = 0V

µA

BOOST Operating Voltage Range

BOOST Operating Voltage Range (Note 5)

BOOST UVLO Threshold (Rising)

BOOST UVLO Threshold Hysteresis

VBOOST – VSW

400

BOOST Supply Current (Note 6)

BOOST Burst Mode Current

BOOST Shutdown Current

VBURST_EN = 0V

VSHDN = 0V

1.4

0.1

µA

http://www.linear.com/product/LT3844#orderinfo

LT3844

3844fc

For more information www.linear.com/LT3844

elecTrical characTerisTics

The l denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. VIN = 20V, VCC = BOOST = BURST_EN = 10V, SHDN = 2V, RSET = 49.9k,

SENSE– = SENSE+ = 10V, SGND = PGND = SW = SYNC = 0V, unless otherwise noted.

PARAMETER CONDITIONS MIN TYP MAX UNITS

VCC Operating Voltage Range (Note 5)

VCC Output Voltage

VCC UVLO Threshold (Rising)

VCC UVLO Threshold Hysteresis

Over Full Line and Load Range

6.25

500

8.3

VCC Supply Current (Note 6)

VCC Burst Mode Current

VCC Shutdown Current

VCC Current Limit

VBURST_EN = 0V

VSHDN = 0V

–40

1.7

–120

2.1 mA

µA

Error Amp Reference Voltage Measured at VFB Pin

1.224

1.215

1.231 1.238

1.245

VFB Pin Input Current VFB = 1.231V 25 nA

SHDN Enable Threshold (Rising)

SHDN Threshold Hysteresis

l1.3 1.35

120

1.4 V

Sense Pins Common Mode Range

Current Limit Sense Voltage

VSENSE+ – VSENSE–

100

115

Input Current

(ISENSE+ + ISENSE–)

VSENSE(CM) = 0V

VSENSE(CM) = 2V

VSENSE(CM) > 4V

350

–25

–170

µA

Operating Frequency

290

270

300 310

330

kHz

Minimum Programmable Frequency

Maximum Programmable Frequency

500

100 kHz

kHz

External Sync Frequency Range l100 600 kHz

SYNC Input Resistance 40 kΩ

SYNC Voltage Threshold l1.4 2 V

Soft-Start Capacitor Control Current 2 µA

Error Amp Transconductance l270 340 410 µS

Error Amp DC Voltage Gain 62 dB

Error Amp Sink/Source Current ±30 µA

TG Drive On Voltage (Note 7)

TG Drive Off Voltage

CLOAD = 2200pF

9.8

0.1

TG Drive Rise/Fall Time 10% to 90% or 90% to 10%, CLOAD = 2200pF 40 ns

Minimum TG Off Time l350 500 ns

Minimum TG On Time l250 350 ns

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 2: The LT3844 includes overtemperature protection that is intended

to protect the device during momentary overload conditions. Junction

temperature will exceed 125°C when overtemperature protection is active.

Continuous operation above the specified maximum operating junction

temperature may impair device reliability.

Note 3: The LT3844E is guaranteed to meet performance specifications from

0°C to 125°C junction temperature. Specifications over the –40°C to 125°C

operating junction temperature range are assured by design, characterization

and correlation with statistical process controls. The LT3844I is guaranteed

over the full –40°C to 125°C operating junction temperature range.

Note 4: VIN voltages below the start-up threshold (7.5V) are only

supported when the VCC is externally driven above 6.5V.

Note 5: Operating range is dictated by MOSFET absolute maximum VGS.

Note 6: Supply current specification does not include switch drive

currents. Actual supply currents will be higher.

Note 7: DC measurement of gate drive output “ON” voltage is typically

8.6V. Internal dynamic bootstrap operation yields typical gate “ON”

voltages of 9.8V during standard switching operation. Standard operation

gate “ON” voltage is not tested but guaranteed by design.

Note 8: The –1V absolute maximum on the SW pin is a transient condition.

It is guaranteed by design and not subject to test.

LT3844

3844fc

For more information www.linear.com/LT3844

3844 G01

SHUTDOWN THRESHOLD, RISING (V)

1.38

1.37

1.36

1.35

1.34

1.33

1.32

TEMPERATURE (°C)

–50 25 75

3844 G02

–25 0 50 100 125

TEMPERATURE (°C)

–50

SHUTDOWN THRESHOLD, FALLING (V)

1.26

1.25

1.24

1.23

1.22

1.21

1.20 25 75–25 0 50 100 125

3844 G05

VIN (V)

VCC (V)

34689

5 7 10 11 12

3844 G07

TEMPERATURE (°C)

–50 25 75–25 0 50 100 125

3844 G08

VCC UVLO THRESHOLD, RISING (V)

6.5

6.4

6.3

6.2

6.1

6.0

TEMPERATURE (°C)

–50

ERROR AMP TRANSCONDUCTANCE (µS)

350

345

340

335

330

325

320 25 75

3844 G09

–25 0 50 100 125

ICC = 20mA

TA = 25°C

TEMPERATURE (°C)

–50 25 75–25 0 50 100 125

3844 G03

8.2

8.1

8.0

7.9

7.8

7.7

7.6

7.5

VCC (V)

ICC = 20mA

–50 –25 1000 50 12525 75

TEMPERATURE (°C)

200

175

150

125

100

3844 G06

ICC CURRENT LIMIT (mA)

ICC(LOAD) (mA)

(V)

3844 G04

10 20 30

8.05

8.00

7.95

7.90

7.85 5 15 25 35

TA = 25°C

VCC (V)

ICC (mA)

02468 10 12 14 18 20

TA = 25°C

Typical perForMance characTerisTics

Shutdown Threshold (Rising)

vs Temperature

VCC vs Temperature

VCC vs ICC(LOAD) VCC vs VIN ICC Current Limit vs Temperature

VCC UVLO Threshold (Rising)

vs Temperature ICC vs VCC (SHDN = 0V)

Error Amp Transconductance

vs Temperature

Shutdown Threshold (Falling)

vs Temperature

LT3844

3844fc

For more information www.linear.com/LT3844

VSENSE (CM) (V)

I(SENSE+ + SENSE–) (µA)

400

300

200

100

–100

–200 0.5 1.0 1.5 2.0

3844 G10

2.5 4.53.5 5.04.03.0

TEMPERATURE (°C)

–50 25 75

3844 G12

–25 0 50 100 125

TEMPERATURE (°C)

–50 25 75

–25 0 50 100 125

TEMPERATURE (°C)

–50 25 75

–25 0 50 100 125

1.234

1.233

1.232

1.231

1.230

1.229

1.228

1.227

ERROR AMP REFERENCE (V)

3844 G14 3844 G15

4.54

4.52

4.50

4.48

4.46

4.44

4.42

4.40

VIN UVLO THRESHOLD, RISING (V)

VIN UVLO THRESHOLD, FALLING (V)

3.86

3.84

3.82

3.80

3.78

3.76

TA = 25°C

TEMPERATURE (°C)

–50

CURRENT SENSE THRESHOLD (mV)

102

104

106

25 75

3844 G16

100

–25 0 50 100 125

TEMPERATURE (°C)

–50

290

OPERATING FREQUENCY (kHz)

292

296

298

300

308

3844 G17

294

–25 75 100

25 125

302

304

306

Typical perForMance characTerisTics

I(SENSE+ + SENSE–) vs

VSENSE(CM)

Operating Frequency

vs Temperature

Error Amp Reference

vs Temperature

Maximum Current Sense

Threshold vs Temperature

VIN UVLO Threshold (Rising)

vs Temperature

VIN UVLO Threshold (Falling)

vs Temperature

LT3844

3844fc

For more information www.linear.com/LT3844

pin FuncTions

VIN (Pin 1): The VIN pin is the main supply pin and should

be decoupled to SGND with a low ESR capacitor located

close to the pin.

SHDN (Pin 2): The SHDN pin has a precision IC enable

threshold of 1.35V (rising) with 120mV of hysteresis. It is

used to implement an undervoltage lockout (UVLO) circuit.

See Applications Information section for implementing

a UVLO function. When the SHDN pin is pulled below

a transistor VBE (0.7V), a low current shutdown mode

is entered, all internal circuitry is disabled and the VIN

supply current is reduced to approximately 9µA. Typical

pin input bias current is <10µA and the pin is internally

clamped to 6V.

CSS (Pin 3): The soft-start pin is used to program the

supply soft-start function. Use the following formula to

calculate CSS for a given output voltage slew rate:

CSS = 2µA(tSS/1.231V)

The pin should be left unconnected when not using the

soft-start function.

BURST_EN (Pin 4): The BURST_EN pin is used to enable

or disable Burst Mode operation. Connect the BURST_EN

pin to ground to enable the burst mode function. Connect

the pin to VFB or VCC to disable the Burst Mode function.

VFB (Pin 5): The output voltage feedback pin, VFB, is exter-

nally connected to the supply output voltage via a resistive

divider. The VFB pin is internally connected to the inverting

input of the error amplifier. In regulation, VFB is 1.231V.

VC (Pin 6): The VC pin is the output of the error amplifier

whose voltage corresponds to the maximum (peak) switch

current per oscillator cycle. The error amplifier is typically

configured as an integrator circuit by connecting an RC

network from the VC pin to SGND. This circuit creates the

dominant pole for the converter regulation control loop.

Specific integrator characteristics can be configured to

optimize transient response. When Burst Mode operation

is enabled (see Pin 4 description), an internal low imped-

ance clamp on the VC pin is set at 100mV below the burst

threshold, which limits the negative excursion of the pin

voltage. Therefore, this pin cannot be pulled low with a

low impedance source. If the VC pin must be externally

manipulated, do so through a 1k series resistance.

SYNC (Pin 7): The SYNC pin provides an external clock

input for synchronization of the internal oscillator. RSET

is set such that the internal oscillator frequency is 10%

to 25% below the external clock frequency. If unused the

SYNC pin is connected to SGND. For more information see

“Oscillator Sync” in the Applications Information section

of this data sheet.

fSET (Pin 8): The fSET pin programs the oscillator frequency

with an external resistor, RSET

. The resistor is required

even when supplying external sync clock signal. See the

Applications Information section for resistor value selec-

tion details.

SGND (Pin 9, 17): The SGND pin is the low noise ground

reference. It should be connected to the –VOUT side of the

output capacitors. Careful layout of the PCB is necessary

to keep high currents away from this SGND connection.

See the Applications Information section for helpful hints

on PCB layout of grounds.

SENSE– (Pin 10): The SENSE– pin is the negative input for

the current sense amplifier and is connected to the VOUT

side of the sense resistor for step-down applications. The

sensed inductor current limit is set to 100mV across the

SENSE inputs.

SENSE+ (Pin 11): The SENSE+ pin is the positive input for

the current sense amplifier and is connected to the induc-

tor side of the sense resistor for step-down applications.

The sensed inductor current limit is set to 100mV across

the SENSE inputs.

PGND (Pin 12): The PGND pin is the high current ground

reference for internal low side switch and the VCC regulator

circuit. Connect the pin directly to the negative terminal of

the VCC decoupling capacitor. See the Applications Informa-

tion section for helpful hints on PCB layout of grounds.

LT3844

3844fc

For more information www.linear.com/LT3844

pin FuncTions

VCC (Pin 13): The VCC pin is the internal bias supply

decoupling node. Use a low ESR 1µF or greater ceramic

capacitor to decouple this node to PGND. Most internal IC

functions are powered from this bias supply. An external

diode connected from VCC to the BOOST pin charges the

bootstrapped capacitor during the off-time of the main

power switch. Back driving the VCC pin from an external

DC voltage source, such as the VOUT output of the regula-

tor supply, increases overall efficiency and reduces power

dissipation in the IC. In shutdown mode this pin sinks

20µA until the pin voltage is discharged to 0V.

SW (Pin 14): In step-down applications the SW pin is

connected to the cathode of an external clamping Schottky

diode, the drain of the power MOSFET and the inductor.

The SW node voltage swing is from VIN during the on-

time of the power MOSFET, to a Schottky voltage drop

below ground during the off-time of the power MOSFET.

In start-up and in operating modes where there is insuf-

ficient inductor current to freewheel the Schottky diode, an

internal switch is turned on to pull the SW pin to ground

so that the BOOST pin capacitor can be charged. Give

careful consideration in choosing the Schottky diode to

limit the negative voltage swing on the SW pin.

TG (Pin 15): The TG pin is the bootstrapped gate drive for

the top N-Channel MOSFET. Since very fast high currents

are driven from this pin, connect it to the gate of the power

MOSFET with a short and wide, typically 0.02" width, PCB

trace to minimize inductance.

BOOST (Pin 16): The BOOST pin is the supply for the

bootstrapped gate drive and is externally connected to a

low ESR ceramic boost capacitor referenced to SW pin.

The recommended value of the BOOST capacitor, CBOOST

is 50 times greater than the total input capacitance of the

topside MOSFET. In most applications 0.1µF is adequate.

The maximum voltage that this pin sees is VIN + VCC,

ground referred.

Exposed Pad (Pin 17): SGND. The exposed leadframe is

internally connected to the SGND pin. Solder the exposed

pad to the PCB ground for electrical contact and optimal

thermal performance.

LT3844

3844fc

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FuncTional DiagraM

–+

–

VIN

UVLO

(<4V)

BST

UVLO

REGULATOR

FEEDBACK

REFERENCE

–

1.231V

3.8V

REGULATOR INTERNAL

SUPPLY RAIL

VIN VCC

UVLO

(<6V)

SHDN

DRIVE

CONTROL

NOL

SWITCH

LOGIC

DRIVE

CONTROL

BURST_EN

CSS

SENSE–

VFB

0.5V

2µA

50µA

Burst Mode

OPERATION

OSCILLATOR

SLOPE COMP

GENERATOR

BOOST

VCC

PGND

SYNC

fSET

RSET

SENSE+

SGND

BOOSTED SWITCH

DRIVER

CURRENT

SENSE

COMPARATOR

100mV

VREF + 200mV

SOFT-START

BURST DISABLE

DRIVER

FAULT CONDITION:

VIN UVLO

VCC UVLO

VSHDN UVLO

CSS CLAMPED TO

VFB + VBE

VREF

~1V

–

ERROR

AMP

–+

(OPTIONAL)

RSENSE

3844 FD

CC2

CC1

VIN

CIN

CBOOST

VOUT

COUT

CVCC

LT3844

3844fc

For more information www.linear.com/LT3844

operaTion

(Refer to Functional Diagram)

The LT3844 is a PWM controller with a constant frequency,

current mode control architecture. It is designed for low

to medium power, switching regulator applications. Its

high operating voltage capability allows it to step up

or down input voltages up to 60V without the need for

a transformer. The LT3844 is used in nonsynchronous

applications, meaning that a freewheeling rectifier diode

(D1 of Function Diagram) is used instead of a bottom

side MOSFET. For circuit operation, please refer to the

Functional Diagram of the IC and Typical Application on

the front page of the data sheet. The LT3800 is a similar

part that uses synchronous rectification, replacing the

diode with a MOSFET in a step-down application.

Main Control Loop

During normal operation, the external N-channel MOSFET

switch is turned on at the beginning of each cycle. The

switch stays on until the current in the inductor exceeds

a current threshold set by the DC control voltage, VC,

which is the output of the voltage control loop. The voltage

control loop monitors the output voltage, via the VFB pin

voltage, and compares it to an internal 1.231V reference.

It increases the current threshold when the VFB voltage

is below the reference voltage and decreases the current

threshold when the VFB voltage is above the reference

voltage. For instance, when an increase in the load current

occurs, the output voltage drops causing the VFB volt-

age to drop relative to the 1.231V reference. The voltage

control loop senses the drop and increases the current

threshold. The peak inductor current is increased until the

average inductor current equals the new load current and

the output voltage returns to regulation.

Current Limit/Short-Circuit

The inductor current is measured with a series sense

resistor (see the Typical Application on the front page).

When the voltage across the sense resistor reaches the

maximum current sense threshold, typically 100mV, the

TG MOSFET driver is disabled for the remainder of that

cycle. If the maximum current sense threshold is still ex-

ceeded at the beginning of the next cycle, the entire cycle

is skipped. Cycle skipping keeps the inductor currents to a

reasonable value during a short-circuit, particularly when

VIN is high. Setting the sense resistor value is discussed

in the “Application Information” section.

VCC/Boosted Supply

An internal VCC regulator provides VIN derived gate-drive

power for start-up under all operating conditions with

MOSFET gate charge loads up to 90nC. The regulator can

operate continuously in applications with VIN voltages up

to 60V, provided the power dissipation of the regulator

does not exceed 250mW. The power dissipation is cal-

culated as follows:

Pd(REG) = (VIN – 8V) • fSW • QG

where QG is the MOSFET gate charge.

In applications where these conditions are exceeded, VCC

must be derived from an external source after start-up.

Maximum continuous regulator power dissipation may be

exceeded for short duration VIN transients.

For higher converter efficiency and less power dissipa-

tion in the IC, VCC can also be supplied from an external

supply such as the converter output. When an external

supply back drives the internal VCC regulator through an

external diode and the VCC voltage is pulled to a diode

above its regulation voltage, the internal regulator is dis-

abled and goes into a low current mode. VCC is the bias

supply for most of the internal IC functions and is also

used to charge the bootstrapped capacitor (CBOOST) via an

external diode. The external MOSFET switch is biased from

the bootstrapped capacitor. While the external MOSFET

switch is off, an internal BJT switch, whose collector is

connected to the SW pin and emitter is connected to the

PGND pin, is turned on to pull the SW node to PGND and

recharge the bootstrap capacitor. The switch stays on until

either the start of the next cycle or until the bootstrapped

capacitor is fully charged.

MOSFET Driver

The LT3844 contains a high speed boosted driver to turn

on and off an external N-channel MOSFET switch. The

MOSFET driver derives its power from the boost capacitor

which is referenced to the SW pin and the source of the

MOSFET. The driver provides a large pulse of current to turn

on the MOSFET fast to minimize transition times. Multiple

MOSFETs can be paralleled for higher current operation.

LT3844

3844fc

For more information www.linear.com/LT3844

operaTion

(Refer to Functional Diagram)

To eliminate the possibility of shoot through between the

MOSFET and the internal SW pull-down switch, an adap-

tive nonoverlap circuit ensures that the internal pull-down

switch does not turn on until the gate of the MOSFET is

below its turn on threshold.

Low Current Operation (Burst Mode Operation)

To increase low current load efficiency, the LT3844 is

capable of operating in Linear Technology’s proprietary

Burst Mode operation where the external MOSFET operates

intermittently based on load current demand. The Burst

Mode function is disabled by connecting the BURST_EN pin

to VCC or VFB and enabled by connecting the pin to SGND.

When the required switch current, sensed via the VC pin

voltage, is below 15% of maximum, Burst Mode operation

is employed and that level of sense current is latched onto

the IC control path. If the output load requires less than

this latched current level, the converter will overdrive the

output slightly during each switch cycle. This overdrive

condition is sensed internally and forces the voltage on the

VC pin to continue to drop. When the voltage on VC drops

150mV below the 15% load level, switching is disabled,

and the LT3844 shuts down most of its internal circuitry,

reducing total quiescent current to 120µA. When the

converter output begins to fall, the VC pin voltage begins

to climb. When the voltage on the VC pin climbs back to

the 15% load level, the IC returns to normal operation

and switching resumes. An internal clamp on the VC pin

is set at 100mV below the output disable threshold, which

limits the negative excursion of the pin voltage, minimizing

the converter output ripple during Burst Mode operation.

During Burst Mode operation, the VIN pin current is 20µA

and the VCC current is reduced to 100µA. If no external

drive is provided for VCC, all VCC bias currents originate

from the VIN pin, giving a total VIN current of 120µA. Burst

current can be reduced further when VCC is driven using

an output derived source, as the VCC component of VIN

current is then reduced by the converter duty cycle ratio.

Start-Up

The following section describes the start-up of the supply

and operation down to 4V once the step-down supply is

up and running. For the protection of the LT3844 and the

switching supply, there are internal undervoltage lockout

(UVLO) circuits with hysteresis on VIN, VCC and VBOOST

as shown in the Electrical Characteristics table. Start-up

and continuous operation require that all three of these

undervoltage lockout conditions be satisfied because the

TG MOSFET driver is disabled during any UVLO fault con-

dition. In start-up, for most applications, VCC is powered

from VIN through the high voltage linear regulator of the

LT3844. This requires VIN to be high enough to drive the

VCC voltage above its undervoltage lockout threshold.

VCC, in turn, has to be high enough to charge the BOOST

capacitor through an external diode so that the BOOST

voltage is above its undervoltage lockout threshold. There

is an NPN switch that pulls the SW node to ground each

cycle during the TG power MOSFET off-time, ensuring the

BOOST capacitor is kept fully charged. Once the supply

is up and running, the output voltage of the supply can

backdrive VCC through an external diode. Internal circuitry

disables the high voltage regulator to conserve VIN supply

current. Output voltages that are too low or too high to

backdrive VCC require additional circuitry such as a voltage

doubler or linear regulator. Once VCC is backdriven from

a supply other than VIN, VIN can be reduced to 4V with

normal operation maintained.

Soft-Start

The soft-start function controls the slew rate of the power

supply output voltage during start-up. A controlled output

voltage ramp minimizes output voltage overshoot, reduces

inrush current from the VIN supply, and facilitates supply

sequencing. A capacitor, CSS, connected from the CSS pin

to SGND, programs the slew rate. The capacitor is charged

from an internal 2µA current source producing a ramped

voltage. The capacitor voltage overrides the internal refer-

ence to the error amplifier. If the VFB pin voltage exceeds

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the CSS pin voltage then the current threshold set by the

DC control voltage, VC, is decreased and the inductor cur-

rent is lowered. This in turn decreases the output voltage

slew rate allowing the CSS pin voltage ramp to catch up

to the VFB pin voltage. An internal 100mV offset is added

to the VFB pin voltage relative to the to CSS pin voltage

so that at start-up the soft-start circuit will discharge the

VC pin voltage below the DC control voltage equivalent to

zero inductor current. This will reduce the input supply

inrush current. The soft-start circuit is disabled once the

CSS pin voltage has been charged to 200mV above the

internal reference of 1.231V.

During a VIN UVLO, VCC UVLO or SHDN UVLO event, the

CSS pin voltage is discharged with a 50µA current source.

In normal operation the CSS pin voltage is clamped to a

diode above the VFB pin voltage. Therefore, the value of

the CSS capacitor is relevant in how long of a fault event

will retrigger a soft-start. In other words, if any of the

above UVLO conditions occur, the CSS pin voltage will be

discharged with a 50µA current source. There is a diode

worth of voltage headroom to ride through the fault before

the CSS pin voltage enters its active region and the soft-

start function is enabled.

Also, since the CSS pin voltage is clamped to a diode above

the VFB pin voltage, during a short circuit the CSS pin volt-

age is pulled low because the VFB pin voltage is low. Once

the short has been removed the VFB pin voltage starts to

recover. The soft-start circuit takes control of the output

voltage slew rate once the VFB pin voltage has exceeded

the slowly ramping CSS pin voltage, reducing the output

voltage overshoot during a short-circuit recovery.

Slope/Antislope Compensation

The IC incorporates slope compensation to eliminate

potential subharmonic oscillations in the current control

loop. The IC’s slope compensation circuit imposes an

artificial ramp on the sensed current to increase the rising

slope as duty cycle increases.

Typically, this additional ramp affects the sensed current

value, thereby reducing the achievable current limit value

by the same amount as the added ramp represents. As

such, the current limit is typically reduced as the duty cycle

increases. The LT3844, however, contains antislope com-

pensation circuitry to eliminate the current limit reduction

associated with slope compensation. As the slope com-

pensation ramp is added to the sensed current, a similar

ramp is added to the current limit threshold. The end result

is that the current limit is not compromised so the LT3844

can provide full power regardless of required duty cycle.

Shutdown

The LT3844 includes a shutdown mode where all the

internal IC functions are disabled and the VIN current

is reduced to less than 10µA. The shutdown pin can be

used for undervoltage lockout with hysteresis, micro-

power shutdown or as a general purpose on/off control

of the converter output. The shutdown function has two

thresholds. The first threshold, a precision 1.23V threshold

with 120mV of hysteresis, disables the converter from

switching. The second threshold, approximately a 0.7V

referenced to SGND, completely disables all internal cir-

cuitry and reduces the VIN current to less than 10µA. See

the Application Information section for more information.

applicaTions inForMaTion

The basic LT3844 step-down (buck) application, shown

in the Typical Application on the front page, converts a

larger positive input voltage to a lower positive or negative

output voltage. This Application Information section assists

selection of external components for the requirements of

the power supply.

RSENSE Selection

The current sense resistor, RSENSE, monitors the induc-

tor current of the supply (See Typical Application on

front page). Its value is chosen based on the maximum

required output load current. The LT3844 current sense

amplifier has a maximum voltage threshold of, typically,

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100mV. Therefore, the peak inductor current is 100mV/

RSENSE. The maximum output load current, IOUT(MAX), is

the peak inductor current minus half the peak-to-peak

ripple current, ∆IL.

Allowing adequate margin for ripple current and exter-

nal component tolerances, RSENSE can be calculated as

follows:

RSENSE =

70mV

IOUT(MAX)

Typical values for RSENSE are in the range of 0.005Ω to

0.05Ω.

Operating Frequency

The choice of operating frequency and inductor value is

a trade off between efficiency and component size. Low

frequency operation improves efficiency by reducing MOS-

FET switching losses and gate charge losses. However,

lower frequency operation requires more inductance for a

given amount of ripple current, resulting in a larger induc-

tor size and higher cost. If the ripple current is allowed

to increase, larger output capacitors may be required to

maintain the same output ripple. For converters with high

step-down VIN-to-VOUT ratios, another consideration is

the minimum on-time of the LT3844 (see the Minimum

On-time Considerations section). A final consideration

for operating frequency is that in noise-sensitive com-

munications systems, it is often desirable to keep the

switching noise out of a sensitive frequency band. The

LT3844 uses a constant frequency architecture that can

be programmed over a 100kHz to 500kHz range with a

single resistor from the fSET pin to ground, as shown in

Figure 1. The nominal voltage on the fSET pin is 1V and

the current that flows from this pin is used to charge an

internal oscillator capacitor. The value of RSET for a given

operating frequency can be chosen from Figure 4 or from

the following equation:

SET

(kΩ)=8.4 •10

•f

SW (

−

1.31)

Table 1 lists typical resistor values for common operating

frequencies:

Table 1. Recommended 1% Standard Values

RSET fSW

191kΩ 100kHz

118kΩ 150kHz

80.6kΩ 200kHz

63.4kΩ 250kHz

49.9kΩ 300kHz

40.2kΩ 350kHz

33.2kΩ 400kHz

27.4kΩ 450kHz

23.2kΩ 500kHz

Step-Down Converter: Inductor Selection

The critical parameters for selection of an inductor are

minimum inductance value, volt-second product, satura-

tion current and/or RMS current.

For a given ∆I, The minimum inductance value is calcu-

lated as follows:

L≥VOUT •

IN(MAX) −

OUT

fSW •V

IN(MAX) •∆IL

fSW is the switch frequency.

FREQUENCY (kHz)

RSET (kΩ)

100

120

400

200

3844 G19

200

100 500

300 600

140

160

180

Figure 1. Timing Resistor (RSET) Value

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The typical range of values for ∆IL is (0.2 • IOUT(MAX)) to

(0.5 • IOUT(MAX)), where IOUT(MAX) is the maximum load

current of the supply. Using ∆IL = 0.3 • IOUT(MAX) yields a

good design compromise between inductor performance

versus inductor size and cost. A value of ∆IL = 0.3 • IOUT(MAX)

produces a ±15% of IOUT(MAX) ripple current around the DC

output current of the supply. Lower values of ∆IL require

larger and more costly magnetics. Higher values of ∆IL

will increase the peak currents, requiring more filtering

on the input and output of the supply. If ∆IL is too high,

the slope compensation circuit is ineffective and current

mode instability may occur at duty cycles greater than

50%. To satisfy slope compensation requirements the

minimum inductance is calculated as follows:

L>VOUT •

2DC

MAX −

DCMAX

•

SENSE

•8.33

Some magnetics vendors specify a volt-second product

in their data sheet. If they do not, consult the magnetics

vendor to make sure the specification is not being exceeded

by your design. The volt-second product is calculated as

follows:

Volt-second (µs) =

IN(MAX) −

OUT

)•V

OUT

IN(MAX) •fSW

The magnetics vendors specify either the saturation cur-

rent, the RMS current or both. When selecting an inductor

based on inductor saturation current, use the peak cur-

rent through the inductor, IOUT(MAX) + ∆IL/2. The inductor

saturation current specification is the current at which

the inductance, measured at zero current, decreases by

a specified amount, typically 30%.

When selecting an inductor based on RMS current rating,

use the average current through the inductor, IOUT(MAX).

The RMS current specification is the RMS current at which

the part has a specific temperature rise, typically 40°C,

above 25°C ambient.

After calculating the minimum inductance value, the

volt-second product, the saturation current and the RMS

current for your design, select an off-the-shelf inductor.

Contact the Application group at Linear Technology for

further support.

For more detailed information on selecting an inductor,

please see the “Inductor Selection” section of Linear

Technology Application Note 44.

Step-Down Converter: MOSFET Selection

The selection criteria of the external N-channel standard

level power MOSFET include on resistance(RDS(ON)), re-

verse transfer capacitance (CRSS), maximum drain source

voltage (VDSS), total gate charge (QG) and maximum

continuous drain current.

For maximum efficiency, minimize RDS(ON) and CRSS.

Low RDS(ON) minimizes conduction losses while low CRSS

minimizes transition losses. The problem is that RDS(ON) is

inversely related to CRSS. Balancing the transition losses

with the conduction losses is a good idea in sizing the

MOSFET. Select the MOSFET to balance the two losses.

Calculate the maximum conduction losses of the MOSFET:

COND =(IOUT(MAX))2VOUT

⎛

⎝

⎜⎞

⎠

⎟(RDS(ON))

Note that RDS(ON) has a large positive temperature depen-

dence. The MOSFET manufacturer’s data sheet contains

a curve, RDS(ON) vs Temperature.

Calculate the maximum transition losses:

PTRAN = (k)(VIN)2 (IOUT(MAX))(CRSS)(fSW)

where k is a constant inversely related to the gate driver

current, approximated by k = 2 for LT3844 applications.

The total maximum power dissipation of the MOSFET is

the sum of these two loss terms:

PFET(TOTAL) = PCOND + PTRAN

To achieve high supply efficiency, keep the PFET(TOTAL) to

less than 3% of the total output power. Also, complete

a thermal analysis to ensure that the MOSFET junction

temperature is not exceeded.

TJ = TA + PFET(TOTAL) • θJA

where θJA is the package thermal resistance and TA is the

ambient temperature. Keep the calculated TJ below the

maximum specified junction temperature, typically 150°C.

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Note that when VIN is high and fSW is high, the transition

losses may dominate. A MOSFET with higher RDS(ON)

and lower CRSS may provide higher efficiency. MOSFETs

with higher voltage VDSS specification usually have higher

RDS(ON) and lower CRSS.

Choose the MOSFET VDSS specification to exceed the

maximum voltage across the drain to the source of the

MOSFET, which is VIN(MAX) plus any additional ringing

on the switch node. Ringing on the switch node can be

greatly reduced with good PCB layout and, if necessary,

an RC snubber.

The internal VCC regulator is capable of sourcing up to

40mA which limits the maximum total MOSFET gate

charge, QG, to 40mA/fSW

. The QG vs VGS specification is

typically provided in the MOSFET data sheet. Use QG at

VGS of 8V. If VCC is back driven from an external supply,

the MOSFET drive current is not sourced from the internal

regulator of the LT3844 and the QG of the MOSFET is not

limited by the IC. However, note that the MOSFET drive

current is supplied by the internal regulator when the

external supply back driving VCC is not available such as

during start-up or short-circuit.

The manufacturer’s maximum continuous drain current

specification should exceed the peak switch current,

IOUT(MAX) + ∆IL/2.

During the supply start-up, the gate drive levels are set by

the VCC voltage regulator, which is approximately 8V. Once

the supply is up and running, the VCC can be back driven

by an auxiliary supply such as VOUT

. It is important not to

exceed the manufacturer’s maximum VGS specification.

A standard level threshold MOSFET typically has a VGS

maximum of 20V.

Step-Down Converter: Rectifier Selection

The rectifier diode (D1 on the Functional Diagram) in a

buck converter generates a current path for the inductor

current when the main power switch is turned off. The

rectifier is selected based upon the forward voltage, re-

verse voltage and maximum current. A Schottky diode is

recommended. Its low forward voltage yields the lowest

power loss and highest efficiency. The maximum reverse

voltage that the diode will see is VIN(MAX).

In continuous mode operation, the average diode cur-

rent is calculated at maximum output load current and

maximum VIN:

IDIODE(AVG) =IOUT(MAX)

IN(MAX) −

OUT

IN(MAX)

To improve efficiency and to provide adequate margin

for short-circuit operation, a diode rated at 1.5 to 2

times the maximum average diode current, IDIODE(AVG),

is recommended.

Step-Down Converter: Input Capacitor Selection

A local input bypass capacitor is required for buck convert-

ers because the input current is pulsed with fast rise and

fall times. The input capacitor selection criteria are based

on the bulk capacitance and RMS current capability. The

bulk capacitance will determine the supply input ripple

voltage. The RMS current capability is used to keep from

overheating the capacitor.

The bulk capacitance is calculated based on maximum

input ripple, ∆VIN:

CIN(BULK) =

OUT(MAX)

•V

OUT

∆V

IN •fSW •V

IN(MIN)

∆VIN is typically chosen at a level acceptable to the user.

100mV to 200mV is a good starting point. Aluminum elec-

trolytic capacitors are a good choice for high voltage, bulk

capacitance due to their high capacitance per unit area.

The capacitor’s RMS current is:

ICIN(RMS) =IOUT

VOUT(V

IN −VOUT )

IN )2

If applicable, calculate it at the worst-case condition,

VIN = 2VOUT

. The RMS current rating of the capacitor

is specified by the manufacturer and should exceed the

calculated ICIN(RMS). Due to their low ESR (equivalent

series resistance), ceramic capacitors are a good choice

for high voltage, high RMS current handling. Note that the

ripple current ratings from aluminum electrolytic capacitor

manufacturers are based on 2000 hours of life. This makes

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it advisable to further derate the capacitor or to choose

a capacitor rated at a higher temperature than required.

The combination of aluminum electrolytic capacitors and

ceramic capacitors is an economical approach to meet-

ing the input capacitor requirements. The capacitor volt-

age rating must be rated greater than VIN(MAX). Multiple

capacitors may also be paralleled to meet size or height

requirements in the design. Locate the capacitor very close

to the MOSFET switch and use short, wide PCB traces to

minimize parasitic inductance.

Step-Down Converter: Output Capacitor Selection

The output capacitance, COUT

, selection is based on the

design’s output voltage ripple, ∆VOUT and transient load

requirements. ∆VOUT is a function of ∆IL and the COUT

ESR. It is calculated by:

∆VOUT = ∆IL•ESR+1

(8 •f

•C

OUT

)

⎛

⎝

⎜

⎞

⎠

⎟

The maximum ESR required to meet a ∆VOUT design

requirement can be calculated by:

ESR(MAX) =

(

∆

OUT

)(L)(f

)

VOUT •1−VOUT

IN(MAX)

⎛

⎝

⎜

⎞

⎠

⎟

Worst-case ∆VOUT occurs at highest input voltage. Use

paralleled multiple capacitors to meet the ESR require-

ments. Increasing the inductance is an option to lower

the ESR requirements. For extremely low ∆VOUT

, an ad-

ditional LC filter stage can be added to the output of the

supply. Application Note 44 has some good tips on sizing

an additional output filter.

Output Voltage Programming

A resistive divider sets the DC output voltage according

to the following formula:

R2 =R1 VOUT

1.231V −1

⎛

⎝

⎜⎞

⎠

⎟

The external resistor divider is connected to the output

of the converter as shown in Figure 2. Tolerance of the

feedback resistors will add additional error to the output

voltage.

Example: VOUT = 12V; R1 = 10k

R2 =10k 12V

1.231V −1

⎛

⎝

⎜⎞

⎠

⎟=87.48k −use 86.6k 1%

The VFB pin input bias current is typically 25nA, so use

of extremely high value feedback resistors could cause a

converter output that is slightly higher than expected. Bias

current error at the output can be estimated as:

∆VOUT(BIAS) = 25nA • R2

Supply UVLO and Shutdown

The SHDN pin has a precision voltage threshold with

hysteresis which can be used as an undervoltage lockout

threshold (UVLO) for the power supply. Undervoltage

lockout keeps the LT3844 in shutdown until the supply

input voltage is above a certain voltage programmed by

the user. The hysteresis voltage prevents noise from falsely

tripping UVLO.

Resistors are chosen by first selecting RB. Then:

RA=RB•VSUPPLY(ON)

1.35V −1

⎛

⎝

⎜⎞

⎠

⎟

VFB PIN

VOUT

COUT

3844 F02

SHDN PIN

SUPPLY

3844 F03

Figure 2. Output Voltage Feedback Divider Figure 3. Undervoltage Lockout Circuit

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VSUPPLY(ON) is the input voltage at which the undervoltage

lockout is disabled and the supply turns on.

Example: Select RB = 49.9kΩ, VSUPPLY(ON) = 14.5V (based

on a 15V minimum input voltage)

RA=49.9k •14.5V

1.35V −1

⎛

⎝

⎜⎞

⎠

⎟

= 486.1k (499k resistor is selected)

If low supply current in standby mode is required, select

a higher value of RB.

The supply turn off voltage is 9% below turn on. In the

example the VSUPPLY(OFF) would be 13.2V.

If additional hysteresis is desired for the enable function,

an external positive feedback resistor can be used from

the LT3844 regulator output.

The shutdown function can be disabled by connecting the

SHDN pin to the VIN through a large value pull-up resistor.

This pin contains a low impedance clamp at 6V, so the

SHDN pin will sink current from the pull-up resistor(RPU):

ISHDN =

IN −

Because this arrangement will clamp the SHDN pin to

the 6V, it will violate the 5V absolute maximum voltage

rating of the pin. This is permitted, however, as long as

the absolute maximum input current rating of 1mA is not

exceeded. Input SHDN pin currents of <100µA are recom-

mended: a 1M or greater pull-up resistor is typically used

for this configuration.

Soft-Start

The desired soft-start time (tSS) is programmed via the

CSS capacitor as follows:

CSS =

2µA •t

1.231V

The amount of time in which the power supply can withstand

a VIN, VCC or VSHDN UVLO fault condition (tFAULT) before

the CSS pin voltage enters its active region is approximated

by the following formula:

tFAULT =

•0.65V

50µA

Oscillator SYNC

The oscillator can be synchronized to an external clock.

Set the RSET resistor at least 10% below the desired sync

frequency.

It is recommended that the SYNC pin be driven with a

square wave that has amplitude greater than 2V, pulse

width greater than 1ms and rise time less than 500ns. The

rising edge of the sync wave form triggers the discharge

of the internal oscillator capacitor.

Efficiency Considerations

The efficiency of a switching regulator is equal to the output

power divided by the input power times 100%. Express

percent efficiency as:

% Efficiency = 100% - (L1 + L2 + L3 + ...)

where L1, L2, etc. are individual loss terms as a percent-

age of input power.

Although all dissipative elements in the circuit produce

losses, four main contributors usually account for most

of the losses in LT3844 circuits:

1. LT3844 VIN and VCC current loss

2. I2R conduction losses

3. MOSFET transition loss

4. Schottky diode conduction loss

1. The VIN and VCC currents are the sum of the quiescent

currents of the LT3844 and the MOSFET drive currents.

The quiescent currents are in the LT3844 Electrical Char-

acteristics table. The MOSFET drive current is a result

of charging the gate capacitance of the power MOSFET

each cycle with a packet of charge, QG. QG is found in

the MOSFET data sheet. The average charging current is

calculated as QG • fSW

. The power loss term due to these

currents can be reduced by backdriving VCC with a lower

voltage than VIN such as VOUT

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2. I2R losses are calculated from the DC resistances of the

MOSFET, the inductor, the sense resistor and the input

and output capacitors. In continuous conduction mode

the average output current flows through the inductor

and RSENSE but is chopped between the MOSFET and

the Schottky diode. The resistances of the MOSFET

(RDS(ON)) and the RSENSE multiplied by the duty cycle

can be summed with the resistances of the inductor

and RSENSE to obtain the total series resistance of the

circuit. The total conduction power loss is proportional

to this resistance and usually accounts for between 2%

to 5% loss in efficiency.

3. Transition losses of the MOSFET can be substantial with

input voltages greater than 20V. See MOSFET Selection

section.

4. The Schottky diode can be a major contributor of power

loss especially at high input to output voltage ratios (low

duty cycles) where the diode conducts for the majority

of the switch period. Lower Vf reduces the losses. Note

that oversizing the diode does not always help because

as the diode heats up the Vf is reduced and the diode

loss term is decreased.

I2R losses and the Schottky diode loss dominate at

high load currents. Other losses including CIN and

COUT ESR dissipative losses and inductor core losses

generally account for less than 2% total additional loss

in efficiency.

PCB Layout Checklist

When laying out the printed circuit board, the following

checklist should be used to ensure proper operation. These

items are illustrated graphically in the layout diagram of

Figure 3.

1. Keep the signal and power grounds separate. The signal

ground consists of the LT3844 SGND pin, the Exposed

Pad on the backside of the LT3844 IC and the (–) terminal

of VOUT

. The signal ground is the quiet ground and does

not contain any high, fast currents. The power ground

consists of the Schottky diode anode, the (–) terminal

of the input capacitor and the ground return of the VCC

capacitor. This ground has very fast high currents and

is considered the noisy ground. The two grounds are

connected to each other only at the (–) terminal of VOUT

2. Use short wide traces in the loop formed by the MOSFET,

the Schottky diode and the input capacitor to minimize

high frequency noise and voltage stress from parasitic

inductance. Surface mount components are preferred.

3. Connect the VFB pin directly to the feedback resistors

independent of any other nodes, such as the SENSE–

pin. Connect the feedback resistors between the (+) and

(–) terminals of COUT

. Locate the feedback resistors in

close proximity to the LT3844 to keep the high imped-

ance node, VFB, as short as possible.

4. Route the SENSE– and SENSE+ traces together and

keep as short as possible.

5. Locate the VCC and BOOST capacitors in close proximity

to the IC. These capacitors carry the MOSFET driver’s

high peak currents. Place the small-signal components

away from high frequency switching nodes (BOOST, SW

and TG). In the layout shown in Figure 3, place all the

small-signal components on one side of the IC and all

the power components on the other. This helps to keep

the signal and power grounds separate.

6. A small decoupling capacitor (100pF) is sometimes

useful for filtering high frequency noise on the feedback

and sense nodes. If used, locate as close to the IC as

possible.

7. The LT3844 packaging will efficiently remove heat

from the IC through the Exposed Pad on the backside

of the part. The Exposed Pad is soldered to a copper

footprint on the PCB. Make this footprint as large as

possible to improve the thermal resistance of the IC

case to ambient air. This helps to keep the LT3844 at a

lower temperature.

8. Make the trace connecting the gate of MOSFET M1 to

the TG pin of the LT3844 short and wide.

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CBOOST

RSENSE

RSET

VIN–

VIN

SHDN

SYNC

fSET

CSS

BURST_EN

VFB

SGND

BOOST

VCC

PGND

SENSE+

SENSE–

–

3844 F04

LT3844

CVCC

CIN

COUT VOUT

CC2

CSS

CC1

Figure 4. LT3844 Layout Diagram (See PCB Layout Checklist)

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Minimum On-Time Considerations (Buck Mode)

Minimum on-time, tON(MIN), is the smallest amount of time

that the LT3844 is capable of turning the top MOSFET on

and off again. It is determined by internal timing delays

and the amount of gate charge required turning on the

top MOSFET. Low duty cycle applications may approach

this minimum on-time limit and care should be taken to

ensure that:

tON =

OUT

IN •f

>tON(MIN)

where tON(MIN) is typically 350ns worst case.

If the duty cycle falls below what can be accommodated

by the minimum on-time, the LT3844 will begin to skip

cycles. The output will be regulated, but the ripple current

and ripple voltage will increase. If lower frequency opera-

tion is acceptable, the on-time can be increased above

tON(MIN) for the same step-down ratio.

Boost Converter Design

The LT3844 can be used to configure a boost converter

to step-up voltages to as high as hundreds of volts. An

example of a boost converter circuit schematic is shown

in the Typical Applications section. The following sections

are a guide to designing a boost converter:

The maximum duty cycle of the main switch is:

DCMAX =

OUT −

IN(MIN)

OUT

Boost Converter: Inductor Selection

The critical parameters for selection of an inductor are

minimum inductance value and saturation current. The

minimum inductance value is calculated as follows:

LMIN =V

IN(MIN)

∆I

•f

•DCMAX

fSW is the switch frequency.

Similar to the buck converter, the typical range of values

for ∆IL is (0.2 • IL(MAX)) to (0.5 • IL(MAX)), where IL(MAX)

is the maximum average inductor current.

IL(MAX) =IOUT(MAX) •

OUT

IN(MIN)

Using ∆IL = 0.3 • IL(MAX) yields a good design compromise

between inductor performance versus inductor size and

cost.

The inductor must not saturate at the peak operating

current, IL(MAX) + ∆IL/2. The inductor saturation current

specification is the current at which the inductance, mea-

sured at zero current, decreases by a specified amount,

typically 30%.

One drawback of boost regulators is that they cannot be

current limited for output shorts because the current steer-

ing diode makes a direct connection between input and

output. Therefore, the inductor current during an output

short circuit is only limited by the available current of the

input supply.

After calculating the minimum inductance value and the

saturation current for your design, select an off-the-shelf

inductor. For more detailed information on selecting an

inductor, please see the “Inductor Selection” section of

Linear Technology Application Note 19.

Boost Converter: MOSFET Selection

The selection criteria of the external N-channel standard

level power MOSFET include on resistance (RDS(ON)), re-

verse transfer capacitance (CRSS), maximum drain source

voltage (VDSS), total gate charge (QG) and maximum

continuous drain current.

For maximum efficiency, minimize RDS(ON) and CRSS.

Low RDS(ON) minimizes conduction losses while low CRSS

minimizes transition losses. The problem is that RDS(ON) is

inversely related to CRSS. Balancing the transition losses

with the conduction losses is a good idea in sizing the

LT3844

3844fc

For more information www.linear.com/LT3844

applicaTions inForMaTion

MOSFET. Select the MOSFET to balance the two losses.

Calculate the maximum conduction losses of the MOSFET:

COND =DCMAX

IOUT(MAX)

1−DCMAX

⎛

⎝

⎜⎞

⎠

⎟•RDS(ON)

Note that RDS(ON) has large positive temperature depen-

dence. The MOSFET manufacturer’s data sheet contains

a curve, RDS(ON) vs Temperature. Calculate the maximum

transition losses:

PTRAN =(k)(VOUT )2(IOUT(MAX))(CRSS )(f

SW )

(1−DC

MAX

)

where k is a constant inversely related to the gate driver

current, approximated by k = 2 for LT3844 applications.

The total maximum power dissipation of the MOSFET is

the sum of these two loss terms:

PFET(TOTAL) = PCOND + PTRAN

To achieve high supply efficiency, keep the PFET(TOTAL) to

less than 3% of the total output power. Also, complete

a thermal analysis to ensure that the MOSFET junction

temperature is not exceeded.

TJ = TA + PFET(TOTAL) • θJA

where θJA is the package thermal resistance and TA is the

ambient temperature. Keep the calculated TJ below the

maximum specified junction temperature, typically 150°C.

Note that when VOUT is high (>20V), the transition losses

may dominate. A MOSFET with higher RDS(ON) and lower

CRSS may provide higher efficiency. MOSFETs with higher

voltage VDSS specification usually have higher RDS(ON)

and lower CRSS.

Choose the MOSFET VDSS specification to exceed the

maximum voltage across the drain to the source of the

MOSFET, which is VOUT plus the forward voltage of the

rectifier, typically less than 1V.

The internal VCC regulator is capable of sourcing up to

40mA which limits the maximum total MOSFET gate

charge, QG, to 40mA / fSW

. The QG vs VGS specification

is typically provided in the MOSFET data sheet. Use QG at

VGS of 8V. If VCC is back driven from an external supply,

the MOSFET drive current is not sourced from the internal

regulator of the LT3844 and the QG of the MOSFET is not

limited by the IC. However, note that the MOSFET drive

current is supplied by the internal regulator when the

external supply back driving VCC is not available such as

during start-up or short-circuit.

The manufacturer’s maximum continuous drain current

specification should exceed the peak switch current which

is the same as the inductor peak current, IL(MAX) + ∆IL/2.

During the supply start-up, the gate drive levels are set by

the VCC voltage regulator, which is approximately 8V. Once

the supply is up and running, the VCC can be back driven

by an auxiliary supply such as VOUT

. It is important not to

exceed the manufacturer’s maximum VGS specification.

A standard level threshold MOSFET typically has a VGS

maximum of 20V.

Boost Converter: Rectifier Selection

The rectifier is selected based upon the forward voltage,

reverse voltage and maximum current. A Schottky diode

is recommended for its low forward voltage and yields the

lowest power loss and highest efficiency. The maximum

reverse voltage that the diode will see is VOUT

. The average

diode current is equal to the maximum output load current,

IOUT(MAX). A diode rated at 1.5 to 2 times the maximum

average diode current is recommended. Remember boost

converters are not short-circuit protected.

Boost Converter: Output Capacitor Selection

In boost mode, the output capacitor requirements are

more demanding due to the fact that the current waveform

is pulsed instead of continuous as in a buck converter.

The choice of component(s) is driven by the acceptable

ripple voltage which is affected by the ESR, ESL and bulk

capacitance. The total output ripple voltage is:

∆VOUT =IOUT(MAX)

fSW •COUT

+ESR

1−DCMAX

⎛

⎝

⎜

⎞

⎠

⎟

where the first term is due to the bulk capacitance and the

second term due to the ESR.

LT3844

3844fc

For more information www.linear.com/LT3844

applicaTions inForMaTion

The choice of output capacitor is also driven by the RMS

ripple current requirement. The RMS ripple current is:

IRMS(COUT) =IOUT(MAX) •VOUT −V

IN(MIN)

At lower output voltages (<30V) it may be possible to sat-

isfy both the output ripple voltage and RMS requirements

with one or more capacitors of a single type. However, at

output voltages above 30V where capacitors with both low

ESR and high bulk capacitance are hard to find, the best

approach is to use a combination of aluminum electrolytic

and ceramic capacitors. The low ESR ceramic capacitor

will minimize the ESR while the Aluminum Electrolytic

capacitor will supply the required bulk capacitance.

Boost Converter: Input Capacitor Selection

The input capacitor of a boost converter is less critical

than the output capacitor, due to the fact that the inductor

is in series with the input and the input current waveform

is continuous. The input voltage source impedance de-

termines the size of the input capacitor, which is typically

in the range of 10µF to 100µF. A low ESR capacitor is

recommended though not as critical as with the output

capacitor. The RMS input capacitor ripple current for a

boost converter is:

IRMS(CIN) =0.3 •

IN(MIN)

L•f

•DCMAX

Please note that the input capacitor can see a very high

surge current when a battery is suddenly connected to

the input of the converter and solid tantalum capacitors

can fail catastrophically under these conditions. Be sure

to specify surge-tested capacitors.

Boost Converter: RSENSE Selection

The boost application in the Typical Applications section

has the location of the current sense resistor in series with

the inductor with one side referenced to VIN. This location

was chosen for two reasons. Firstly, the circulating current

is always monitored so in the case of an output overvoltage

or input overcurrent condition the main switch will skip

cycles to protect the circuitry. Secondly, the VIN node can

be considered low noise since it is heavily filtered and the

input current is not pulsed but continuous.

In the case where the input voltage exceeds the voltage

limits on the LT3844 Sense pins, the sense resistor can

be moved to the source of the MOSFET. In both cases the

resistor value is the calculated using the same formula.

The LT3844 current comparator has a maximum threshold

of 100mV/RSENSE. The current comparator threshold sets

the peak of the inductor current. Allowing adequate margin

for ripple current and external component tolerances,

RSENSE can be calculated as follows:

RSENSE =

70mV

IL(MAX)

Where IL(MAX) is the maximum average inductor current

as calculated in the Boost Converter: Inductor Selection

section.

LT3844

3844fc

For more information www.linear.com/LT3844

Typical applicaTions

All Ceramic Capacitor Application, 24V to 3.3V at 5A, fSW = 250kHz

VIN

SHDN

CSS

BURST_EN

VFB

SYNC

fSET

BOOST

VCC

PGND

SENSE+

SENSE–

SGND

LT3844

3844 TA02

63.4k

OUT

3.3V

680pF

100pF

CIN

22µF

×3

10k

2200pF

0.22µF

2.2µF

6.8µH

COUT

100µF

×2

RSENSE

0.01Ω

3.32k

5.62k

IN4148

24V

L1 = VISHAY, IHLP5050FD-01

M1 = VISHAY, SI7852DP

D1 = DIODES INC, PDS760

COUT = TDK, C4532X5R0J107K

CIN = TDK, C4532X7R2A225K

(VOLTAGE

TRANSIENTS

UP TO 60V)

8V to 20V to 8V, 25W SEPIC Application

VIN

SHDN

CSS

BURST_EN

VFB

SYNC

fSET

BOOST

VCC

PGND

SENSE+

SENSE–

SGND

LT3844

3844 TA03

49.9k

OUT

25W

100pF

680pF

CIN2

1µF

25V

CIN1

22µF

25V

×3

3300pF

22µF

25V

×3

1µF

25V

10k

54.9k

RSENSE

0.01Ω

40.2k

10Ω

12V

L1 = COILTRONICS, VERSAPAC VP5-0083

CIN, C5, COUT2 = TDK, C4532X7R1E226M

D2 = ONSEMI, MBRD660

COUT = SANYO OS-CON, 16SVP330M

= VISHAY, Si7852DP

56pF

OUT2

22µF

25V

COUT1

330µF

16V

•

LT3844

3844fc

For more information www.linear.com/LT3844

Typical applicaTions

Two Phase Spread Spectrum 24V Input to 12V, 6A Output

VIN

SHDN

CSS

BURST_EN

VFB

SYNC

fSET

BOOST

VCC

PGND

SENSE+

SENSE–

SGND

LT3844

49.9k

R21

49.9k

R22

10k

OUT

12V

680pF

C13

47pF

4.99k

2200pF

0.22µF

16V

2.2µF

15µH

COUT

22µF

25V

RSENSE

0.02Ω

10k

CIN

6.8µF

50V

×3

87.5k

270k

R13

R16

270k

D1a

BAV70

D11a

BAV70

D1b

BAV70

VIN

SHDN

CSS

BURST_EN

VFB

SYNC

fSET

BOOST

VCC

PGND

SENSE+

SENSE–

SGND

LT3844

3844 TA04

R15

49.9k

C11

2200pF

C15

0.22µF

16V

C14

2.2µF

15µH

RSENSE

0.02Ω

R11

10k

R12

87.5k

D11

M11

L1, L11 = VISHAY, IHLP5050FD-01

M1, M11 = VISHAY, Si7850DP

D1, D11 = DIODES INC, PDS760

COUT = TDK, C4532X7R1E226K

CIN = TDK, C4532X7R1H685K

V+

DIV

OUT1

SET

MOD

GND

OUT2

OUT VIN

GND

LTC6902

LT1121-5

VIN

18V

36V

D11b

BAV70

SYNC1

SYNC2

SYNC1

SYNC2

LT3844

3844fc

For more information www.linear.com/LT3844

package DescripTion

FE Package

16-Lead Plastic TSSOP (4.4mm)

(Reference LTC DWG # 05-08-1663 Rev K)

Exposed Pad Variation BC

FE16 (BC) TSSOP REV K 1013

0.09 – 0.20

(.0035 – .0079)

0° – 8°

0.25

REF

0.50 – 0.75

(.020 – .030)

4.30 – 4.50*

(.169 – .177)

1 3 4 5678

DETAIL B IS THE PART OF

THE LEAD FRAME FEATURE

FOR REFERENCE ONLY

NO MEASUREMENT PURPOSE

4.90 – 5.10*

(.193 – .201)

16 1514 13 12 11

1.10

(.0433)

MAX

0.05 – 0.15

(.002 – .006)

0.65

(.0256)

BSC

2.94

(.116)

0.48

(.019)

REF

0.51

(.020)

REF

0.195 – 0.30

(.0077 – .0118)

TYP

RECOMMENDED SOLDER PAD LAYOUT

0.45 ±0.05

0.65 BSC

4.50 ±0.10

6.60 ±0.10

1.05 ±0.10

2.94

(.116)

3.58

(.141)

3.58

(.141)

MILLIMETERS

(INCHES)

*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.150mm (.006") PER SIDE

NOTE:

1. CONTROLLING DIMENSION: MILLIMETERS

2. DIMENSIONS ARE IN

3. DRAWING NOT TO SCALE

4. RECOMMENDED MINIMUM PCB METAL SIZE

FOR EXPOSED PAD ATTACHMENT

SEE NOTE 4

SEE NOTE 5

5. BOTTOM EXPOSED PADDLE MAY HAVE METAL

PROTRUSION IN THIS AREA. THIS REGION MUST

BE FREE OF ANY EXPOSED TRACES OR VIAS ON

PCB LAYOUT

6.40

(.252)

BSC

FE Package

16-Lead Plastic TSSOP (4.4mm)

(Reference LTC DWG # 05-08-1663 Rev K)

Exposed Pad Variation BC

DETAIL B

4.60

Please refer to http://www.linear.com/product/LT3844#packaging for the most recent package drawings.

LT3844

3844fc

For more information www.linear.com/LT3844

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

revision hisTory

REV DATE DESCRIPTION PAGE NUMBER

C 04/16 Modified the Block Diagram

Changed lower schematic C5 value

(Revision history begins at Rev C)

LT3844

3844fc

For more information www.linear.com/LT3844

 LINEAR TECHNOLOGY CORPORATION 2006

LT 0416 REV C • PRINTED IN USA

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LT3844

relaTeD parTs

Typical applicaTion

12V to 48V 50W Step-Up Converter with 400kHz Switching Frequency

40k

383k

10k

VIN

CSS

BURST_EN

VFB

SYNC

fSET

BOOST

VCC

PGND

SENSE+

SENSE–

SGND

3844 TA05

LT3844

SHDN

4700pF

120pF

2.2µF

25V

4700pF

0.1µF

25V

COUT1

330µF

COUT2

220µF

M1 = VISHAY, Si7370DP

L1 = VISHAY, IHLP5050FD-01

D2 = DIODES INC., PDS560

CIN = SANYO, 25SVP33M

COUT1 = SANYO, 63CE220FST

COUT2 = TDK, C4532X7R2A225K

SENSE = IRC, LRF2512-01-R010-F

CIN

33µF

25V

×2

VIN

12V

SENSE

0.01Ω

6.8µH

VOUT

48V

50W

BAV99

4.7M

33.2k

PART NUMBER DESCRIPTION COMMENTS

LT3840 Wide Input Range Synchronous Regulator Controller

with Integrated Buck-Boost Bias Voltage requlator

Synchronizable Fixed Frequency 100kHz to 600kHz, 2.5V ≤ VIN ≤ 60V,

1.23V ≤ VOUT ≤ 60V, IQ = 75µA, TSSOP-28, 4mm × 6mm QFN-38

LT3845A 60V, Low IQ, Single Output Synchronous Step-Down

DC/DC Controller

Synchronizable Fixed Frequency 100kHz to 600kHz, 4V ≤ VIN ≤ 60V,

1.23V ≤ VOUT ≤ 36V, IQ = 120µA, TSSOP-16

LT3800 60V, Low IQ, Single Output Synchronous Step-Down

DC/DC Controller

Fixed 200kHz Frequency, 4V≤ VIN ≤ 60V, 1.23V ≤ VOUT ≤ 36V, IQ = 120µA,

TSSOP-16

LT C

3891 60V, Low IQ, Synchronous Step-Down DC/DC Controller Phase-Lockable Fixed Frequency, 50kHz to 900kHz 4V ≤ VIN ≤ 60V,

0.8V ≤ VOUT ≤ 24V, IQ = 50µA

LTC3864 60V, Low IQ, Step-Down DC/DC Controller 100% Duty

Cycle Capability

Selectable Fixed Frequency, 200kHz to 600kHz 3.5V≤ VIN ≤ 60V,

0.8V ≤ VOUT ≤ VIN, IQ = 40µA, MSOP-10E

LTC3892 60V, Low IQ, Dual 2-Phase Synchronous Step-Down

DC/DC Controller with Adjustable Gate Drive Voltage

Phase-Lockable Fixed Frequency, 50kHz to 900kHz, 4V ≤ VIN ≤ 60V,

0.8V ≤ VOUT ≤ 0.99VIN, IQ = 29µA

LT8709 Negative Input Synchronous Multi-Topology DC/DC

Control

–80V ≤ VIN ≤ –4.5V, Up to 400kHz Programmable Operating Frequency,

TSSOP-20

LT8710 Synchronous SEPIC/Inverting/Boost Controller with

Output Current Control

4.5V ≤ VIN ≤ 80V, 100kHz to 1MHz Programmable Operating Frequency,

TSSOP-20

LT8705 80V VIN and VOUT Synchronous 4-Switch Buck- Boost

DC/DC Controller

2.8V ≤ VIN ≤ 80V, 100kHz to 400kHz Programmable Operating Frequency,

5mm × 7mm QFN-38 and TSSOP-38