LM4960
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LM4960 Piezoelectric Speaker Driver
Check for Samples: LM4960
1FEATURES DESCRIPTION
The LM4960 utilizes a switching regulator to drive a
23• Low Current Shutdown Mode dual audio power amplifier. It delivers 24VP-P mono-
• "Click and Pop" Suppression Circuitry BTL to a ceramic speaker with less than 1.0%
• Low Quiescent Current THD+N while operating on a 3.0V power supply.
• Unity-Gain Stable Audio Amplifiers The LM4960's switching regulator is a current-mode
• External Gain Configuration Capability boost converter operating at a fixed frequency of
1.6MHz.
• Thermal Shutdown Protection Circuitry Boomer™ audio power amplifiers were designed
• Wide Input Voltage Range (3.0V - 7V) specifically to provide high quality output power with a
• 1.6MHz Switching Frequency minimal amount of external components. The
LM4960 does not require output coupling capacitors
APPLICATIONS or bootstrap capacitors, and therefore is ideally suited
• Mobile Phone for mobile phone and other low voltage applications
where minimal power consumption is a primary
• PDA's requirement.
KEY SPECIFICATIONS The LM4960 features a low-power consumption
externally controlled micropower shutdown mode.
• VOUT @ VDD = 3.0 THD+N ≤1%: 24 VP-P (typ) Additionally, the LM4960 features and internal
• Power Supply Range: 3.0 to 7 V thermal shutdown protection mechanism along with a
short circuit protection.
• Switching Frequency: 1.6 MHz (typ)
The LM4960 is unity-gain stable and can be
configured by external gain-setting resistors.
Connection Diagram
Figure 1. 28-Pin WQFN (Top View)
See RSG0028A Package
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2Boomer is a trademark of Texas Instruments.
3All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 2004–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Bypass
Amp Shutdown
Reg Shutdown
GND
VDD SW
FB
V1
GND
20k
200k
82p
0.039 PF
Audio In
0.22 PF
4.7PF
L1
10 PH
4.7 PF
13k
115k
4.7 PF
230 pF
VDD
150k
VIN B
10
10
Ceramic
Speaker
S/D 6
1
7,12,24,25
20
2
23
28
7,12,24,25
27
8
4 11
20k
20k
VOUT A
VOUT B
VIN A
21
800 nF
LM4960
SNAS221C –OCTOBER 2004–REVISED MAY 2013
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Typical Application
Figure 2. Typical Audio Amplifier Application Circuit
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
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ABSOLUTE MAXIMUM RATINGS (1)(2)(3)
Supply Voltage (VDD) 8.5V
Supply Voltage (V1)
(Pin 27 referred to GND) 18V
Storage Temperature −65°C to +150°C
Input Voltage −0.3V to VDD + 0.3V
Power Dissipation (4) Internally limited
ESD Susceptibility (5) 2000V
ESD Susceptibility (6) 200V
Junction Temperature 150°C
Thermal Resistance θJA (WQFN) °C/W
See SNOA401 'Leadless Leadframe Packaging (LLP).'
(1) All voltages are measured with respect to the GND pin, unless otherwise specified.
(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which ensure specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not ensured for parameters where no limit is given, however, the typical value is a good indication
of device performance.
(3) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
(4) The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX,θJA, and the ambient temperature,
TA. The maximum allowable power dissipation is PDMAX = (TJMAX −TA) / θJA or the given in Absolute Maximum Ratings, whichever is
lower. For the LM4960 typical application (shown in Figure 2) with VDD = 12V, RL= 4Ωstereo operation the total power dissipation is
3.65W. θJA = 35°C/W.
(5) Human body model, 100pF discharged through a 1.5kΩresistor.
(6) Machine Model, 220pF–240pF discharged through all pins.
OPERATING RATINGS
Temperature Range TMIN ≤TA≤TMAX −40°C ≤TA≤+85°C
Supply Voltage (VDD) 3.0V ≤VDD ≤7V
Supply Voltage (V1) 9.6V ≤V1≤16V
ELECTRICAL CHARACTERISTICS VDD = 3.0V (1)(2)
The following specifications apply for VDD = 3V, AV= 10, RL= 800nF+20Ω, V1 = 12V unless otherwise specified. Limits apply
for TA= 25°C. LM4960 Units
Symbol Parameter Conditions (Limits)
Typical (3) Limit (4)(5)
IDD Quiescent Power Supply Current VIN = GND, No Load 85 150 mA (max)
ISD Shutdown Current VSHUTDOWN = GND (6) 30 100 µA (max)
VOS Output Offset Voltage 5 40 mV (max)
VSDIH Shutdown Voltage Input High 2 V (max)
VSDIL Shutdown Voltage Input Low 0.4 V (min)
TWU Wake-up Time CB= 0.22µF 50 ms
150 °C (min)
TSD Thermal Shutdown Temperature 170 190 °C (max)
VOOutput Voltage THD = 1% (max); f = 1kHz 24 20 VP-P (min)
(1) All voltages are measured with respect to the GND pin, unless otherwise specified.
(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which ensure specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not ensured for parameters where no limit is given, however, the typical value is a good indication
of device performance.
(3) Typicals are measured at 25°C and represent the parametric norm.
(4) Limits are ensured to AOQL (Average Outgoing Quality Level).
(5) Datasheet min/max specification limits are ensured by design, test, or statistical analysis.
(6) Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to VDD for
minimum shutdown current.
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ELECTRICAL CHARACTERISTICS VDD = 3.0V (1)(2) (continued)
The following specifications apply for VDD = 3V, AV= 10, RL= 800nF+20Ω, V1 = 12V unless otherwise specified. Limits apply
for TA= 25°C. LM4960 Units
Symbol Parameter Conditions (Limits)
Typical (3) Limit (4)(5)
THD+N Total Harmomic Distortion + Noise VO= 3Wrms; f = 1kHz 0.04 %
εOS Output Noise A-Weighted Filter, VIN = 0V 90 µV
PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 1kHz 55 50 dB (min)
VFB Feedback Pin Reference Voltage 1.23 V (max)
ELECTRICAL CHARACTERISTICS VDD = 5.0V (1)(2)
The following specifications apply for VDD = 5V, AV= 10, RL= 800nF+20Ωunless otherwise specified. Limits apply for TA=
25°C.Symbol Parameter Conditions LM4960 Units
(Limits)
Typical (3) Limit (4) (5)
IDD Quiescent Power Supply Current VIN = GND, No Load 45 mA (max)
ISD Shutdown Current VSHUTDOWN = GND (6) 55 100 µA (max)
VSDIH Shutdown Voltage Input High 2 V (max)
VSDIL Shutdown Voltage Input Low 0.4 V (min)
TWU Wake-up Time CB= 0.22µF 50 ms
150 °C (min)
TSD Thermal Shutdown Temperature 170 190 °C (max)
THD = 1% (max); f = 1kHz
VOOutput Voltage 24 20 VP-P (min)
RL= Ceramic Speaker
THD+N Total Harmomic Distortion + Noise VO= 3Wrms; f = 1kHz 0.04 %
εOS Output Noise A-Weighted Filter, VIN = 0V 90 µV
PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 1kHz 60 dB (min)
VFB Feedback Pin Reference Voltage 1.23 V (max)
(1) All voltages are measured with respect to the GND pin, unless otherwise specified.
(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which ensure specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not ensured for parameters where no limit is given, however, the typical value is a good indication
of device performance.
(3) Typicals are measured at 25°C and represent the parametric norm.
(4) Limits are ensured to AOQL (Average Outgoing Quality Level).
(5) Datasheet min/max specification limits are ensured by design, test, or statistical analysis.
(6) Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to VDD for
minimum shutdown current.
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20 200 2k 20k
0.001
0.01
0.1
1
10
THD+N (%)
FREQUENCY (Hz)
20 200 2k 20k
0.001
0.01
0.1
1
10
THD+N (%)
FREQUENCY (Hz)
20 200 2k 20k
0.001
0.01
0.1
1
10
THD+N (%)
FREQUENCY (Hz)
20 200 2k 20k
0.001
0.01
0.1
1
10
THD+N (%)
FREQUENCY (Hz)
20 200 2k 20k
0.001
0.01
0.1
1
10
THD+N (%)
FREQUENCY (Hz)
20 200 2k 20k
0.001
0.01
0.1
1
10
THD+N (%)
FREQUENCY (Hz)
LM4960
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SNAS221C –OCTOBER 2004–REVISED MAY 2013
TYPICAL PERFORMANCE CHARACTERISTICS
THD+N vs Frequency THD+N vs Frequency
VDD = 3V, V1= 9.6V, V0= 3Vrms snas2202906 VDD = 3V, V1= 12V, V0= 3Vrms
Figure 3. Figure 4.
THD+N vs Frequency THD+N vs Frequency
VDD = 3V, V1= 15V, V0= 3Vrms VDD = 5V, V1= 9.6V, V0= 3Vrms
Figure 5. Figure 6.
THD+N vs Frequency THD+N vs Frequency
VDD = 5V, V1=12V, V0= 3Vrms VDD = 5V, V1=15V, V0= 3Vrms
Figure 7. Figure 8.
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2.83 9.05 21.5 33.94
0.001
0.01
0.1
1
10
THD+N (%)
OUTPUT VOLTAGE (Vrms)
27.7215.27
2.83 9.05 21.5 33.94
0.001
0.01
0.1
1
10
THD+N (%)
OUTPUT VOLTAGE (Vrms)
27.7215.27
2.83 9.05 21.5 33.94
0.001
0.01
0.1
1
10
THD+N (%)
OUTPUT VOLTAGE (Vrms)
27.7215.27
2.83 9.05 21.5 33.94
0.001
0.01
0.1
1
10
THD+N (%)
OUTPUT VOLTAGE (Vrms)
27.7215.27
2.83 9.05 21.5 33.94
0.001
0.01
0.1
1
10
THD+N (%)
OUTPUT VOLTAGE (Vrms)
27.7215.27
2.83 9.05 21.5 33.94
0.001
0.01
0.1
1
10
THD+N (%)
OUTPUT VOLTAGE (Vrms)
27.7215.27
LM4960
SNAS221C –OCTOBER 2004–REVISED MAY 2013
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
THD+N vs Output Power THD+N vs Output Power
VDD = 3V, V1= 9.6V, VDD = 3V, V1= 12V,
f = 100Hz, 1kHz, 10kHz f = 100Hz, 1kHz, 10kHz
Figure 9. Figure 10.
THD+N vs Output Power THD+N vs Output Power
VDD = 3V, V1= 15V, VDD = 5V, V1= 9.6V,
f = 100Hz, 1kHz, 10kHz f = 100Hz, 1kHz, 10kHz
Figure 11. Figure 12.
THD+N vs Output Power THD+N vs Output Power
VDD = 5V, V1= 12V, VDD = 5V, V1= 15V,
f = 100Hz, 1kHz, 10kHz f = 100Hz, 1kHz, 10kHz
Figure 13. Figure 14.
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20 200 2k 20k
-100
-80
-60
-40
-20
0
PSRR (dB)
FREQUENCY (Hz)
23 6 8
50
150
300
350
400
SUPPLY CURRENT (mA)
SUPPLY VOLTAGE (V)
250
200
100
10
054 97
20 200 2k 20k
-100
-80
-60
-40
-20
0
PSRR (dB)
FREQUENCY (Hz)
010 20 30
100
300
600
700
800
POWER DISSIPATION (mW)
OUTPUT VOLTAGE (Vp-p)
500
400
200
40
0
0 10 20 30
0
400
1000
1200
1400
POWER DISSIPATION (mW)
OUTPUT VOLTAGE (Vp-p)
800
600
200
40
LM4960
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SNAS221C –OCTOBER 2004–REVISED MAY 2013
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Power Dissipation vs Output Voltage Power Dissipation vs Output Voltage
VDD = 3V, from top to bottom: VDD = 5V, from top to bottom:
V1= 15V, V1= 12V, V1= 9.6V V1= 15V, V1= 12V, V1= 9.6V
Figure 15. Figure 16.
Supply Current vs Supply Voltage
from top to bottom: Power Supply Rejection Ratio
VDD = 15V, VDD = 12V, VDD = 9.6V VDD = 3V
Figure 17. Figure 18.
Power Supply Rejection Ratio
VDD = 5V
Figure 19.
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APPLICATION INFORMATION
BRIDGE CONFIGURATION EXPLANATION
The Audio Amplifier portion of the LM4960 has two internal amplifiers allowing different amplifier configurations.
The first amplifier’s gain is externally configurable, whereas the second amplifier is internally fixed in a unity-gain,
inverting configuration. The closed-loop gain of the first amplifier is set by selecting the ratio of Rf to Ri while the
second amplifier’s gain is fixed by the two internal 20kΩresistors. Figure 2 shows that the output of amplifier one
serves as the input to amplifier two. This results in both amplifiers producing signals identical in magnitude, but
out of phase by 180°. Consequently, the differential gain for the Audio Amplifier is
AVD = 2 *(Rf/Ri) (1)
By driving the load differentially through outputs Vo1 and Vo2, an amplifier configuration commonly referred to as
“bridged mode” is established. Bridged mode operation is different from the classic single-ended amplifier
configuration where one side of the load is connected to ground.
A bridge amplifier design has a few distinct advantages over the single-ended configuration. It provides
differential drive to the load, thus doubling the output swing for a specified supply voltage. Four times the output
power is possible as compared to a single-ended amplifier under the same conditions. This increase in attainable
output power assumes that the amplifier is not current limited or clipped.
The bridge configuration also creates a second advantage over single-ended amplifiers. Since the differential
outputs, Vo1 and Vo2, are biased at half-supply, no net DC voltage exists across the load. This eliminates the
need for an output coupling capacitor which is required in a single supply, single-ended amplifier configuration.
Without an output coupling capacitor, the half-supply bias across the load would result in both increased internal
IC power dissipation and also possible loudspeaker damage.
AMPLIFIER POWER DISSIPATION
Power dissipation is a major concern when designing a successful amplifier, whether the amplifier is bridged or
single-ended. A direct consequence of the increased power delivered to the load by a bridge amplifier is an
increase in internal power dissipation. Since the amplifier portion of the LM4960 has two operational amplifiers,
the maximum internal power dissipation is 4 times that of a single-ended amplifier. The maximum power
dissipation for a given BTL application can be derived from Equation (2).
PDMAX(AMP) = 4(VDD)2/ (2π2ZL)
where
• ZL= Ro1 + Ro2 +1/2πfc (2)
BOOST CONVERTER POWER DISSIPATION
At higher duty cycles, the increased ON-time of the switch FET means the maximum output current will be
determined by power dissipation within the LM2731 FET switch. The switch power dissipation from ON-time
conduction is calculated by Equation (3).
PDMAX(SWITCH) = DC x IIND(AVE)2x RDS(ON)
where
• DC is the duty cycle (3)
There will be some switching losses as well, so some derating needs to be applied when calculating IC power
dissipation.
TOTAL POWER DISSIPATION
The total power dissipation for the LM4960 can be calculated by adding Equation (2) and Equation (3) together
to establish Equation (4):
PDMAX(TOTAL) = [4*(VDD)2/2π2ZL] + [DC x IIND(AVE)2xRDS(ON)] (4)
The result from Equation (4) must not be greater than the power dissipation that results from Equation (5):
PDMAX = (TJMAX - TA) / θJA (5)
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For the LQA28A, θJA = 59°C/W. TJMAX = 125°C for the LM4960. Depending on the ambient temperature, TA, of
the system surroundings, Equation (5) can be used to find the maximum internal power dissipation supported by
the IC packaging. If the result of Equation (4) is greater than that of Equation (5), then either the supply voltage
must be increased, the load impedance increased or TAreduced. For the typical application of a 3V power
supply, with V1 set to 12V and a 800nF + 20Ωload, the maximum ambient temperature possible without violating
the maximum junction temperature is approximately 118°C provided that device operation is around the
maximum power dissipation point. Thus, for typical applications, power dissipation is not an issue. Power
dissipation is a function of output power and thus, if typical operation is not around the maximum power
dissipation point, the ambient temperature may be increased accordingly. Refer to the TYPICAL
PERFORMANCE CHARACTERISTICS curves for power dissipation information for lower output levels.
EXPOSED-DAP PACKAGE PCB MOUNTING CONSIDERATIONS
The LM4960’s exposed-DAP (die attach paddle) package (WQFN) provides a low thermal resistance between
the die and the PCB to which the part is mounted and soldered. The low thermal resistance allows rapid heat
transfer from the die to the surrounding PCB copper traces, ground plane, and surrounding air. The WQFN
package should have its DAP soldered to a copper pad on the PCB. The DAP’s PCB copper pad may be
connected to a large plane of continuous unbroken copper. This plane forms a thermal mass, heat sink, and
radiation area. Further detailed and specific information concerning PCB layout, fabrication, and mounting a
WQFN package is found in Texas Instruments' Package Engineering Group under application note SNOA401.
SHUTDOWN FUNCTION
In many applications, a microcontroller or microprocessor output is used to control the shutdown circuitry to
provide a quick, smooth transition into shutdown. Another solution is to use a single-pole, single-throw switch,
and a pull-up resistor. One terminal of the switch is connected to GND. The other side is connected to the two
shutdown pins and the terminal of the pull-up resistor. The remaining resistance terminal is connected to VDD. If
the switch is open, then the external pull-up resistor connected to VDD will enable the LM4960. This scheme
ensures that the shutdown pins will not float thus preventing unwanted state changes.
PROPER SELECTION OF EXTERNAL COMPONENTS
Proper selection of external components in applications using integrated power amplifiers, and switching DC-DC
converters, is critical for optimizing device and system performance. Consideration to component values must be
used to maximize overall system quality.
The best capacitors for use with the switching converter portion of the LM4960 are multi-layer ceramic
capacitors. They have the lowest ESR (equivalent series resistance) and highest resonance frequency, which
makes them optimum for high frequency switching converters.
When selecting a ceramic capacitor, only X5R and X7R dielectric types should be used. Other types such as
Z5U and Y5F have such severe loss of capacitance due to effects of temperature variation and applied voltage,
they may provide as little as 20% of rated capacitance in many typical applications. Always consult capacitor
manufacturer’s data curves before selecting a capacitor. High-quality ceramic capacitors can be obtained from
Taiyo-Yuden, AVX, and Murata.
POWER SUPPLY BYPASSING
As with any amplifier, proper supply bypassing is critical for low noise performance and high power supply
rejection. The capacitor location on both V1 and VDD pins should be as close to the device as possible.
SELECTING INPUT CAPACITOR FOR AUDIO AMPLIFIER
One of the major considerations is the closedloop bandwidth of the amplifier. To a large extent, the bandwidth is
dictated by the choice of external components shown in Figure 2. The input coupling capacitor, Ci, forms a first
order high pass filter which limits low frequency response. This value should be chosen based on needed
frequency response for a few distinct reasons.
High value input capacitors are both expensive and space hungry in portable designs. Clearly, a certain value
capacitor is needed to couple in low frequencies without severe attenuation. But ceramic speakers used in
portable systems, whether internal or external, have little ability to reproduce signals below 100Hz to 150Hz.
Thus, using a high value input capacitor may not increase actual system performance.
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In addition to system cost and size, click and pop performance is affected by the value of the input coupling
capacitor, Ci. A high value input coupling capacitor requires more charge to reach its quiescent DC voltage
(nominally 1/2 VDD). This charge comes from the output via the feedback and is apt to create pops upon device
enable. Thus, by minimizing the capacitor value based on desired low frequency response, turn-on pops can be
minimized.
SELECTING BYPASS CAPACITOR FOR AUDIO AMPLIFIER
Besides minimizing the input capacitor value, careful consideration should be paid to the bypass capacitor value.
Bypass capacitor, CB, is the most critical component to minimize turn-on pops since it determines how fast the
amplifer turns on. The slower the amplifier’s outputs ramp to their quiescent DC voltage (nominally 1/2 VDD), the
smaller the turn-on pop. Choosing CBequal to 1.0µF along with a small value of Ci(in the range of 0.039µF to
0.39µF), should produce a virtually clickless and popless shutdown function. Although the device will function
properly, (no oscillations or motorboating), with CBequal to 0.1µF, the device will be much more susceptible to
turn-on clicks and pops. Thus, a value of CBequal to 1.0µF is recommended in all but the most cost sensitive
designs.
SELECTING FEEDBACK CAPACITOR FOR AUDIO AMPLIFIER
The LM4960 is unity-gain stable which gives the designer maximum system flexability. However, to drive ceramic
speakers, a typical application requires a closed-loop differential gain of 10. In this case a feedback capacitor
(Cf2) will be needed as shown in Figure 2 to bandwidth limit the amplifier.
This feedback capacitor creates a low pass filter that eliminates possible high frequency oscillations. Care should
be taken when calculating the -3dB frequency because an incorrect combination of Rfand Cf2 will cause rolloff
before the desired frequency
SELECTING OUTPUT CAPACITOR (CO) FOR BOOST CONVERTER
A single 4.7µF to 10µF ceramic capacitor will provide sufficient output capacitance for most applications. If larger
amounts of capacitance are desired for improved line support and transient response, tantalum capacitors can
be used. Aluminum electrolytics with ultra low ESR such as Sanyo Oscon can be used, but are usually
prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies above 500
kHz because of significant ringing and temperature rise due to self-heating from ripple current. An output
capacitor with excessive ESR can also reduce phase margin and cause instability.
In general, if electrolytics are used, we recommended that they be paralleled with ceramic capacitors to reduce
ringing, switching losses, and output voltage ripple.
SELECTING INPUT CAPACITOR (Cs1) FOR BOOST CONVERTER
An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each
time the switch turns ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We
recommend a nominal value of 4.7µF, but larger values can be used. Since this capacitor reduces the amount of
voltage ripple seen at the input pin, it also reduces the amount of EMI passed back along that line to other
circuitry.
SETTING THE OUTPUT VOLTAGE (V1) OF BOOST CONVERTER
The output voltage is set using the external resistors R1 and R2 (see Figure 2). A value of approximately 13.3kΩ
is recommended for R2 to establish a divider current of approximately 92µA. R1 is calculated using the formula:
R1 = R2 X (V2/1.23 −1) (6)
FEED-FORWARD COMPENSATION FOR BOOST CONVERTER
Although the LM4960's internal Boost converter is internally compensated, the external feed-forward capacitor Cf
is required for stability (see Figure 2). Adding this capacitor puts a zero in the loop response of the converter.
The recommended frequency for the zero fz should be approximately 6kHz. Cf1 can be calculated using the
formula:
Cf1 = 1 / (2 X R1 X fz) (7)
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SELECTING DIODES
The external diode used in Figure 2 should be a Schottky diode. A 20V diode such as the MBR0520 is
recommended.
The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications
exceeding 0.5A average but less than 1A, a Microsemi UPS5817 can be used.
DUTY CYCLE
The maximum duty cycle of the boost converter determines the maximum boost ratio of output-to-input voltage
that the converter can attain in continuous mode of operation. The duty cycle for a given boost application is
defined as:
Duty Cycle = VOUT + VDIODE - VIN/VOUT + VDIODE - VSW
This applies for continuous mode operation.
INDUCTANCE VALUE
The first question we are usually asked is: “How small can I make the inductor.” (because they are the largest
sized component and usually the most costly). The answer is not simple and involves trade-offs in performance.
Larger inductors mean less inductor ripple current, which typically means less output voltage ripple (for a given
size of output capacitor). Larger inductors also mean more load power can be delivered because the energy
stored during each switching cycle is:
E = L/2 X (lp)2
where
• “lp” is the peak inductor current. (8)
An important point to observe is that the LM4960 will limit its switch current based on peak current. This means
that since lp(max) is fixed, increasing L will increase the maximum amount of power available to the load.
Conversely, using too little inductance may limit the amount of load current which can be drawn from the output.
Best performance is usually obtained when the converter is operated in “continuous” mode at the load current
range of interest, typically giving better load regulation and less output ripple. Continuous operation is defined as
not allowing the inductor current to drop to zero during the cycle. It should be noted that all boost converters shift
over to discontinuous operation as the output load is reduced far enough, but a larger inductor stays “continuous”
over a wider load current range.
To better understand these trade-offs, a typical application circuit (5V to 12V boost with a 10µH inductor) will be
analyzed. We will assume:
VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V (9)
Since the frequency is 1.6MHz (nominal), the period is approximately 0.625µs. The duty cycle will be 62.5%,
which means the ON-time of the switch is 0.390µs. It should be noted that when the switch is ON, the voltage
across the inductor is approximately 4.5V. Using the equation:
V = L (di/dt) (10)
We can then calculate the di/dt rate of the inductor which is found to be 0.45 A/µs during the ON-time. Using
these facts, we can then show what the inductor current will look like during operation:
Figure 20. 10μH Inductor Current
5V - 12V Boost (LM4960)
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During the 0.390µs ON-time, the inductor current ramps up 0.176A and ramps down an equal amount during the
OFF-time. This is defined as the inductor “ripple current”. It can also be seen that if the load current drops to
about 33mA, the inductor current will begin touching the zero axis which means it will be in discontinuous mode.
A similar analysis can be performed on any boost converter, to make sure the ripple current is reasonable and
continuous operation will be maintained at the typical load current values.
MAXIMUM SWITCH CURRENT
The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the
application. This is illustrated in a graph in the TYPICAL PERFORMANCE CHARACTERISTICS section which
shows typical values of switch current as a function of effective (actual) duty cycle.
CALCULATING OUTPUT CURRENT OF BOOST CONVERTER (IAMP)
As shown in Figure 20 which depicts inductor current, the load current is related to the average inductor current
by the relation:
ILOAD = IIND(AVG) x (1 - DC)
where
• "DC" is the duty cycle of the application. (11)
The switch current can be found by:
ISW = IIND(AVG) + 1/2 (IRIPPLE) (12)
Inductor ripple current is dependent on inductance, duty cycle, input voltage and frequency:
IRIPPLE = DC x (VIN-VSW) / (f x L) (13)
Combining all terms, we can develop an expression which allows the maximum available load current to be
calculated:
ILOAD(max) = (1–DC)x(ISW(max)–DC(VIN-VSW))/fL (14)
The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF
switching losses of the FET and diode.
DESIGN PARAMETERS VSW AND ISW
The value of the FET "ON" voltage (referred to as VSW in Equation (11) through Equation (14)) is dependent on
load current. A good approximation can be obtained by multiplying the "ON Resistance" of the FET times the
average inductor current.
FET on resistance increases at VIN values below 5V, since the internal N-FET has less gate voltage in this input
voltage range (see TYPICAL PERFORMANCE CHARACTERISTICS curves). Above VIN = 5V, the FET gate
voltage is internally clamped to 5V.
The maximum peak switch current the device can deliver is dependent on duty cycle. For higher duty cycles, see
TYPICAL PERFORMANCE CHARACTERISTICS curves.
INDUCTOR SUPPLIERS
Recommended suppliers of inductors for the LM4960 include, but are not limited to Taiyo-Yuden, Sumida,
Coilcraft, Panasonic, TDK and Murata. When selecting an inductor, make certain that the continuous current
rating is high enough to avoid saturation at peak currents. A suitable core type must be used to minimize core
(switching) losses, and wire power losses must be considered when selecting the current rating.
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SNAS221C –OCTOBER 2004–REVISED MAY 2013
PCB LAYOUT GUIDELINES
High frequency boost converters require very careful layout of components in order to get stable operation and
low noise. All components must be as close as possible to the LM4802 device. It is recommended that a 4-layer
PCB be used so that internal ground planes are available.
Some additional guidelines to be observed:
1. Keep the path between L1, D1, and Co extremely short. Parasitic trace inductance in series with D1 and Co
will increase noise and ringing.
2. The feedback components R1, R2 and Cf1 must be kept close to the FB pin of U1 to prevent noise injection
on the FB pin trace.
3. If internal ground planes are available (recommended) use vias to connect directly to ground at pin 2 of U1,
as well as the negative sides of capacitors Cs1 and Co.
GENERAL MIXED-SIGNAL LAYOUT RECOMMENDATION
This section provides practical guidelines for mixed signal PCB layout that involves various digital/analog power
and ground traces. Designers should note that these are only "rule-of-thumb" recommendations and the actual
results will depend heavily on the final layout.
Power and Ground Circuits
For 2 layer mixed signal design, it is important to isolate the digital power and ground trace paths from the
analog power and ground trace paths. Star trace routing techniques (bringing individual traces back to a central
point rather than daisy chaining traces together in a serial manner) can have a major impact on low level signal
performance. Star trace routing refers to using individual traces to feed power and ground to each circuit or even
device. This technique will take require a greater amount of design time but will not increase the final price of the
board. The only extra parts required may be some jumpers.
Single-Point Power / Ground Connection
The analog power traces should be connected to the digital traces through a single point (link). A "Pi-filter" can
be helpful in minimizing high frequency noise coupling between the analog and digital sections. It is further
recommended to place digital and analog power traces over the corresponding digital and analog ground traces
to minimize noise coupling.
Placement of Digital and Analog Components
All digital components and high-speed digital signals traces should be located as far away as possible from
analog components and circuit traces.
Avoiding Typical Design / Layout Problems
Avoid ground loops or running digital and analog traces parallel to each other (side-by-side) on the same PCB
layer. When traces must cross over each other do it at 90 degrees. Running digital and analog traces at 90
degrees to each other from the top to the bottom side as much as possible will minimize capacitive noise
coupling and crosstalk.
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REVISION HISTORY
Changes from Revision B (May 2013) to Revision C Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 13
14 Submit Documentation Feedback Copyright © 2004–2013, Texas Instruments Incorporated
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PACKAGE OPTION ADDENDUM
www.ti.com 16-Oct-2015
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM4960SQ/NOPB LIFEBUY WQFN RSG 28 1000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM L4960SQ
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
PACKAGE OPTION ADDENDUM
www.ti.com 16-Oct-2015
Addendum-Page 2
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM4960SQ/NOPB WQFN RSG 28 1000 178.0 12.4 5.3 5.3 1.3 8.0 12.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 11-Oct-2013
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM4960SQ/NOPB WQFN RSG 28 1000 210.0 185.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 11-Oct-2013
Pack Materials-Page 2
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