LM48520TL/NOPB - NATIONAL SEMICONDUCTOR

LM48520

LM48520 Boosted Stereo Class D Audio Power Amplifier with Output

Speaker Protection and Spread Spectrum

Literature Number: SNAS367B

March 20, 2008

LM48520

Boosted Stereo Class D Audio Power Amplifier with Output

Speaker Protection and Spread Spectrum

General Description

The LM48520 integrates a boost converter with a high effi-

ciency Class D stereo audio power amplifier to provide up to

1W/ch continuous power into an 8Ω speaker when operating

from 2.7V to 5.0V power supply with boost voltage (PV1) of

5.0V. The LM48520 utilizes a proprietary spread spectrum

pulse width modulation technique that lowers RF interference

and EMI levels. The Class D amplifier is a low noise, filterless

PWM architecture that eliminates the output filter, reducing

external component count, board area, power consumption,

system cost, and simplifying design.

The LM48520 is designed for use in mobile phones and other

portable communication devices. The high (78%) efficiency

extends battery life when compared to Boosted Class AB am-

plifiers. The LM48520 features a low-power consumption

shutdown mode. Shutdown may be enabled by driving the

Shutdown pin to a logic low (GND). Also, external leakage is

minimized via control of the ground reference via the SW-

OUT pin .

The LM48520 has 4 gain options which are pin selectable via

Gain0 and Gain1 pins. Output short circuit prevents the de-

vice from damage during fault conditions. Superior click and

pop suppression eliminates audible transients during power-

up and shutdown.

Key Specifications

■ Quiescent Power Supply Current 11.5mA (typ)

■ Output Power

(RL = 8Ω, THD+N ≤ 1%,

VDD = 3.3V,PV1 = 5.0V) 1.1W (typ)

■ Shutdown Current 0.04μA (typ)

Features

■Click and Pop Suppression

■Low 0.04μA Shutdown Current

■78% Efficiency

■Filterless Class D

■2.7V - 5.0V operation

■4 Adjustable Gain settings

■Adjustable output swing limiter with Soft Clipping

■Speaker Protection

■Short circuit protection on Audio Amps

■Independent Boost and Amplifier shutdown pins

Applications

■Mobile Phones

■PDAs

■Portable media

■Cameras

■Handheld games

Boomer® is a registered trademark of National Semiconductor Corporation.

LM48520 Boosted Stereo Class D Audio Power Amplifier with Output Speaker Protection and

Spread Spectrum

Typical Application

20198701

FIGURE 1. Typical LM48520 Audio Amplifier Application Circuit

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LM48520

Connection Diagrams

LM48520TL

20198702

Top View

Order Number LM48520TL

See NS Package Number TLA25AAA

Micro SMD Marking

20198721

Top View

X — Date Code

T — Die Traceability

G — Boomer Family

I5 — LM48520TL

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LM48520

Pin Descriptions

Pin Designator Pin Name Pin Function

A1 VDD Power Supply

A2 BstFB Regulator Feedback Input. Connect BstFB to an external resistive voltage

divider to set the boost output voltage.

A3 Soft Start Soft start capacitor

A4 SW_GND Booster ground

A5 SW Drain of the Internal FET switch

B1 INR+ Non-inverting right channel input

B2 INR- Inverting right channel input

B3 FB_GND Ground return for R1, R2 resistor divider

B4 INL- Inverting left channel input

B5 INL+ Non-inverting left channel input

C1 V1 Amplifier supply voltage. Connect to PV1.

C2 BstSD Regulator active low shutdown

C3 GND Ground

C4 Gain0 Gain setting input 0

C5 PV1 Amplifier H-bridge power supply. Connect to V1.

D1 AmpSD Amplifier active low shutdown

D2 OUTR+ Non-inverting right channel output

D3 NC No connect

D4 OUTL+ Non-inverting left channel output

D5 Gain1 Gain setting input 1

E1 VLimit Set to control output clipping level

E2 OUTR- Inverting right channel output

E3 PGND Power ground

E4 OUTL- Inverting left channel output

E5 NC No connect

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LM48520

Absolute Maximum Ratings (Note 2)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Supply Voltage (VDD, V1)6V

Storage Temperature −65°C to +150°C

Input Voltage −0.3V to VDD + 0.3V

Power Dissipation (Note 3) Internally limited

ESD Susceptibility (Note 4) 2000V

ESD Susceptibility (Note 5) 200V

Junction Temperature 150°C

Thermal Resistance

θJA (TL) 40.5 °C/W

Operating Ratings

Temperature Range

TMIN ≤ TA ≤ TMAX −40°C ≤ TA ≤ +85°C

Supply Voltage (VDD) 2.7V ≤ VDD ≤ 5.0V

Amplifier Voltage (V1 )

Not under Boosted Condition 2.4V ≤ V1 ≤ 5.5V

Amplifier Voltage (PV1 )

Under Boosted Condition 3.0V ≤ PV1 ≤ 5.0V

Electrical Characteristics VDD = 3.3V (Notes 1, 2)

The following specifications apply for VDD = 3.3V, AV = 6dB, RL = 15µH + 8Ω +15µH, fIN = 1kHz, unless otherwise specified. Limits

apply for TA = 25°C, R1 = 40.2kΩ, R2 = 16.2kΩ, V1 = PV1 = 5V, Vlimit = GND. All electrical specifications are for amplifier and booster.

Symbol Parameter Conditions LM48520 Units

(Limits)

Typical

(Note 6)

Limit

(Notes 7, 8)

IDD Quiescent Power Supply Current

VIN = 0, RLOAD = ∞

VDD = 2.7V 14.8

VDD = 3.3V 11.5 15.5 mA (max)

VDD = 5.0V 8.0

ISD Shutdown Current VSHUTDOWN = GND 0.04 1.0 μA (max)

VSDIH Shutdown Voltage Input High For SD Boost, SD Amp 1.4 V

VSDIL Shutdown Voltage Input Low For SD Boost, SD Amp 0.4 V

TWU Wake-up Time Amplifier + Booster Wakeup 3 ms

VOS Output Offset Voltage 5 mV

AVGain

G0, G1 = GND

RL = ∞6 dB

G0 = VDD, G1 = GND

RL = ∞12 dB

G0 = GND, G1 = VDD

RL = ∞18 dB

G0, G1 = VDD

RL = ∞24 dB

POOutput Power

RL = 15μH + 8Ω + 15μH

THD+N = 1% (max),

f = 1kHz, 22kHz, BW

VDD = 3.3V 1.1 0.87 W (min)

RL = 15μH + 8Ω + 15μH

THD+N = 10% (max),

f = 1kHz, 22kHz, BW

VDD = 3.3V 1.3

THD+N Total Harmonic Distortion + Noise

PO = 500mW, f = 1kHz,

RL = 15μH + 8Ω + 15μH,

VDD = 3.3V

0.04

εOS Output Noise VDD = 3.6V, f = 20Hz – 20kHz

Inputs to AC GND, A weighted 32 µVRMS

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LM48520

Symbol Parameter Conditions LM48520 Units

(Limits)

Typical

(Note 6)

Limit

(Notes 7, 8)

PSRR Power Supply Rejection Ratio

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 217Hz 82 dB

VRIPPLE = 200mVP-P Sine,

fRIPPLE = = 1kHz 79 dB

CMRR Common Mode Rejection Ratio VRIPPLE = 1VP-P, fRIPPLE = 217Hz 67 dB

ηEfficiency

PO = 1W, f = 1kHz,

RL = 15μH + 8Ω + 15μH

VDD = 3.3V

VDD = 4.2V 78

VFB Feedback Pin Reference Voltage (Note 11) 1.24 V

Vout clipped

Output Voltage in clipped state with

soft clip activated

Vlimit = 2V, RL = 8Ω, VIN = 2VP

Vout clipped = 8/3 (PV1 – 2Vlimit)2.5 1.9

3.2

Vpk (min)

Vpk (max)

Note 1: All voltages are measured with respect to the GND pin, unless otherwise specified.

Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions

which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters

where no limit is given, however, the typical value is a good indication of device performance.

Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, θJA, and the ambient temperature, TA. The maximum

allowable power dissipation is PDMAX = (TJMAX − TA) / θJA or the given in Absolute Maximum Ratings, whichever is lower.

Note 4: Human body model, 100pF discharged through a 1.5kΩ resistor.

Note 5: Machine Model, 220pF–240pF discharged through all pins.

Note 6: Typicals are measured at 25°C and represent the parametric norm.

Note 7: Limits are guaranteed to National's AOQL (Average Outgoing Quality Level).

Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.

Note 9: Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to Vin for minimum shutdown

current.

Note 10: Shutdown current is measured with components R1 and R2 removed.

Note 11: Feedback pin reference voltage is measured with the Audio Amplifier disconnected from the Boost converter (the Boost converter is unloaded).

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LM48520

Typical Performance Characteristics

THD+N vs Frequency

VDD = 2.7V, POUT = 800mW, RL = 8Ω

20198703

THD+N vs Frequency

VDD = 3.3V, POUT = 900mW, RL = 8Ω

20198704

THD+N vs Frequency

VDD = 5.0V, POUT = 1W, RL = 8Ω

20198705

THD+N vs Output Power

VDD = 2.7V, RL = 8Ω

20198706

THD+N vs Output Power

VDD = 3.3V, RL = 8Ω

20198707

THD+N vs Output Power

VDD = 5.0V, RL = 8Ω

20198708

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LM48520

Power Dissipation vs Output Power

VDD = 2.7V, RL = 8Ω, f = 1kHz

20198712

Power Dissipation vs Output Power

VDD = 3.3V, RL = 8Ω, f = 1kHz

20198713

Power Dissipation vs Output Power

VDD = 5.0V, RL = 8Ω, f = 1kHz

20198714

Efficiency vs Output Power

VDD = 2.7V, RL = 8Ω, f = 1kHz

20198709

Efficiency vs Output Power

VDD = 3.3V, RL = 8Ω, f = 1kHz

20198710

Efficiency vs Output Power

VDD = 5.0V, RL = 8Ω, f = 1kHz

20198711

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LM48520

CMRR vs Frequency

VDD =3.3V, VRIPPLE = 1VP-P, RL = 8Ω

20198722

PSRR vs Frequency

VDD =3.3V, VRIPPLE = 200mVP-P, RL = 8Ω

20198723

Supply Current vs Supply Voltage

No Load

20198724

Output Power vs Supply Voltage

RL = 8Ω, f = 1kHz

20198725

Boost Output Voltage vs Load Current

VDD = 2.7V

20198726

Boost Output Voltage vs Load Current

VDD = 3.3V

20198727

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LM48520

Boost Output Voltage vs Load Current

VDD = 5.0V

20198728

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LM48520

Application Information

GENERAL AMPLIFIER FUNCTION

The LM48520 features a Class D audio power amplifier that

utilizes a filterless modulation scheme, reducing external

component count, conserving board space and reducing sys-

tem cost. The outputs of the device transition from PV1 to

GND with a 300kHz switching frequency. With no signal ap-

plied, the outputs (VLS+ and VLS-) switch with a 50% duty

cycle, in phase, causing the two outputs to cancel. This can-

cellation results in no net voltage across the speaker, thus

there is no current to the load in the idle state.

With the input signal applied, the duty cycle (pulse width) of

the LM48520 outputs changes. For increasing output voltage,

the duty cycle of VLS+ increases, while the duty cycle of VLS-

decreases. For decreasing output voltages, the converse

occurs. The difference between the two pulse widths yields

the differential output voltage.

DIFFERENTIAL AMPLIFIER EXPLANATION

The amplifier portion of the LM48520 is a fully differential am-

plifier that features differential input and output stages. A

differential amplifier amplifies the difference between the two

input signals. Traditional audio power amplifiers have typical-

ly offered only single-ended inputs resulting in a 6dB reduc-

tion in signal to noise ratio relative to differential inputs. The

amplifier also offers the possibility of DC input coupling which

eliminates the two external AC coupling, DC blocking capac-

itors. The amplifier can be used, however, as a single ended

input amplifier while still retaining it's fully differential benefits.

In fact, completely unrelated signals may be placed on the

input pins. The amplifier portion of the LM48520 simply am-

plifies the difference between the signals. A major benefit of

a differential amplifier is the improved common mode rejec-

tion ratio (CMRR) over single input amplifiers. The common-

mode rejection characteristic of the differential amplifier

reduces sensitivity to ground offset related noise injection,

especially important in high noise applications.

AMPLIFIER DISSIPATION AND EFFICIENCY

The major benefit of a Class D amplifier is increased efficiency

versus a Class AB. The efficiency of the LM48520 is attributed

to the region of operation of the transistors in the output stage.

The Class D output stage acts as current steering switches,

consuming negligible amounts of power compared to their

Class AB counterparts. Most of the power loss associated

with the output stage is due to the IR loss of the MOSFET on-

resistance, along with switching losses due to gate charge.

REGULATOR POWER DISSIPATION

At higher duty cycles, the increased ON-time of the switch

FET means the maximum output current will be determined

by power dissipation within the LM48520 FET switch. The

switch power dissipation from ON-time conduction is calcu-

lated by:

PD(SWITCH) = DC x (IINDUCTOR(AVE))2 x RDS(ON) (W) (1)

Where DC is the duty cycle.

SHUTDOWN FUNCTION

The LM48520 features independent amplifier and regulator

shutdown controls, allowing each portion of the device to be

disabled or enabled independently. AmpSD controls the

Class D amplifiers, while BstSD controls the regulator. Driving

either inputs low disables the corresponding portion of the

device, and reducing supply current.

When the regulator is disabled, both FB_GND switches open,

further reducing shutdown current by eliminating the current

path to GND through the regulator feedback network. With the

regulator disabled, there is still a current path from VDD,

through the inductor and diode, to the amplifier power supply.

This allows the amplifier to operate even when the regulator

is disabled. The voltage at PV1 and V1 will be:

VDD — [VD + (IL x DCR)] (2)

Where VD is the forward voltage of the Schottky diode, IL is

the current through the inductor, and DCR is the DC resis-

tance of the inductor. Additionally, when the regulator is dis-

abled, an external voltage between 2.4V and 5.5V can be

applied directly to PV1 and V1 to power the amplifier.

It is best to switch between ground and VDD for minimum cur-

rent consumption while in shutdown. The LM48520 may be

disabled with shutdown voltages in between GND and VDD,

the idle current will be greater than the typical 0.1µA value.

Increased THD+N may also be observed when a voltage of

less than VDD is applied to AmpSD.

PROPER SELECTION OF EXTERNAL COMPONENTS

Proper selection of external components in applications using

integrated power amplifiers, and switching DC-DC convert-

ers, is critical for optimizing device and system performance.

Consideration to component values must be used to maxi-

mize overall system quality.

The best capacitors for use with the switching converter por-

tion of the LM48520 are multi-layer ceramic capacitors. They

have the lowest ESR (equivalent series resistance) and high-

est resonance frequency, which makes them optimum for

high frequency switching converters.

When selecting a ceramic capacitor, only X5R and X7R di-

electric types should be used. Other types such as Z5U and

Y5F have such severe loss of capacitance due to effects of

temperature variation and applied voltage, they may provide

as little as 20% of rated capacitance in many typical applica-

tions. Always consult capacitor manufacturer’s data curves

before selecting a capacitor. High-quality ceramic capacitors

can be obtained from Taiyo-Yuden, AVX, and Murata.

POWER SUPPLY BYPASSING FOR AMPLIFIER

As with any amplifier, proper supply bypassing is critical for

low noise performance and high power supply rejection. The

capacitor location on both PV1, V1 and VDD pins should be

as close to the device as possible.

SELECTING INPUT CAPACITOR FOR AUDIO AMPLIFIER

Input capacitors, CIN, in conjunction with the input impedance

of the LM48520 forms a high pass filter that removes the DC

bias from an incoming signal. The AC-coupling capacitor al-

lows the amplifier to bias the signal to an optimal DC level.

Assuming zero source impedance, the -3dB point of the high

pass filter is given by:

f(–3dB) = 1/2πRINCIN (3)

Choose CIN such that f-3dB is well below that lowest frequency

of interest. Setting f-3dB too high affects the low-frequency re-

sponses of the amplifier. Use capacitors with low voltage

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LM48520

coefficient dielectrics, such as tantalum or aluminum elec-

trolytic. Capacitors with high-voltage coefficients, such as

ceramics, may result in increased distortion at low frequen-

cies. Other factors to consider when designing the input filter

include the constraints of the overall system. Although high

fidelity audio requires a flat frequency response between

20Hz and 20kHz, portable devices such as cell phones may

only concentrate on the frequency range of the frequency

range of the spoken human voice (typically 300Hz to 4kHz).

In addition, the physical size of the speakers used in such

portable devices limits the low frequency response; in this

case, frequencies below 150Hz may be filtered out.

SELECTING OUTPUT CAPACITOR (CO) FOR BOOST

CONVERTER

A single 100µF low ESR tantalum capacitor provides suffi-

cient output capacitance for most applications. Higher capac-

itor values improve line regulation and transient response.

Typical electrolytic capacitors are not suitable for switching

converters that operate above 500kHz because of significant

ringing and temperature rise due to self-heating from ripple

current. An output capacitor with excessive ESR reduces

phase margin and causes instability.

SELECTING INPUT CAPACITOR (Cs1) FOR BOOST

CONVERTER

An input capacitor is required to serve as an energy reservoir

for the current which must flow into the coil each time the

switch turns ON. This capacitor must have extremely low

ESR, so ceramic is the best choice. We recommend a nomi-

nal value of 2.2µF, but larger values can be used. Since this

capacitor reduces the amount of voltage ripple seen at the

input pin, it also reduces the amount of EMI passed back

along that line to other circuitry.

SELECTING SOFTSTART (CSS) CAPACITOR

The soft-start function charges the boost converter reference

voltage slowly. This allows the output of the boost converter

to ramp up slowly thus limiting the transient current at startup.

Selecting a soft-start capacitor (CSS) value presents a trade

off between the wake-up time and the startup transient cur-

rent. Using a larger capacitor value will increase wake-up time

and decrease startup transient current while the apposite ef-

fect happens with a smaller capacitor value. A general guide-

line is to use a capacitor value 1000 times smaller than the

output capacitance of the boost converter (CO). A 0.1uF soft-

start capacitor is recommended for a typical application.

SETTING THE OUTPUT VOLTAGE (V1) OF BOOST

CONVERTER

The output voltage is set using the external resistors R1 and

R2 (see Figure 1). A value of approximately 13.3kΩ is rec-

ommended for R2 to establish a divider current of approxi-

mately 92µA. R1 is calculated using the formula:

R1 = R2 X (V1/1.23 − 1) (4)

FEED-FORWARD COMPENSATION FOR BOOST

CONVERTER

Although the LM48520's internal Boost converter is internally

compensated, the external feed-forward capacitor Cf is re-

quired for stability (see Figure 1). Adding this capacitor puts

a zero in the loop response of the converter. The recom-

mended frequency for the zero fz should be approximately

6kHz. Cf1 can be calculated using the formula:

Cf = 1 / (2 X R1 X fz) (5)

SELECTING DIODES FOR BOOST

The external diode used in Figure 1 should be a Schottky

diode. A 20V diode such as the MBRS320T3 is recommend-

ed.

The MBRS320T3 series of diodes are designed to handle a

maximum average current of 3A.

DUTY CYCLE

The maximum duty cycle of the boost converter determines

the maximum boost ratio of output-to-input voltage that the

converter can attain in continuous mode of operation. The

duty cycle for a given boost application is defined as:

Duty Cycle = VOUT + VDIODE - VIN / VOUT + VDIODE - VSW

This applies for continuous mode operation.

SELECTING INDUCTOR VALUE

Inductor value involves trade-offs in performance. Larger in-

ductors reduce inductor ripple current, which typically means

less output voltage ripple (for a given size of output capacitor).

Larger inductors also mean more load power can be delivered

because the energy stored during each switching cycle is:

E = L/2 X (IP)2

Where “lp” is the peak inductor current. The LM48520 will limit

its switch current based on peak current. With IP fixed, in-

creasing L will increase the maximum amount of power avail-

able to the load. Conversely, using too little inductance may

limit the amount of load current which can be drawn from the

output. Best performance is usually obtained when the con-

verter is operated in “continuous” mode at the load current

range of interest, typically giving better load regulation and

less output ripple. Continuous operation is defined as not al-

lowing the inductor current to drop to zero during the cycle.

Boost converters shift over to discontinuous operation if the

load is reduced far enough, but a larger inductor stays con-

tinuous over a wider load current range.

During the TBDµs ON-time, the inductor current ramps up

TBDA and ramps down an equal amount during the OFF-

time. This is defined as the inductor “ripple current”. It can also

be seen that if the load current drops to about TBDmA, the

inductor current will begin touching the zero axis which means

it will be in discontinuous mode. A similar analysis can be

performed on any boost converter, to make sure the ripple

current is reasonable and continuous operation will be main-

tained at the typical load current values.

MAXIMUM SWITCH CURRENT

The maximum FET switch current available before the current

limiter cuts in is dependent on duty cycle of the application.

This is illustrated in a graph in the typical performance char-

acterization section which shows typical values of switch

current as a function of effective (actual) duty cycle.

CALCULATING OUTPUT CURRENT OF BOOST

CONVERTER (IAMP)

As shown in Figure 2 which depicts inductor current, the load

current is related to the average inductor current by the rela-

tion:

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LM48520

ILOAD = IIND(AVG) x (1 - DC) (6)

Where "DC" is the duty cycle of the application. The switch

current can be found by:

ISW = IIND(AVG) + 1/2 (IRIPPLE) (7)

Inductor ripple current is dependent on inductance, duty cy-

cle, input voltage and frequency:

IRIPPLE = DC x (VIN-VSW) / (f x L) (8)

combining all terms, we can develop an expression which al-

lows the maximum available load current to be calculated:

ILOAD(max) = (1–DC)x(ISW(max)–DC(VIN-VSW))/fL (9)

The equation shown to calculate maximum load current takes

into account the losses in the inductor or turn-OFF switching

losses of the FET and diode.

DESIGN PARAMETERS VSW AND ISW

The value of the FET "ON" voltage (referred to as VSW in

equations 4 thru 7) is dependent on load current. A good ap-

proximation can be obtained by multiplying the "ON Resis-

tance" of the FET times the average inductor current.

FET on resistance increases at VIN values below 5V, since

the internal N-FET has less gate voltage in this input voltage

range (see Typical Performance Characteristics curves).

Above VIN = 5V, the FET gate voltage is internally clamped to

5V.

The maximum peak switch current the device can deliver is

dependent on duty cycle. For higher duty cycles, see Typical

Performance Characteristics curves.

INDUCTOR SUPPLIERS

The recommended inductor for the LM48520 is the

NR8040T6R8N from Taiyo Yuden. When selecting an induc-

tor, make certain that the continuous current rating is high

enough to avoid saturation at peak currents, where:

IIND = (PV1 / VDD) x ILOAD(BOOST) (10)

A suitable core type must be used to minimize core (switch-

ing) losses, and wire power losses must be considered when

selecting the current rating.

PCB Layout Guidelines

High frequency boost converters require very careful layout

of components in order to get stable operation and low noise.

All components must be as close as possible to the LM48520

device. It is recommended that a four layer PCB be used so

that internal ground planes are available.

Some additional guidelines to be observed (all designators

are referencing Figure 1):

1. Keep the path between L1, D1, and Co extremely short.

Parasitic trace inductance in series with D1 and Co will in-

crease noise and ringing.

2. The feedback components R1, R2 and Cf1 must be kept

close to the FB pin to prevent noise injection on the FB pin

trace.

3. Since the external components of the boost converter

are switching, L1 and D1 should be kept away from the input

traces to prevent the noise from injecting into the input.

4. The power supply bypass capacitors, Cs1 and Cs2

should be placed as close to the LM48520 device as possible.

GROUNDING GUIDELINES

There are three grounds on the LM48520, GND, SW_GND,

and PGND. When laying out the PCB, it is critical to connect

the grounds as close to the device as possible. The simplest

way to do that is to place vias close to the GND, SW_GND,

and PGND bumps and connect the GND, SW_GND, and

PGND vias using a single ground plane in an inner layer of

the PCB.

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LM48520

Output Speaker Protection Function

The LM48520’s output voltage limiter can be used to set a

minimum and maximum output voltage swing magnitude. The

voltage applied to the VLimit pin controls the limit on the out-

put voltage level. The output level is determined by the fol-

lowing equation:

Vout clipped = 8/3 * (PV1 — 2 * Vlimit)

Vout clipped = 1/2 * (PV1 — 3/8 * Vout clipped)

Where, Vout clipped = the desired output level measured in

Vpk, PV1 = Boost output voltage, and Vlimit is the voltage

applied the the Vlimit pin on the LM48520.

To disable the limiter, set Vlimit = 0V.

Figure 2 provides an example of how the output voltage limiter

functions with VDD = 3.3V, AV = 6dB, PV1 = 5V, Vlimit = 2V,

RL = 8Ω, VIN = 2VP.

20198729

FIGURE 2. Soft Clipping vs No Clipping

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LM48520

Revision History

Rev Date Description

1.0 02/27/08 Initial release.

1.01 03/07/08 Added the Soft clipping vs No clipping curve.

1.02 03/12/08 Text edits.

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LM48520

Physical Dimensions inches (millimeters) unless otherwise noted

micro SMD Package

Order Number LM48520TL

NS Package Number TLA25AAA

X1 = 2.49mm, X2 = 2.49mm, X3 = 0.6mm,

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LM48520

Notes

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LM48520

Notes

LM48520 Boosted Stereo Class D Audio Power Amplifier with Output Speaker Protection and

Spread Spectrum

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