Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC THAT 4320 FEATURES APPLICATIONS * Pre-trimmed VCA & RMS detector * Companding noise reduction * Wide supply voltage range: 4.5V~16V * Wireless microphones * Low supply current: 3.7mA typ. (5V) * Wireless instrument packs * Wireless in-ear monitors * Four opamps * Battery operated dynamics processors * One low-noise opamp (<5nV/rt-Hz) * Compressors * On board PTAT reference * Limiters * Wide dynamic range: 120dB as compander * AGCs * De-essers Description technology for noise reduction. However, with 22 active pins, the part is extremely flexible and can be configured for a wide range of applications including single and multi-band companders, compressors, limiters, AGCs, de-essers, etc. The THAT4320 is a single-chip Analog Engine(R) optimized for low-voltage, low-power operation. Incorporating a high-performance voltage- controlled amplifier (VCA), RMS-level sensor, and four opamps, the surface mount part is aimed at battery-operated audio applications such as wireless microphones, wireless instruments and in-ear monitors. The 4320 operates from a single supply voltage down to +4.5Vdc, drawing only 3.7mA. What really sets the 4320 apart is the transparent sound of its Blackmer(R) VCA coupled with its accurate true-RMS level detector. The IC is useful in batterypowered mixers, compressor/limiters, ENG devices and other portable audio products. The part is highly integrated and requires minimal external support circuitry: it even contains an on-board PTAT (proportional to absolute temperature) voltage reference to This IC also works at supply voltages up to 16Vdc, making it useful in line-operated products as well. The VCA is pre-trimmed at wafer stage to deliver low distortion without further adjustment. And, one opamp is quiet enough to be used as a microphone preamp. generate thermally compensated control voltages for thresholds and gain settings. The part was developed specifically for use as a companding noise reduction system, drawing from THAT's long history and experience with dbx(R) 27 26 25 23 21 20 18 17 16 OUT IN EC+ 28 OA3 VCA EC- VEE VCC THAT 4320 15 VCC/2 Buffer 13 RMS OUT IN 1 CT 14 VPTAT 2 3 4 6 7 8 9 VREF GND 11 Figure 1. THAT4320 equivalent block diagram (QSOP-28 pin assignments shown) THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation; Document 600045 Rev 08 Document 600045 Rev 08 Page 2 of 16 THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC SPECIFICATIONS Absolute Maximum Ratings 1 Positive Supply Voltage (VCC) Supply Current (ICC) Operating Temperature Range (TOP) Junction Temperature (TJ) Output Short-Circuit Duration +18V 30mA -40 to +85 C -40 to +125 C 30 sec Power Dissipation (PD) at TA=85 C 400mW Input Voltage Supply Voltage -40 to +125 C Storage Temperature Range (TST) Lead Temperature Range (Soldering, 10 sec) 300 C Electrical Characteristics 2 Parameter Symbol Conditions Min Typ Max Units Positive Supply Voltage VCC Referenced to GND +4.5 - +16 V Negative Supply Voltage (OA1) VEE OA1 only VCC-16 0 0 V VPIN13 When overridden by split supply VCC - 8 VCC / 2 GND + 8 V ICC No Signal Power Supply Resistive Divider Voltage Supply Current IEE VCC=+5 V 3.7 6 mA VCC=+15 V 5 10 mA VCC=+5V, VEE=-5 V 0.6 - mA VCC = +5 V 500 Apeak VCC = +15 V 1 mApeak Voltage Controlled Amplifier (VCA) Max. I/O Signal Current iIN(VCA) + iOUT(VCA) Gain at 0V Control3 G0 0V at +IN of OA2 -1.5 0 +1.5 Gain-Control Constant EC+/Gain (dB) -60 dB < gain < +40 dB - 6.0 - mV/dB Gain-Control Tempco EC/TCHIP Ref TCHIP=27C - +0.33 - %/C Output Offset Voltage Change4 VOFF(OUT) Output Noise eN(OUT) ROUT = 20 k 0 dB gain - 1 15 mV +15 dB gain - 3 30 mV +30 dB gain - 10 50 mV - -98 -95 dBV 0.05 0.1 % 0 +8 mV iIN = 200 nA RMS 1 3 dB iIN = 1 mA RMS 1 3 dB 0 dB gain 22Hz~22kHz, RIN=ROUT=20 k Total Harmonic Distortion3 dB THD VIN= -5dBV, 1kHz, 0V at +IN of OA2 eO(0) iIN = 7.5 A RMS RMS Level Detector Output Voltage at Reference iIN Output Error at Input Extremes eO(RMS)error -8 1. If the devices are subjected to stress above the Absolute Maximum Ratings, permanent damage may result. Sustained operation at or near the Absolute Maximum Ratings conditions is not recommended. In particular, like all semiconductor devices, device reliability declines as operating temperature increases. 2. Unless otherwise noted, T A=25C, VCC=+5V, VEE=0 V. Test circuit is as shown in Figure 2. 3. Assumes OA 2 is configured for unity gain, & includes offset voltage of OA 2. 4. Reference is to output offset with -80 dB VCA gain. THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 3 of 16 THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC Electrical Characteristics (con't) 2 Parameter Symbol Scale Factor Match to VCA Conditions Min Typ Max Units .95 1 1.05 - -20 dB < VCA gain < +20 dB 1 A< iIN(RMS) < 100 A Rectifier Balance 7.5mA DCIN Timing Current IT Filtering Time Constant 1 - 7.5 dB - 3467 X CTIME Output Tempco EO/TCHIP Ref TCHIP = 27 C - Load Resistance RL -250mV < VOUTRMS< +250mV, re:Vref 2 Capacitive Load CL +0.33 A s - %/C k 150 pF Operational Amplifier OA15 Input Offset Voltage VOS - 1 3.5 mV Input Bias Current IB - 500 1200 nA Input Offset Current IOS - 30 120 nA VICR+ 4 4.3 - V VICR- - 0.4 0.6 V Input Common Mode Range Equivalent Input Noise Voltage eN(IN) f = 1 kHz - 4.5 6 nV/Hz Equivalent Input Noise Current iN(IN) f = 1 kHz - 0.9 - pA/Hz GBW f = 50 kHz - 13 - MHz Slew Rate SR G = +10, CL = 100 pF 2.3 4 - V/s Open Loop Gain AVOL RL = 10 k - 95 - dB Output Short Circuit Current ISC+ Output to VCC/2, VID = +0.4 V -2.3 -6.5 -20 mA ISC- Output to VCC/2, VID = -0.4 V 1.5 3.7 12 mA VO+ RL = 10 k to VCC/2, G = +10 Gain Bandwidth Product Output Voltage Range VCC-0.9 VCC-0.75 VEE+0.75 VEE+0.95 VOCapacitive Load Power Supply Rejection Ratio - CL PSRR +5 V < VCC-VEE < +15V V V 150 pF - 105 - dB VOS - 1.5 6 mV Input Bias Current IB - 450 1000 nA Input Offset Current IOS - 25 100 nA Input Common Mode Range VICR -1 +1 V Equivalent Input Noise Voltage eN(IN) f = 1 kHz - 8 - nV/Hz Equivalent Input Noise Current iN(IN) f = 1 kHz - 0.6 - pA/Hz GBW f = 50 kHz, CL= 100 nF, RL= 10 k - 0.012/CL - Hz SR G = +1 ISC/CL - V/s Operational Amplifier OA2 (Control Voltage Buffer) Input Offset Voltage Gain Bandwidth Product Slew Rate THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 4 of 16 THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC Electrical Characteristics (con't) 2 Parameter Open Loop Gain Output Short Circuit Current Power Supply Rejection Ratio Capacitive Load 6 Symbol Conditions Min Typ Max Units AVOL RL = 10 k - 57.5 dB 20*log(.075*RI) dB ISC+ Output to VCC/2, VID = +0.4 V - -4 - mA ISC- Output to VCC/2, VID = -0.4 V - 2.7 - mA PSRR +5 V < VCC < +15 V - 88 - dB CL 22 nF Operational Amplifier OA3 (VCA Current-to-Voltage Converter) Input Offset Voltage VOS 1.5 mV 200 nA Input Bias Current IB - Input Offset Current IOS Only one input is accessible Input Common Mode Range VICR Equivalent Input Noise Voltage eN(IN) f = 1 kHz - 10.5 - nV/Hz Equivalent Input Noise Current iN(IN) f = 1 kHz - 0.3 - pA/Hz GBW f = 50 kHz - 7.3 - MHz Slew Rate SR CL = 100 pF - 3.2 - V/s Open Loop Gain AVOL RL = 10 k - 92 - dB Output Short Circuit Current ISC+ Output to VCC/2 -3.5 - mA 2.5 - mA 4.25 - V 0.75 0.9 V 150 pF Gain Bandwidth Product Not meaningful ISCOutput Voltage Range Capacitive Load RL = 10 k to VCC/2, Rf = 20 k, 0 dB VCA gain VO+ Iin(VCA) = +100 A VO- Iin(VCA) = -100 A 4.1 CL Operational Amplifier OA4 Input Offset Voltage VOS - 1.5 5 mV Input Bias Current IB - 200 500 nA Input Offset Current IOS - 10 50 nA VICR+ 4 4.3 - V VICR- - 0.4 0.6 V Input Common Mode Range Equivalent Input Noise Voltage eN(IN) f = 1 kHz - 10.5 14 nV/Hz Equivalent Input Noise Current iN(IN) f = 1 kHz - 0.3 - pA/Hz GBW f = 50 kHz - 7.3 - MHz Slew Rate SR G = +10, CL = 100 pF 2.0 3.2 - V/s Open Loop Gain AVOL RL = 10 k - 92 - dB Gain Bandwidth Product 5. OA1 is stable for closed-loop gains of 2 or greater. 6. Note - OA2 and the VCC/2 buffer require a capacitve load for stability. THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 5 of 16 THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC Electrical Characteristics (con't) 2 Parameter Output Short Circuit Current Output Voltage Range Symbol Conditions Min Typ Max Units ISC+ Output to VCC/2, VID = +0.4 V -1.3 -3.5 -12 mA ISC- Output to VCC/2, VID = -0.4 V 1 2.5 8 mA VO+ RL = 10 k to VCC/2, G = +10 4.1 VOCapacitive Load 4.25 - V 0.75 0.9 V 150 pF 100 - dB CL Power Supply Rejection Ratio PSRR +5V < VCC < +15 V - VCC/2 Reference Buffer Reference Voltage VREF Voltage Divider Impedance Output Short Circuit Current Output Noise Voltage Capacitive Load 6 No Signal, No load on pin 13, VCC = +5 V, RL= 3 k to VCC or GND 2.4 2.5 2.6 V VCC = +15 V - VCC/2 - V - 20 - k RA, RB IOsc- Output to VCC -3 mA IOsc+ Output to GND 4.5 mA eN(OUT) 22 Hz ~ 22 kHz, CFILT= 22 F CL - -120 -117 dBV 22 nF Proportional To Absolute Temperature (PTAT) Voltage Generator Output Voltage VPTAT RL = 10 k, TCHIP = 25 C VCA Gain Change Caused by VPTAT Maximum Sink Current VREF - 0.072 - V VCA Gain at 1 kHz -11 -12 -13 dB Ref TCHIP = 27 C - +0.33 - VPTAT applied to OA2, AV = +1 (VPTAT-VREF)/TCHIP Output Tempco - ISINK(MAX) Capacitive Load %/C 800 A CL 150 pF Performance as a Compander 7 (through an encode-decode cycle) Dynamic Range (Max signal level) - (No Signal Output Noise) 120 dB THD f = 1 kHz 0.1 % -20 dB re: Max Signal 20 Hz ~ 20 kHz 1.5 dB Distortion Frequency response Package Characteristics Parameter Symbol Surface Mount Packages Thermal Resistance Conditions QSOP-28 QFN-24 See page 16 for pinouts and dimensions JA Environmental Regulation Compliance Package soldered to board. Thermal pad not soldered on QFN 8 90 70 C/W Complies with July 21,2011 RoHS 2 Requirements Soldering Reflow Profile Moisture Sensitivity Level Units JEDEC JESD22-A113-D (260 C) Above-referenced JEDEC soldering profile MSL-1 MSL-1 7. Compressor circuit is as shown in Figure 12, Expander circuit is as shown in Figure 13. 8. For best VCA THD performance, QFN thermal pad should not be soldered to the PCB. THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 6 of 16 DUT VCC OA1 AC/DC IN R14 R2 10k0 VPTAT 1k00 8 K3A 9 VREF OUT R1 3k01 27 C12 22p CM NP0 100k C7 10u C8 22u (CFILT) 13 VREF Filt C1 1 14 Gnd Gnd 22n MY 4320 DUTB 4320 25 + 26 OA1 - 28 C5 5 R19 9k09 15 VPTAT VCC 11 K4A R9 5 DUT VCC DUTE C13 150p 8 THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC 5 22p CM NP0 R8 100k R18 K6A K5A 8 4 8 R13 9k09 1k00 5 RL1 10k0 CL1 150p CM NP0 OA1 VEE 100n CM C11 5 C4 100n CM OA1 OUTPUT K1A 8 DUT VCC 8 4 OA4 AC/DC IN K2A 5 C16 22p VCA AC IN C19 R44 20k0 470n MY OA2 OUT CL2 22n MY R57 6k82 DUTA 4320 5 C20 78n My C22 K12A 22p CM R17 9k09 8 4 R12 1k00 5 + - 2 5 OA3 OUT CL6 150p RL6 20k0 K17A 4 8 5 8 K15A 5 K16A 4 8 8 R4 R10 4 1k00 8 K18A VREF OUT C17 4 8 10n MY 5 R7 100k R56 976 10k0 8 5k 0.1% R50 15k0 5k 0.1% R22 VPTAT 5 2 3 6 OA2 AC/DC IN 8 R25 R43 3k32 150p CM NP0 22p CM NP0 R5 100k R15 K10A K9A 5 CL4 C9 8 RL4 10k0 9k09 RMS OUT 10k0 K14A C15 22p CM NP0 5 8 OA4 OUTPUT 18 R16 9k09 K11A DUT-2D 4320 + 16 17 OA4 - 5 100k 8 R3 OA2 3 K8A R6 5 22p CM NP0 21 23 In VCA 20 OA3 Ec+ + VREF 4 8 1k00 5 K7A R47 15k0 R46 20k0 C10 R11 DUTC 4320 8 RMS Out C14 150p CM NP0 In R24 6 CT 7 5k 0.1% C27 RMS AC IN 22u C18 1u (CTIME) 1 U1A OP-07B VREF OUT K13A Figure 2. 4320 Test Circuit Schematic (QSOP-28 pin assignments shown) REPRESENTATIVE DATA Figure 3. VCA THD vs. Level at 0 dB gain (BW=22kHz) Figure 4. VCA THD vs. Level at +12 dB gain (BW=22kHz) Figure 5. VCA THD vs. Level at -12 dB gain (BW=22kHz) Figure 6. VCA THD vs. Frequency (BW=80kHz) THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 7 of 16 THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC Figure 7. VCA Gain vs. Control Voltage Figure 8. VCA Noise vs. Gain (BW=22kHz) Figure 9. VCA Offset vs. Gain Figure 10. RMS Output vs. Level Figure 11. RMS Frequency Response vs. Level Theory of Operation The THAT 4320 Dynamics Processor combines THAT Corporation's proven Voltage-Controlled Amplifier (VCA) and RMS-Level Detector designs with four general-purpose opamps to produce an Analog Engine useful in a variety of dynamics processor applications. The part is integrated using a proprietary, fully complementary, dielectric-isolation process. This process produces very high-quality bipolar transistors (both NPNs and PNPs) with unusually low collector-substrate capacitances. The 4320 takes advantage of these devices to deliver wide bandwidth and excellent audio performance while consuming very low current and operating over a wide range of power supply voltages. For details of the theory of operation of the VCA and RMS Detector building blocks, the interested reader is referred to THAT Corporation's data sheets on the 2180-Series VCAs and the 2252 RMS Level Detector. Theory of the interconnection of exponentially-controlled VCAs and log-responding level detectors is covered in THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 8 of 16 THAT Corporation's design note DN01, The Mathematics of Log-Based Dynamic Processors. The VCA -- in Brief The VCA in THAT 4320 is based on THAT Corporation's highly successful complementary log-antilog gain cell topology -- The Blackmer(R) VCA -- as used in THAT 2180-Series IC VCAs. VCA symmetry is trimmed during wafer probe for minimum distortion. No external adjustment is allowed. See Figures 3 ~ 6, page 6 for the representative THD data. Input signals are currents in the VCA's IN pin. This pin is a virtual ground with dc level approximately equal to VREF, so in normal operation an input voltage is converted to input current via an appropriately sized resistor (R44 in Figure 2, Page 6). Because the currents associated with dc offsets present at the input pin and any dc offset in preceding stages will be modulated by gain changes (thereby becoming audible as thumps), the input pin is normally ac-coupled (C19 in Figure 2). The VCA output signal is also a current, inverted with respect to the input current. In normal operation, the output current is converted to a voltage via inverter OA3, where the ratio of the conversion is determined by the feedback resistor (R46 or R47, Figure 2) connected between OA3`s output and its inverting input. The signal path through the VCA and OA3 is noninverting. The gain of the VCA is controlled by the voltage applied between EC+ and EC-. Note that EC- is an internal node connected to the VREF generator. Gain (in decibels) is proportional to (EC+ - EC-). See Figure 7 [page 7]. The constant of proportionality is 6.0 mV/dB for the voltage at EC+ (relative to VREF). The VCA's noise performance varies with gain in a predictable way, but due to the way internal bias currents vary with gain, noise at the output is not strictly the product of a static input noise times the voltage gain commanded. Figure 8 [page 7] plots noise (in dBV -- referenced to 1 V -- in a 22 kHz bandwidth) at the output of OA3 vs. VCA gain commands over a range of -100 dB to +30 dB gain. At large attenuation, the noise floor of ~-109 dBV is limited by the input noise of OA3 and its feedback resistor. At 0 dB gain, the noise floor is ~-98 dBV as specified. In the vicinity of 0 dB gain, the noise increases more slowly than the gain: approximately 5 dB noise increase for every 10 dB gain increase. Finally, as gain approaches 30 dB, output noise begins to increase directly with gain. While the 4320's VCA circuitry is very similar to that of the THAT 2180 Series VCAs, there are several important differences, as follows. THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC 1) Supply current for the VCA depends on VCC. At +5 V VCC, approximately 500 A is available for the sum of input and output signal currents. This increases to about 1 mA at +15 V VCC. (Compare this to ~1.8 mA for a 2180 Series VCA when biased as recommended. This is appropriate given the lower supply voltage for the 4320.) 2) The signal current output of the VCA is internally connected to the inverting input of on-chip opamp OA3. In order to provide external feedback around this opamp, this node is brought out to a pin. 3) Only the EC+ node is available for gain control. A SYM control port (similar to that on the 2180 VCA) exists, but is driven from an internally trimmed current generator. The negative control port (EC-) is internally connected to VREF. 4) The control-voltage constant is approximately 6.0 mV/dB, due primarily to the lower internal operating temperature of the 4320 compared to that of the 2180 Series (and the 4301). 5) The OTA used for the VCA's internal opamp in the 4320 uses less emitter degeneration resistance in its output than that of the 2180 VCA. This requires that the source impedance at the VCA's input (which is a summing junction) must be under 5 k at frequencies over 1 MHz. In Figure 2, C16 and R57 accomplish this. See the applications section for an alternative on how to address this issue. The RMS Detector -- in Brief The 4320's detector computes RMS level by rectifying input current signals, converting the rectified current to a logarithmic voltage, and applying that voltage to a log-domain filter. The output signal is a dc voltage proportional to the decibel-level of the RMS value of the input signal current. Some ac component (at twice the input frequency) remains superimposed on the dc output. The ac signal is attenuated by a log-domain filter, which constitutes a single-pole rolloff with cutoff determined by an external capacitor and a programmable dc current. As in the VCA, input signals are currents to the RMS IN pin. This input is a virtual ground with dc level equal to VREF, so a resistor (R24 in Figure 2) is normally used to convert input voltages to the desired current. The level detector is capable of accurately resolving signals well below 10 mV (with a 5 k input resistor). However, if the detector is to accurately track such low-level signals, ac coupling is normally required (C27 in Figure 2). Note also that small, low-voltage electrolytic capacitors used for this purpose may create significant leakage if they support half the supply voltage, as is the case when the source is dc-referenced to ground. To THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 9 of 16 ensure good detector tracking to low levels, a tantalum capacitor or high-voltage electrolytic may be required for input coupling. The log-domain filter cutoff frequency is usually placed well below the frequency range of interest. For an audio-band detector, a typical value would be 5 Hz, or a 32 ms time constant (). The filter's time constant is determined by an external capacitor CTIME attached to the CT pin, and an internal current source (IT) connected to CT. The current source is internally fixed at 7.5 A. The resulting time constant in seconds is approximately equal to 3467 * CTIME. Note that, as a result of the mathematics of RMS detection, the attack and release time constants are fixed in their relationship to each other. The RMS detector is capable of driving large spikes of current into CTIME, particularly when the audio signal input to the RMS detector increases suddenly. This current is drawn from VCC at pin 15 (QFN pin 16), fed through CTIME at pin 7 (QFN pin 10), and returns to the power supply through the ground end of CTIME. If not handled properly through layout and bypassing, these currents can mix with the audio with unpredictable and undesirable results. As noted in the Applications section, local bypassing from the VCC pin to the ground end of CTIME is strongly recommended in order to keep these currents out of the ground structure of the device. The dc output of the detector is scaled with the same constant of proportionality as the VCA gain control: 6.0 mV/dB. See figure 10 [page 7]. The detector's 0 dB reference (iin0), the input current which causes the detector's output to equal VREF), is trimmed during wafer probe to approximately equal 7.5 A. The RMS detector output stage is capable of sinking or sourcing 125 A. It is also capable of driving up to 150 pF of capacitance. Frequency response of the detector extends across the audio band for a wide range of input signal levels. Note, however, that it does fall off at high frequencies at low signal levels. See figure 11 (page 7). Differences between the 4320's RMS Level Detector circuitry and that of the THAT 2252 RMS Detector include the following. 1) The rectifier in the 4320 RMS Detector is internally balanced by design, and cannot be balanced via an external control. The 4320 will typically balance positive and negative halves of the input signal within 10 %, but in extreme cases the mismatch may reach +40, -30 % (3 dB). However, even such extreme-sounding mismatches will not significantly increase rippleinduced distortion in dynamics processors over that caused by signal ripple alone. THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC 2) The time constant of the 4320's RMS detector is determined by the combination of an external capacitor (connected to the CT pin) and an internal current source. The internal current source is set to about 7.5 A. A resistor is not normally connected directly to the CT pin on the 4320. 3) The 0 dB reference point, or level match, is also set to approximately 7.5 A. However, as in the 2252, the level match will be affected by any additional currents drawn from the CT pin. The Opamps -- in Brief The four opamps in the 4320 have been optimized independently to suit each one's intended application. While they all use PNP input stages, they differ in bandwidth, noise level, and compensation scheme depending on their expected uses. Therefore, to get the most out of the 4320, it is useful to know the major differences among these opamps. OA1 - Low Source Impedance Pre-amp OA1, with typical equivalent input noise of 4.5 nV/Hz, is the quietest opamp on the 4320. This opamp is intended for signal conditioning such as preamplification from low-impedance sources. (At source impedances of >5.6 k, the input current noise contribution will surpass the voltage contribution.) OA1 is stable for closed-loop gains of 2 or greater. Its output typically swings to within 0.75 V of VCC or VEE, allowing it to support a 1.2 VRMS sine wave from a single +5 V supply (4.75 VRMS with a +15 V supply). Its typical slew rate is ~ 4 V/s, allowing the part to support maximum level sine waves at up to 360 kHz on a +5 V supply (94 kHz on a +15 V supply). OA1`s output is capable of driving up to 150 pF, so it is possible to directly bypass RF to ground via a small capacitor at OA1`s output, as is often desired in wireless transmitter applications. OA1`s most unusual feature9 is that it's negative power supply connection is brought out separately to VEE at pin 28 (QFN pin 3) to provide additional headroom in certain applications. While VEE is normally connected to the power supply ground (and pins 1 and 14 (QFN pin 4 and 15), which are the ground connections for the rest of the chip), it can be connected to a separate negative supply. OA1`s positive supply connection is internally connected to VCC at pin 15 (QFN pin 16). Therefore, OA1 sees as its supply voltage the difference between VCC and VEE. Note that this difference must not exceed 16 V. To gain an advantage from the separate VEE connection for this opamp, the design must provide a negative 9. THAT has applied for patent coverage on this novel approach. THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 10 of 16 supply below ground to this pin. By doing, so, OA1 can gain additional voltage swing over that available to the rest of the IC. Because OA1 is commonly used as a pre-amp before a noise reduction compressor based on the rest of the chip, headroom is most critical at this point. (The VCA will reduce the audio signal's dynamic range to a more manageable level for subsequent stages.) The rest of the chip can run from +5 V and ground to maintain low power dissipation, while only OA1 is run from, say, a 5 V supply to gain additional headroom. To see how this works in practice, suppose VCC is +5 V. If VEE is set to 0 V (ground), the maximum swing at OA1`s output is typically 3.5 V (typically, OA1 reaches within ~0.75 V of its supply rails), If, instead, VEE is set to -5 V, the maximum swing at OA1`s output increases to 8.5 V -- for a 7.7 dB increase in dynamic range! OA2 - Control Voltage Buffer OA2 is intended as a control voltage buffer, and is the least general purpose of the four opamps. It is externally compensated, and requires at least 22 nF at its output to remain stable. This was a deliberate design choice based on several factors including the relatively limited bandwidth and voltage swing required for the VCA control port and the importance of low noise (and low RF content) at this node. Additionally, the capacitive high-frequency output impedance guarantees stability in the VCA. Because it is intended to handle only the VCA control port signal (consisting primarily of dc with added low frequency content), OA2 is optimized for dc at the expense of ac performance. This opamp has limited input compliance (1 V common mode range), is relatively slow (120 kHz gain-bandwidth product with a typical 100 nF capacitive load), has low open-loop gain (57 dB with the typical 10 k resistive load), and has approximately a 10 output impedance. These characteristics, while limiting in an opamp intended for handling audio signals, are ideal for the control voltage buffer. In particular, compensating the opamp at its output takes advantage of an often-required RF-bypass capacitor to minimize noise pickup at the sensitive VCA control port. OA3 - VCA Current-to-Voltage Converter OA3 is intended to translate the VCA's output currents into voltage signals. It is a unity-gain stable, 7.3 MHz opamp with moderately low input noise of 10.5 nv/Hz. This noise floor complements that of the VCA. Like OA1, because it handles audio signals directly, OA3 is optimized for audio performance. It's output typically swings to within 0.75 V of VCC or ground, THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC allowing it to support a 1.2 VRMS sine wave from a single +5 V supply (4.75 VRMS with a +15 V supply). It's typical slew rate is ~3.2 V/s, allowing the part to support maximum level sine waves at up to 290 kHz on a +5 V supply (75 kHz on a +15 V supply). As with the other opamps, OA3`s output is capable of driving up to 150 pF, so it is possible to directly bypass RF to ground via a small capacitor at OA3`s output. It's output section is capable of supplying at least 1 mA, making it possible to use this opamp directly as the output stage in lightly loaded applications. Note, however, that OA3`s output is not designed to withstand an indefinite short-circuit to a power supply or ground rail, and a resistor should be included in series with such outputs to ensure stability with capacitive loads larger than 150 pF. OA4 - General Purpose OpAmp OA4 is intended for either signal or control voltage applications. It is a unity-gain stable, 7.3 MHz opamp with moderately low input noise voltage of 10.5 nV/Hz, and moderately low input noise current of 0.3 pA/Hz. Because of it's lower current noise, OA4 is a better choice for an audio pre-amp than OA1 in cases where the source impedance feeding it is high. All other characteristics of OA4 are similar to those of OA3. VCC/2 Reference Buffer For single-supply applications, the 4320 requires a center-tap to provide a synthetic "ground" reference for its circuitry. The 4320 contains a built-in resistive divider (at pins 13/14/15), followed by a buffer, to provide a low-impedance source at approximately half VCC. Note that the center tap of the resistive divider is brought out to filter the voltage, thereby minimizing noise in the divider. A large electrolytic capacitor (typically 22 F or greater) is used for this purpose. The output of the buffer is available at pin 11. This is "VREF". The buffer is capable of delivering ~3 mA at its output. Like OA2, it is compensated by capacitance at its output, working against an internal output impedance of approximately 10 ; at least 22 nF should be used to ensure stability, reduce high-frequency output impedance, and attenuate high-frequency noise. VREF may be used to supply a "ground" reference voltage to other sections of circuits beyond the 4320 itself. However, in any such uses, the designer should take care to minimize currents, especially signal currents, that flow through the VREF line. Any signal currents should return to the real circuit ground (GND); VREF should be connected only to relatively high impedance loads (e.g., the positive input of opamps). Where THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 11 of 16 significant currents (signal or otherwise) must be delivered at the VREF dc level, an opamp should be used to buffer the VREF line itself. Another approach to power supply arrangements is to operate the 4320 from symmetrical split supplies (e.g., 5 V and ground). In such cases, the center-tap of the resistive divider at pin 13 (QFN pin 14) should be grounded. This will force VREF to very nearly ground (within the offset of the VCC/2 buffer). A final note on the subject of power supply connections is that both of the 4320's two GND, pins 1 and 14 (QFN pins 4 and 15), must be tied together for proper operation of the device. While these pins are tied together internally on the chip, due to the large size of the die inside the part, the resistance and inductance of the internal connection is not as low as an external PCB trace can provide. The 4320 may not meet all its specifications unless a short PCB connection is made between these two pins. PTAT Voltage Generator The VCA control port and the RMS-level detector output both share a fundamental temperature drift proportional to absolute temperature. Room temperature is approximately 300 K (or 27 C), so near room temperature the drift amounts to +0.33 %/C. The drift is expressed in percent per degree Celsius because the magnitude of the change with temperature depends on the gain control command or detected level being presented. There is no temperature drift at 0 dB gain, or at the RMS' reference level. But, away from either of these 0 dB points, the scale factor of these parameters varies by 0.33 % for each degree Celsius of temperature change. The PTAT voltage generator produces an output that varies directly with absolute temperature. At 25 C, it's output is 72 mV. One end of the generator is connected to VREF, the other (negative end) is buffered and brought out at VPTAT at pin 9 (QFN pin 12). While one application for the voltage on this pin might be to read the temperature of the IC, it has many important practical uses in audio applications based on the 4320. Basically, it provides a voltage that can be used, after appropriate scaling, to supply any gain controls or offsets used to condition the RMS detector output and/or the VCA gain control signals. An example may help make this clear. Suppose a designer wants to provide a potentiometer to control signal gain through the VCA. If the desired gain range is 0 to +20 dB, the VCA control port must be driven from 0 mV (for 0 dB gain) to +120 mV (for +20 dB gain), but only at room temperature. (At room temperature, the gain control constant is 6.0 mV/dB.) If the temperature increases by 10 C, the voltage for 0 dB gain remains the THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC same, but that for 20 dB gain increases by 3.3 %, to 124 mV. If the same 120 mV gain command is applied (because it comes from a source that does not vary with temperature), the gain will be 19.35 dB, not 20 dB. If the supply that feeds the gain-control pot derives from a stable voltage source, the commanded gain will drift with temperature. Alternatively, if the supply can be made to vary with temperature just as the control port's sensitivity drifts, the two can compensate each other and the result will be stable. That is the purpose of the 4320's PTAT voltage generator: to supply a voltage that drifts exactly as the VCA and the RMS detector drifts. The PTAT voltage can be used, with appropriate scaling, to reference all gain controls, gain offsets, and threshold setting amplifiers throughout the levelprocessing side chain. And, because the PTAT generator is integrated on the same IC as its VCA and RMS detector, temperature tracking between these three components is excellent. The No Connection Pins Some pins on the THAT4320 are labeled "No Connection" (N/C). These pins are not internally connected to the 4320 die, so it is acceptable to leave these pins unconnected or to connect these pins to some external circuit nodes. In fact, the placement of the N/C pins was chosen partly to facilitate passive guarding to certain pins which are sensitive to low-level leakage currents (e.g., the RMS and VCA inputs). Because the dc potential at the most sensitive circuit nodes is very close to VREF, THAT Corporation recommends that all the N/C pins be connected to VREF wherever possible. However, layout constraints may preclude such a connection. In this case, either leave the pins open, or choose a slow moving (dc) signal that is close in dc potential to VREF, such as VPTAT. Tying the N/C pins to VCC or GND -- not recommended -- will guard against AC signals, but runs the risk of generating unanticipated dc leakage currents which can spoil the performance of the 4320's VCA and RMS detector. Noise Reduction (Compander) Configurations A primary use of the 4320 is for noise reduction systems, particularly within battery-operated devices. In these applications, one 4320 is configured for use as a compressor to condition audio signals before feeding them into a noisy channel. A second 4320, configured as an expander, is located at the receiver end of the noisy channel. The compressor increases gain in the presence of low-level audio signals, and reduces its gain in the presence of high-level audio signals. The expander works in opposite, complementary fashion to restore the original signal levels present at the input of the compressor. THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 12 of 16 During low-level audio passages, the compressor increases signal levels, bringing them up above the noise floor of the noisy channel. At the receiving end, the expander reduces the signal back to it's original level, in the process attenuating the channel noise. THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC range of levels; the VCA responds accurately to a wide range of gain commands; the detector output and the VCA control input are fully configurable; and the part contains enough opamps to provide many options in signal conditioning. All these features mean that the 4320 will support a wide range of compander designs (and more), including simple 2:1 wide range (levelindependent) systems, level-dependent systems with thresholds and varying compression slopes, systems including noise gating and/or limiting, and systems with varying degrees of pre-emphasis and filtering in both the signal and detector paths. Furthermore, much of this can be accomplished by extensively conditioning the control voltage sidechain rather than the audio signal itself. The audio signal can pass through as little as one VCA and one opamp, and still support multiple ratios, thresholds, and time constants. During high-level audio passages, the compressor decreases signal levels, reducing them to fit within the headroom limits of the noisy channel. The expander increases the signal back to its original level. While the channel noise may be increased in this action, a welldesigned compander will mask the noise floor with the signal itself. The 4320 was designed to facilitate the design of a wide variety of companding noise reduction systems. The RMS detector responds accurately over a wide Optional Clipper/Overload Protection D1 D2 20 kHz Butterworth LPF C5 6dB Static Gain C9 In 10n C3 R3 R1 10k0 10u C8 22p NPO R4 1k10 23 In VCA 3n3 NPO 2.5% 20k0 21 OA3 R5 20 Ec+ 2k05 9k09 2 R13 OA2 3 2k49 U1A 4320 VCC = +5 V + - VREF R8 RMS 8 Out 100n 2k26 U1C 4320 4k99 C11 C12 100n - 16 18 OA4 + C1 1n NPO 2.5% 4320 10 dB / ~ 50 s pre-emphasis C14 R10 R9 Un-used and available for low noise pre-amp or other circuits U1B 4320 VREF 27 + 25 26 OA1 - 28 U1D 17 R6 VREF 4 Decoder Out 4k99 R7 10k0 6dB Static Gain VPTAT CT 7 C10 10u (CTIME) 6 In R2 4k99 VCC 10n C2 470n 60 Hz HPF U1E 4320 15 C4 FILT 22u C13 100n 13 14 VCC VPTAT Filt VREF Gnd Gnd 9 VPTAT 11 VREF 1 C7 22u Figure 12. THAT4320 2:1 Encoder Circuit (QSOP-28 pin assignments shown) THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation C6 100n Document 600045 Rev 08 Page 13 of 16 THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC Applications The 4320 includes so many useful building blocks and operates from such a wide range of supply voltages that it is suitable for a wide variety of dynamics processing applications. Chief among these are wireless companding systems. For this datasheet, we show the part in a simple 2:1 companding noise reduction system that performs as well or better than any analog companding solution on the market today. Many other configurations of the 4320 are possible, but are not shown here. THAT intends to publish additional circuits in forthcoming applications notes. Please check with THAT's applications engineering department to see if your application has been covered yet, and for personalized assistance with specific designs. The encoder Figure 12 shows a simple 2:1 encoder or feedback compressor. The encoder in a wireless companding system is located in the transmitter and generally operates from a battery supply. To optimize signal levels within the voltage limitations of the battery supply, the encoder VCA gain is offset by 6 dB via the ratio of R4 to R1. Additionally, another 6 dB of static gain is injected at the control port opamp, via VPTAT and R7. (A 36 mV dc offset is required to produce 6 dB of static gain. Since VPTAT ~ -72 mV, a gain of -1/2 will create the required 36 mV. Because the PTAT generator voltage tracks in temperature with the VCA gain control constant, this gain will be stable over temperature.) This encoder includes a high-frequency pre-emphasis network at the input of the VCA (R3/C9) that ultimately provides 20 dB of gain at 20 kHz. Its lower corner frequency is at approximately 1.5 kHz (f1); the upper corner is near 15 kHz (f2). Companding noise reduction encoders often include a clipper somewhere in the signal path to prevent overmodulation of the RF channel. The optional antiparallelled diodes D1 and D2, can perform that function in this circuit, and should be placed ahead of the 20 kHz Butterworth low-pass filter composed of OA4 and its surrounding components. This placement helps reduce "spectral splatter" that results from momentary clipping. What clipping takes place is limited in duration to transients only, since the encoder will eventually reduce its gain to below the clip point. The output of the low-pass filter is the output of the encoder. This is where the input to the RMS detector is derived. The input circuit for the RMS detector includes another pre-emphasis network which provides a maximum of 10 dB of pre-emphasis (R10/C14), rising at approximately 2.9 kHz (f3), and stopping at around 6.5 kHz (f4). These frequencies were chosen such that f1 % f2 = f3 % f4 This effectively centers the rising sections of both the RMS and VCA pre-emphasis curves. This network feeds the input of the RMS detector, which is a virtual ground referenced to VREF. As described in the Theory of Operation section "The RMS Detector - In Brief" (on page 9), the RMS detector is capable of driving large spikes of current into the averaging capacitor CTIME. To prevent these currents from upsetting circuit grounds, it is necessary to bypass VCC to a point very near the grounded end of the CTIME with a capacitor (C4 in Figure 12) equal to or greater than the value of CTIME. The grounded ends of these two capacitors should be connected together before being tied to the rest of the ground system. Doing so will ensure that the current spikes flow within the local loop consisting of the two capacitors, and stay out of the ground system. This requirement applies to the decoder and other applications of the THAT4320 as well. The output of the RMS detector is zero volts when the RMS input current is equal to the timing current (internally set to ~7.5 A). A low-frequency voltage level of -26 dBu was chosen as the desired zero dB reference since this, in conjunction with the applied static gain, makes optimal use of the available gain in the VCA. Then, the RMS detector's low-frequency input resistance can be calculated as: R2 = -26 0.775 % 10 20 7.5 A { 4.99 k From the desired 10 dB pre-emphasis, the value for R10 can then be calculated to be 2.26 k. C14 is calculated based on the desired pre-emphasis starting frequency. In THAT Corporation's design note DN03, A Signal Limiter for Power Amplifiers, the compression ratio for a feedback compressor was derived using the analytical technique described in DN01A, The Mathematics of Log Based Dynamics Processors. This technique is referred to as `working in the log domain'. Using these methods, it can be shown that the compression ratio (C.R.) of a feedback compressor is C.R. = 1 + A, where A is the absolute value of the gain of the side chain. The RMS detector's output is connected to the VCA gain control port (EC+) through OA2, configured for an THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 14 of 16 inverting gain of one. This fixes the compression ratio at 2:1. Note that the negative sign in the side chain gain makes this circuit a compressor. THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC and C9 provide the necessary compensation to maintain stability. OA1 is used to implement another 20 kHz Butterworth low-pass filter. This ensures that noise picked up in the transmission channel will not cause mistracking between the detectors in the encoder and decoder. The output of this filter feeds the RMS detector input, which in turn has the same pre-emphasis network as in the encoder RMS detector. The decoder Figure 13 shows the THAT4320 configured as a 2:1 expander, an arrangement intended to complement the encoder in Figure 12. This circuit is optimized for low-voltage operation, as might be the case for a decoder in an in-ear monitoring system which will run from battery power. Using the same log based mathematics described earlier, the expansion ratio of a feedforward expander can be shown to be The pre-emphasis network from the VCA input in Figure 12 is now in the feedback loop of OA3; This provides de-emphasis. The VCA is set up with -12 dB of static gain to keep output signal levels low for battery operation. Because the VCA is not stable unless it sees a high frequency source impedance of 5 k or less, R5 E.R. = 1 + A OA2 is configured as a gain-of-one follower. This reverses the polarity of the control signal relative to the encoder, and makes this circuit a 2:1 expander. 20 dB / ~ 100 s de-emphasis C5 22p NPO C11 C9 R5 10n R8 4k99 47p 10k0 C3 R7 2u2 40k2 -12dB Static Gain 23 In Encoder Out C6 R1 9k09 R3 100k 1u VREF R6 27 2k05 C8 1n U1A 4320 + 15 13 C12 22u 14 C13 100n VCC VPTAT Filt VREF Gnd Gnd 9 + 2 C14 100n 25 C14 R8 10n C2 2k26 470n 4k99 VPTAT VREF 11 1 + VREF 4 R2 60 Hz HPF U1C 4320 6 In VCC Out 20 - - U1E 4320 - OA3 Ec+ OA2 3 U1B 4320 26 OA1 28 21 VCA 20 kHz Butterworth LPF C1 3n3 NPO R4 1k10 RMS 8 Out CT 7 Un-used C10 10u 17 C7 100n C4 22u 18 U1D 16 OA4 + 4320 VREF Figure 13. THAT4320 2:1 Decoder Circuit (QSOP-28 pin assignments shown) THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Document 600045 Rev 08 Page 15 of 16 General Dynamics Processor Configurations The same distinguishing features that make the 4320 so applicable to companding noise reduction systems also qualify it for application to dynamics processors of all types. This is even more so when the application must run from battery power. The 4320 is versatile enough to be used as the heart of a compressor, expander, noise gate, AGC, de-esser, frequency-sensitive compressor, and many other dynamics processors. It is beyond the scope of this data sheet to provide specific advice about any of these functional classes. We refer the interested reader to THAT's applications notebooks volumes 1 and 2, which contain many circuits based on THAT's other VCAs and RMS level detectors, but are largely applicable to the THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC 4320 with only minor variations. Of course, look for more applications information aimed specifically at the 4320 in the future. Where to go from here The design of compander systems and dynamics processors is a very intricate art: witness the proliferation of first analog, then digital companding systems, and the many different dynamics processors available in the market today. In the applications section of this data sheet, we offer a single example of a compander as a starting point only. THAT Corporation's applications engineering department is ready to assist customers with suggestions for tailoring and extending these basic circuits to meet specific needs. Ordering Information Package 28 pin QSOP Order Number 4320Q28-U 24 pin QFN (5x5) 4320N24-U Table 1. Ordering information For sales: Tel: +1 (508) 634-9922 Fax: +1 (508)634-6698 E-mail: sales@thatcorp.com Revision History Revision ECO Date Changes 05 2258 04/06/09 Fixed errors in specification table and Figure 4. -- 06 2377 02/23/10 Added QFN-24 package. Corrected equation. -- 07 2856 03/18/14 Clarified the minimum gain for stability of OA1 08 2933 07/22/15 Corrected Package Characteristics table and Figure 11. THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Page 3, 4, 9 5, 7 Document 600045 Rev 08 Page 16 of 16 THAT4320 Pre-trimmed Low-voltage Low-power Analog Engine(R) Dynamics Processor IC Package Information Pin Name GND OA2 +IN OA2 -IN OA2 OUT No Connection RMS IN CAP RMS OUT VPTAT No Connection VREF No Connection FILTER GND VCC OA4 OUT OA4 -IN OA4 +IN No Connection OA3 OUT VCA OUT No Connection VCA IN No Connection OA1 OUT OA1 -IN OA1 +IN OA1 VEE Pin Number 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 1 D A E B C Pin Number 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24 THERMAL PAD (25) Table 3. QFN-24 (5x5) pin assignments * For best VCA THD performance the QFN's thermal pad should not be soldered to the PCB. H 0-8 I ITEM A B C D E G H I J MILLIMETERS 9.80 - 9.98 3.81 - 3.99 5.79 - 6.20 0.20 - 0.30 0.635 BSC 1.35 - 1.75 0.10 - 0.25 0.40 - 1.27 0.19 - 0.25 INCHES 0.386 - 0.393 0.150 - 0.157 0.228 - 0.244 0.008 - 0.012 0.025 BSC 0.0532 - 0.0688 0.004 - 0.010 0.016 - 0.050 0.0075 - 0.0098 Figure 14. QSOP-28 package drawing Table 2. QSOP-28 pin assignments Pin Name OA1 -IN OA1 +IN OA1 VEE GND OA2 +IN OA2 -IN OA2 OUT No Connection RMS IN CAP RMS OUT VPTAT VREF FILTER GND VCC OA4 OUT OA4 -IN OA4 +IN No Connection OA3 OUT VCA OUT VCA IN OA1 OUT GND* G J Exposed Thermal Pad I 13 18 12 D 19 J B K 24 7 6 1 E F G BOTTOM VIEW C H 0 A ITEM A B C D E F G H I J K MILLIMETERS 5.00 0.10 5.00 0.10 0.90 0.05 0.25 0.05 0.65 0.05 0.40 0.05 0.00 ~ 0.05 0.20 0.05 3.40 0.05 3.40 0.05 C' 0.4 x 45 INCHES 0.197 0.004 0.197 0.004 0.035 0.002 0.010 0.002 0.026 0.002 0.016 0.002 0.000 ~ 0.020 0.008 0.002 0.134 0.002 0.134 0.002 C` 0.016 x 45 Figure 15. QFN-24 (5x5) package drawing THAT Corporation; 45 Sumner Street; Milford, MA 01757-1656; USA Tel: +1 508 478 9200; Fax: +1 508 478 0990; Email: info@thatcorp.com; Web: www.thatcorp.com Copyright (c) 2015, THAT Corporation Mouser Electronics Authorized Distributor Click to View Pricing, Inventory, Delivery & Lifecycle Information: THAT Corporation: 4320Q28-U 4320N24-U 4320Q28-UR 4320N24-UR