FN8386 Rev 8.00 Page 1 of 32
February 4, 2016
FN8386
Rev 8.00
February 4, 2016
ISL28022
Precision Digital Power Monitor
DATASHEET
The ISL28022 is a bidirectional high-side and low-side digital
current sense and voltage monitor with serial interface. The
device monitors current and voltage and provides the results
digitally along with calculated power. The ISL28022 provides
tight accuracy of less than 0.3% for both voltage and current
monitoring over the entire input range. The digital power
monitor has configurable fault thresholds and measurable
ADC gain ranges.
The ISL28022 handles common-mode input voltage ranging
from 0V to 60V. The wide range permits the device to handle
telecom, automotive and industrial applications with minimal
external circuitry. Both high- and low-side ground sensing
applications are easily handled with the flexible architecture.
The ISL28022 consumes an average current of just 700µA and is
available in a 10 Ld MSOP package. The ISL28022 is also offered
in a space saving 16 Ld QFN package. The part operates across
the extended temperature range from -40°C to +125°C.
Related Literature
•AN1955, “Design Ideas for Intersil Digital Power Monitors”
•AN1875, “ISL28022 Digital Power Monitor Evaluation Kit
(ISL28022EVKIT1Z)
•AN1811, “ISL28022 Digital Power Monitor 8 Site Evaluation
Kit”
Features
• Bus voltage sense range . . . . . . . . . . . . . . . . . . . . . . 0V to 60V
•16-bit ∑∆ADC monitors current and voltage
• Voltage measuring error . . . . . . . . . . . . . . . . . . . . . . . . . <0.3%
• Current measuring error . . . . . . . . . . . . . . . . . . . . . . . . . <0.3%
• Handles negative system voltage
• Overvoltage/undervoltage and current fault monitoring
•I
2C/SMBus interface
•Wide V
CC range . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3V to 5.5V
• ESD (HBM). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8kV
• Supports high speed I2C . . . . . . . . . . . . . . . . . . . . . . . 3.4MHz
Applications
•Routers and servers
• DC/DC, AC/DC converters
• Battery management/charging
• Automotive power
•Power distribution
• Medical and test equipment
FIGURE 1. TYPICAL APPLICATION
A1
SMBCLK/SCL
SMBDAT/SDA
VINP
VINM
GND
RSH ADC
16-BIT
SW
MUX
TO µC
VOLTAGE
REGULATOR
VOUT
EN
ECLK/INT
VIN = 0V TO 60V
REG
MAP
VCC
LOAD
VBUS
A0
VCC
I2C
SMBUS
ISL28022
FN8386 Rev 8.00 Page 2 of 32
February 4, 2016
Table of Contents
Block Diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Ordering Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Pin Configurations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Pin Descriptions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Absolute Maximum Ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Thermal Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Recommended Operating Conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Electrical Specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Typical Performance Curves . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Functional Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Detailed Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Functional Pin Descriptions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Register Descriptions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Serial Interface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Protocol Conventions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
SMBus Support . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Device Addressing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Write Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Read Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Broadcast Addressing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
I2C Clock Speed . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Signal Integrity. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Measurement Stability vs Acquisition Time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Fast Transients. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
External Clock. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Over-Ranging . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Shunt Resistor Selection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Lossless Current Sensing (DCR). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
A Trace as a Sense Resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Revision History. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
About Intersil . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Package Outline Drawing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
M10.118 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
L16.3x3B . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
ISL28022
FN8386 Rev 8.00 Page 3 of 32
February 4, 2016
Block Diagram
FIGURE 2. BLOCK DIAGRAM
DIGITAL
CONTROL
LOGIC
I2C
SM BUS
A0
16
A1
SMBCLK
SMBDAT
OSC
VINM
VINP
VCC
GND
ADC
16-BIT REG
MAP
REF
CLOCK
DIV
SW
MUX
CM = 0 TO 60V
ECLK/INT
VBUS
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
PART
MARKING
TEMP RANGE
(°C)
PACKAGE
(RoHS Compliant)
PKG.
DWG. #
ISL28022FUZ 8022F -40 to +125 10 Ld MSOP M10.118
ISL28022FRZ 022F -40 to +125 16 Ld QFN L16.3x3B
ISL28022EVKIT1Z ISL28022 Evaluation Kit (Includes Dongle Board, Generic Evaluation Board, RLOAD Board)
ISL28022MBEV1Z ISL28022 Generic Evaluation Board
ISL28022EV1Z ISL28022 8-site Evaluation Board
NOTES:
1. Add “-T” suffix for QFN 6k or MSOP 2.5k units tape and reel options. Add “-T7A” suffix for 250 units tape and reel options. Please refer to TB347 for
details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL28022. For more information on MSL please see tech brief TB363.
ISL28022
FN8386 Rev 8.00 Page 4 of 32
February 4, 2016
Pin Configurations
ISL28022
(10 LD MSOP)
TOP VIEW
ISL28022
(16 LD QFN)
TOP VIEW
A1
A0
EXT_CLK/INT
SDA/SMBDAT
SCL/SMBCLK
1
2
3
4
5
10
9
8
7
6VCC
GND
VBUS
VINM
VINP
1
3
4
15
A1
A0
EXT_CLK/INT
SDA/SMBDAT
NC
NC
NC
VINP
16 14 13
2
12
10
9
11
6578
VINM
VBUS
GND
VCC
SCL/SMBCLK
NC
NC
NC
GND
Pin Descriptions
MSOP
PIN
NUMBER
QFN PIN
NUMBER
PIN
NAME DESCRIPTION
11 A1I
2C address, Bit 1
22 A0I
2C address, Bit 0
3 3 EXT_CLK/INT External ADC clock input or CPU interrupt output signal. When the pin is configured as an interrupt, the
output is an open drain.
4 4 SDA/SMBDAT I2C serial data input/output.
5 5 SCL/SMBCLK I2C clock input
6 9 VCC Positive power pin. The positive power supply to the part.
7 10 GND Negative power pin. Can be connected to ground or a negative voltage.
8 11 VBUS VBUS power voltage sense.
9 12 VINM Current sense minus input.
10 13 VINP Current sense plus input.
6, 7, 8, 14,
15, 16
NC No connect. No internal connection.
Epad GND Negative power pin. Can be connected to ground or a negative voltage.
ISL28022
FN8386 Rev 8.00 Page 5 of 32
February 4, 2016
TABLE 1. DPM PORTFOLIO COMPARISON - ISL28022 vs ISL28023 vs ISL28025
DESCRIPTION
BASIC DIGITAL
POWER MONITOR
FULL FEATURE
DIGITAL POWER MONITOR
DIGITAL POWER MONITOR
IN TINY PACKAGE
PART NUMBER ISL28022 ISL28023 ISL28025
PACKAGE MSOP10, QFN16 QFN24 WLCSP-16
Temperature Range -40°C to +125°C -40°C to +125°C -40°C to +125°C
0V to 60V Input Range 0V to 60V Opt 1: 0V to 60V
Opt 2: 0V to 16V
Opt 1: 0V to 60V
Opt 2: 0V to 16V
ADC 16-bit 16-bit 16-bit
+25°C Gain Error 0.30% 0.25% 0.25%
Current Measure LSB Step 10µV 2.5µV 2.5µV
+25°C Offset 75µV 30µV 30µV
Primary Differential Shunt Input X X X
Channel Independent Bus Voltage X X X
LV Aux Differential Shunt Input X
Channel Independent Bus Voltage X X
VBus LSB Step Low Voltage Bus 0.25mV 0.25mV
High Voltage Bus 4mV 1mV/0.25mV 1mV/0.25mV
External Temperature Sensor Input X
HV Internal Regulator (3.3VOUT)XX
Fast OC/OV/UV Alert Outputs 2 Outputs 2 Outputs
Margin DAC X
Internal Temperature Sensor X X
User Select Conversion Mode/Sample Rate X X X
Peak Min/Max Current Registers X X
Slave Address Locations 16 Addresses 55 Addresses 55 Addresses
I2C Level Translators XX
PMBus XX
I2C/SMBus X X X
High Speed (3.4MHz) I2C Mode X X X
External Clock Input X X X
Power Shutdown Mode X X X
ISL28022
FN8386 Rev 8.00 Page 6 of 32
February 4, 2016
Absolute Maximum Ratings Thermal Information
VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6.0V
VBUS Voltage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63V
Common-Mode Input Voltage (VINP, VINM) . . . . . . . . . . . . . . . . . . . . . . 63V
Differential Input Voltage (VINP, VINM) . . . . . . . . . . . . . . . . . . . . . . . . . ±63V
Input Voltage (Digital Pins) . . . . . . . . . . . . . . . . . . . . . . (GND - 0.3V) to 5.5V
Output Voltage (Digital Pins) . . . . . . . . . . . . . . . . (GND - 0.3V) to VCC + 0.3V
Open-Drain Output Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10mA
Open-Drain Voltage (Interrupt) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24V
ESD Rating
Human Body Model (Tested per JESD22-A114) . . . . . . . . . . . . . . . . . 8kV
Machine Model (Tested per JESD22-A115). . . . . . . . . . . . . . . . . . . . 400V
Charged Device Model (Tested per JESD22-C101). . . . . . . . . . . . . . . 2kV
Latch-Up (Tested per JESD-78B) . . . . . . . . . . . . . . . . . . . . . . 60V at +125°C
Thermal Resistance (Typical) JA (°C/W) JC (°C/W)
16 Ld QFN (Notes 4, 5) . . . . . . . . . . . . . . . . 52 6.5
10 Ld MSOP (Notes 6, 7) . . . . . . . . . . . . . . . 150 55
Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C
Maximum Junction Temperature (TJMAX) . . . . . . . . . . . . . . . . . . . . .+150°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see TB493
Recommended Operating Conditions
Ambient Temperature Range (TA) . . . . . . . . . . . . . . . . . . .-40°C to +125°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
5. For JC, the “case temp” location is the center of the exposed metal pad on the package underside.
6. JA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
7. For JC, the “case temp” location is taken at the package top center.
Electrical Specifications TA = +25°C, VCC = 3.3, VINP = VBUS = 12V, VSENSE = VINP-VINM = 32mV, unless otherwise specified. All
voltages with respect to GND pin.
PARAMETER DESCRIPTION TEST CONDITIONS
MIN
(Note 8)TYP
MAX
(Note 8)UNIT
INPUTS
VSENSEDIFF Useful Full-Scale Current Sense
Differential Voltage Range (VINP-VINM)
PGA gain = /1 0 ±40 mV
PGA gain = /2 0 ±80 mV
PGA gain = /4 0 ±160 mV
PGA gain = /8 0 ±320 mV
VSHUNT_step LSB Step Size, Shunt Voltage 10 µV
VCMSENSE Current Sense Common-Mode
(VINP, VINM)
060V
VOS VSENSE Offset Voltage PGA gain = /1, /2, /4, /8;
ADC setting = 1111
±10 ±75 µV
VOSTC VSENSE Offset Voltage Temperature
Coefficient
0.15 µV/°C
CMRR VSENSE VOS vs Common-Mode VBUS = 0V to 60V; BRNG = 2, 3 110 130 dB
PSRR VSENSE VOS vs Power Supply VCC = 3V to 5V 105 dB
ACS Current Sense Gain Error ±40 m%
ACSTC Current Sense Gain Error Temperature
Coefficient
±1 m%/°C
IVINACT Input Leakage, VIN Pins Active mode
(for both VINP and VINM pins)
±20 µA
IVINACT Input Leakage, VIN Pins Power-down mode
(for both VINP and VINM pins)
±0.1 ±0.5 µA
VBUS Useful Bus Voltage Range BRNG = 0 0 16 V
BRNG = 1 0 32 V
BRNG = 2, 3 0 60 V
VBUS_Step LSB Step Size, Bus Voltage BRNG = 0 4 mV
VBUS_VCO VBUS Voltage Coefficient 50 ppm/V
RVBACT Input Impedance, VBUS Pin Active mode 600 kΩ
ISL28022
FN8386 Rev 8.00 Page 7 of 32
February 4, 2016
DC ACCURACY
ADC Resolution (Native) PGA gain = /1, VSENSE = ±320mV 16 Bits
Current Measurement Error TA = +25°C ±0.2 ±0.3 %
Current Measurement Error
Over-Temperature
TA = -40°C to +85°C ±0.5 %
TA = -40°C to +125°C ±1 %
Bus Voltage Measurement Error TA = +25°C ±0.2 ±0.3 %
Bus Voltage Measurement Error
Over-Temperature
TA = -40°C to +85°C ±0.5 %
TA = -40°C to +125°C ±1 %
ADC TIMING SPECS
tsADC Conversion Time
Mode = 5 or 6
ADC setting = 0000 72.0 79.2 µs
ADC setting = 0001 132.0 145.2 µs
ADC setting = 0010 258.0 283.8 µs
ADC setting = 0011 508.0 558.8 µs
ADC setting = 1001 1.01 1.11 ms
ADC setting = 1010 2.01 2.21 ms
ADC setting = 1011 4.01 4.41 ms
ADC setting = 1100 8.01 8.81 ms
ADC setting = 1101 16.01 17.61 ms
ADC setting = 1110 32.01 35.21 ms
ADC setting = 1111 64.01 70.41 ms
I2C INTERFACE SPECIFICATIONS
VIL SDA and SCL Input Buffer LOW Voltage -0.3 0.3 x VCC V
VIH SDA and SCL Input Buffer HIGH Voltage 0.7 x VCC VCC + 0.3 V
Hysteresis SDA and SCL Input Buffer Hysteresis 0.05 x VCC V
VOL SDA Output Buffer LOW Voltage, Sinking
3mA
VCC = 5V, IOL = 3mA 0 0.02 0.40 V
CPIN SDA and SCL Pin Capacitance TA = +25°C, f = 1MHz,
VCC = 5V, VIN = 0V,
VOUT = 0V
10 pF
fSCL SCL Frequency 400 kHz
tIN Pulse Width Suppression Time at SDA
and SCL Inputs
Any pulse narrower than the
maximum spec is suppressed
50 ns
tAA SCL Falling Edge to SDA Output Data
Valid
SCL falling edge crossing 30% of VCC,
until SDA exits the 30% to 70% of VCC
window.
900 ns
tBUF Time the Bus Must be Free Before the
Start of a New Transmission
SDA crossing 70% of VCC during a
STOP condition, to SDA crossing 70%
of VCC during the following START
condition.
1300 ns
tLOW Clock LOW Time Measured at the 30% of VCC crossing 1300 ns
tHIGH Clock HIGH Time Measured at the 70% of VCC crossing 600 ns
tSU:STA START Condition Setup Time SCL rising edge to SDA falling edge
Both crossing 70% of VCC
600 ns
tHD:STA START Condition Hold Time From SDA falling edge crossing 30%
of VCC to SCL falling edge crossing
70% of VCC
600 ns
Electrical Specifications TA = +25°C, VCC = 3.3, VINP = VBUS = 12V, VSENSE = VINP-VINM = 32mV, unless otherwise specified. All
voltages with respect to GND pin. (Continued)
PARAMETER DESCRIPTION TEST CONDITIONS
MIN
(Note 8)TYP
MAX
(Note 8)UNIT
ISL28022
FN8386 Rev 8.00 Page 8 of 32
February 4, 2016
tSU:DAT Input Data Setup Time From SDA exiting the 30% to 70% of
VCC window, to SCL rising edge
crossing 30% of VCC
100 ns
tHD:DAT Input Data Hold Time From SCL falling edge crossing 30%
of VCC to SDA entering the 30% to
70% of VCC window
20 900 ns
tSU:STO STOP Condition Setup Time From SCL rising edge crossing 70% of
VCC, to SDA rising edge crossing 30%
of VCC
600 ns
tHD:STO STOP Condition Hold Time From SDA rising edge to SCL falling
edge. Both crossing 70% of VCC.
600 ns
tDH Output Data Hold Time From SCL falling edge crossing 30%
of VCC, until SDA enters the 30% to
70% of VCC window
0ns
tRSDA and SCL Rise Time From 30% to 70% of VCC 20 + 0.1
x Cb
300 ns
tFSDA and SCL Fall Time From 70% to 30% of VCC 20 + 0.1
x Cb
300 ns
Cb Capacitive Loading of SDA or SCL Total on-chip and off-chip 75 pF
RPU SDA and SCL Bus Pull-Up Resistor
Off-Chip
Maximum is determined by tR and tF
For Cb = 400pF, maximum is about
2kΩ~2.5kΩ
For Cb = 40pF, maximum is about
15kΩ~20kΩ
1k
POWER SUPPLY
Operating Supply Voltage Range 3 5.5 V
ICCEXT Power Supply Current On VCC Pin, Active
Mode
External power supply mode,
VCC = 5V
0.7 1.0 mA
ICCPD Power Supply Current On VCC Pin,
Power-Down Mode
External power supply mode,
VCC = 5V
515µA
NOTE:
8. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
Electrical Specifications TA = +25°C, VCC = 3.3, VINP = VBUS = 12V, VSENSE = VINP-VINM = 32mV, unless otherwise specified. All
voltages with respect to GND pin. (Continued)
PARAMETER DESCRIPTION TEST CONDITIONS
MIN
(Note 8)TYP
MAX
(Note 8)UNIT
ISL28022
FN8386 Rev 8.00 Page 9 of 32
February 4, 2016
Typical Performance Curves TA = +25°C, VCC = 3.3V, VINP = VBUS = 12V, S(B)ADC = 15;
unless otherwise specified.
FIGURE 3. VSHUNT VOS FIGURE 4. VSHUNT VOS vs TEMPERATURE
FIGURE 5. VSHUNT MEASUREMENT ERROR FIGURE 6. VSHUNT MEASUREMENT ERROR vs VSHUNT INPUT
FIGURE 7. VSHUNT GAIN vs TEMPERATURE FIGURE 8. VBUS MEASUREMENT ERROR DISTRIBUTION
0
5
10
15
20
25
-75 -60 -45 -30 -15 0 15 30 45 60 75
HITS
VSHUNT VOS (µV)
-0.0750
-0.0625
-0.0500
-0.0375
-0.0250
-0.0125
0
0.0125
0.0250
0.0375
0.0500
0.0625
0.0750
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
VOS (mV)
VCC = 3.3V
VCC = 3V VCC = 5V
SADC = 15
0
5
10
15
20
25
30
35
40
-0.30 -0.25 -0.20 -0.15 -0.10 -0.05 0 0.05 0.10 0.15 0.20 0.25 0.30
HITS
VSHUNT MEASUREMENT ERROR (%)
-0.3
-0.2
-0.1
0
0.1
0.2
0.3
-0.30 -0.25 -0.20 -0.15 -0.10 -0.05 0 0.05 0.10 0.15 0.20 0.25 0.30
VSHUNT (V)
VSHUNT MEASUREMENT ERROR (%)
VCC = 5.5V
VCC = 3.3V VCC = 3V
T = +25°C
VSHUNT (CMV) = 12V
SADC = 15
-1.0
-0.8
-0.6
-0.4
-0.2
0
0.2
0.4
0.6
0.8
1.0
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
GAIN ERROR (%)
VCC = 5.5V
VCC = 3.3V
VCC = 3V
VSHUNT (DIFF) = 32mV
VSHUNT (CMV) = 12V
SADC = 15
0
10
20
30
40
50
60
70
-0.30 -0.25 -0.20 -0.15 -0.10 -0.05 0 0.05 0.10 0.15 0.20 0.25 0.30
HITS
VBUS MEASUREMENT ERROR (%)
ISL28022
FN8386 Rev 8.00 Page 10 of 32
February 4, 2016
FIGURE 9. VBUS MEASUREMENT ERROR vs VBUS (TA = +25°C) FIGURE 10. VBUS MEASUREMENT ERROR vs TEMPERATURE
FIGURE 11. CMRR vs TEMPERATURE FIGURE 12. SUPPLY CURRENT vs MODE vs TEMPERATURE
FIGURE 13. SUPPLY CURRENT vs MODE vs VCC FIGURE 14. SUPPLY CURRENT vs MODE 0 vs TEMPERATURE
Typical Performance Curves TA = +25°C, VCC = 3.3V, VINP = VBUS = 12V, S(B)ADC = 15;
unless otherwise specified. (Continued)
-1.0
-0.8
-0.6
-0.4
-0.2
0
0.2
0.4
0.6
0.8
1.0
0 8 16 24 32 40 48 56 64
VBUS (V)
VBUS MEASUREMENT ERROR (%)
VCC = 5.5V
VCC = 3.3V
VCC = 3V
-1.0
-0.8
-0.6
-0.4
-0.2
0
0.2
0.4
0.6
0.8
1.0
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
VBUS MEASUREMENT ERROR (%)
VCC = 5.5V
VCC = 3.3V
VCC = 3V
120
125
130
135
140
145
150
155
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
VCC = 3V
VCC = 3.3V
VCC = 5V
VSHUNT (DCMV) = 0V TO 60V
SADC = 15
CMRR (dB)
300
350
400
450
500
550
600
650
700
750
800
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
SUPPLY CURRENT (µA)
MODE = 7
MODE = 4
300
350
400
450
500
550
600
650
700
750
800
3.0 3.5 4.0 4.5 5.0 5.5 6.0
VCC (V)
SUPPLY CURRENT (µA)
MODE = 7
MODE = 4
0
2
4
6
8
10
12
14
16
18
20
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
SUPPLY CURRENT (µA)
ISL28022
FN8386 Rev 8.00 Page 11 of 32
February 4, 2016
FIGURE 15. SUPPLY CURRENT vs MODE 0 vs VCC FIGURE 16. SHUNT IVIN vs TEMPERATURE (MODE 5)
FIGURE 17. SHUNT IVIN vs COMMON-MODE VOLTAGE (MODE 5) FIGURE 18. SHUNT IVIN vs TEMPERATURE (MODE 0, 4)
FIGURE 19. SHUNT IVIN vs COMMON-MODE VOLTAGE (MODE 0, 4) FIGURE 20. SHUNT IOS vs TEMPERATURE (MODE 5)
Typical Performance Curves TA = +25°C, VCC = 3.3V, VINP = VBUS = 12V, S(B)ADC = 15;
unless otherwise specified. (Continued)
0
2
4
6
8
10
12
14
16
18
20
3.0 3.5 4.0 4.5 5.0 5.5 6.0
VCC (V)
SUPPLY CURRENT (µA)
5
7
9
11
13
15
17
19
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
IVIN (µA)
5
6
7
8
9
10
11
12
13
14
15
0 8 16 24 32 40 48 56 64
VCM (V)
IVIN (µA)
0
0.005
0.010
0.015
0.020
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
IVIN (µA)
MODE = 4
MODE = 0
0
0.005
0.010
0.015
0.020
0 8 16 24 32 40 48 56 64
VCM (V)
MODE = 0
MODE = 4
IVIN (µA)
-0.5
-0.4
-0.3
-0.2
-0.1
0
0.1
0.2
0.3
0.4
0.5
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
IOS (µA)
ISL28022
FN8386 Rev 8.00 Page 12 of 32
February 4, 2016
FIGURE 21. SHUNT IOS vs COMMON-MODE VOLTAGE (MODE 5) FIGURE 22. SHUNT IOS vs TEMPERATURE (MODE 0, 4)
FIGURE 23. SHUNT IOS vs COMMON-MODE VOLTAGE (MODE 0, 4) FIGURE 24. VSHUNT BANDWIDTH vs SADC MODE
FIGURE 25. VSHUNT BANDWIDTH vs EXTERNAL CLOCK FREQUENCY FIGURE 26. INTERRUPT TIMING
Typical Performance Curves TA = +25°C, VCC = 3.3V, VINP = VBUS = 12V, S(B)ADC = 15;
unless otherwise specified. (Continued)
-0.5
-0.4
-0.3
-0.2
-0.1
0
0.1
0.2
0.3
0.4
0.5
0 8 16 24 32 40 48 56 64
VCM (V)
IOS (µA)
-0.0020
-0.0015
-0.0010
-0.0005
0
0.0005
0.0010
0.0015
0.0020
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
IOS (µA)
MODE = 4
MODE = 0
-0.0020
-0.0015
-0.0010
-0.0005
0
0.0005
0.0010
0.0015
0.0020
0 8 16 24 32 40 48 56 64
VCM (V)
IOS (µA)
MODE = 4
MODE = 0
-50
-40
-30
-20
-10
0
10
10 100 1k 10k
FREQUENCY (Hz)
GAIN (dB)
SADC = 1
SADC = 0
SADC = 2
SADC = 3
VIN = 200mVP-P SINE WAVE
-50
-40
-30
-20
-10
0
10
10 100 1k 10k
FREQUENCY (Hz)
SADC = 3
F_EXTCLK = OFF
SADC = 3
F_EXTCLK = 768kHz
SADC = 3
F_EXTCLK = 384kHz
VIN = 200mVP-P SINE WAVE
GAIN (dB)
-0.2 -0.1 00.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8
TIME (ms)
S(B)ADC = 508µs
S(B)ADC = 256µs
S(B)ADC = 72µs
S(B)ADC = 132µs
MODE = 5 OR 6
INPUT
SIGNAL
ISL28022
FN8386 Rev 8.00 Page 13 of 32
February 4, 2016
Functional Description
Overview
The ISL28022 is a Digital Power Monitor (DPM) device that is
capable of measuring bidirectional currents while monitoring the
bus voltage.
The DPM requires an external shunt resistor to enable current
measurements. The shunt resistor translates the bus current to a
voltage. The DPM measures the voltage across the shunt
resistors and reports the measured value out digitally via an I2C
interface. A register within the DPM is reserved to store the value
of the shunt resistor. The stored current sense resistor value
allows the DPM to output the current value to an external digital
device.
The ISL28022 measures bus voltage and current sequentially.
The device has a power measurement functionality that
multiplies current and voltage measured values. The power
calculation is stored in a unique register. The power
measurement allows the user to monitor power to or from the
load in addition to current and voltage.
The ISL28022 can monitor supplies from 0V to 60V while
operating on a chip supply ranging from 3V to 5.5V.
The ISL28022 ADC sample rate can be configured to an internal
oscillator (500kHz) or a user can provide a synchronized clock.
Detailed Description
The ISL28022 consists of a two channel analog front end
multiplexer, a 16-bit sigma delta ADC and digital signal
processing/serial communication circuitry.
The main block within the device is a 3rd order Sigma Delta ADC.
The input signal bandwidth is 1kHz, wide enough for power
monitoring applications. The main block includes an internal
1.2V bandgap voltage reference that is used to drive the ADC.
The analog front end multiplexer selects the input to the ADC.
The selection to the input of the ADC is either a single-ended
VBUS measurement or a fully differential measurement across a
shunt resistor.
The digital block contains controllable registers, I2C serial
communication circuitry and a state machine. The state machine
controls the behavior of the ADC acquisition, whether the
acquisition is triggered or continuous. A more detailed
description of the state machine states can be found in “MODE:
Operating Mode” on page 15.
Functional Pin Descriptions
A1
A1 is the address select pin. A1 is one of two I2C/SMBus slave
address select pins that are multilogic programmable for a total
of 16 different address combinations.
There are four selectable levels for A1, VCC, GND, SCL/SMBCLK,
and SDA/SMBDAT. See Table 22 for more details in setting the
slave address of the device.
A0
A0 is the address select pin. A0 is one of two I2C/SMBus slave
address select pins that are multilogic programmable for a total
of 16 different address combinations.
There are four selectable levels for A0, VCC, GND, SCL/SMBCLK,
and SDA/SMBDAT. See Table 22 for more details in setting the
slave address of the device.
EXT_CLK/INT
EXT_CLK/INT is the External/Interrupt clock pin. EXT_CLK/INT is
a bidirectional pin. The pin provides a connection to the system
clock. The system clock is connected to the ADC. The
acquisitions rate of the ADC can be varied through the
EXT_CLK/INT pin. The pin functionality is set through a control
register bit.
When the EXT_CLK/INT pin is configured as an output, the pin
functionality becomes an interrupt flag to connecting devices.
EXT_CLK/INT pin as an output requires a pull-up resistor to a
power supply, up to 20V, for proper operation. The internal
threshold detectors (OVsh/UVsh/OVb/UVb) signal level relative to
the measured value determines the state of the INT pin.
SDA/SMBDAT
SDA/SMBDAT is the serial data input/output pin. SDA/SMBDAT
is a bidirectional pin used to transfer data to and from the device.
The pin is an open-drain output and may be wired with other
open-drain/collector outputs. The open-drain output requires a
pull-up resistor for proper functionality. The pull-up resistor
should be connected to VCC of the device.
SCL/SMBCLK
SCL/SMBCLK is the serial clock input pin. The SCL/SMBCLK
input is responsible for clocking in all data to and from the
device.
VCC
VCC is the positive supply voltage pin. VCC is an analog power
pin. VCC supplies power to the device.
GND
GND is the ground pin. All voltages internal to the chip are
referenced to ground. GND should be tied to 0V for single supply
applications. For dual supply applications, the pin should be
connected to the most negative voltage in the application.
VBUS
VBUS is the power bus voltage input pin. The pin should be
connected to the desired power supply bus to be monitored.
VINP
VINP is the shunt voltage monitor positive input pin. The pin
connects to the most positive voltage of the current shunt
resistor.
VINM
VINM is the shunt voltage monitor negative input pin. The pin
connects to the most negative voltage of the current shunt
resistor.
ISL28022
FN8386 Rev 8.00 Page 14 of 32
February 4, 2016
Register Descriptions
Table 2 is the register map for the device. The table describes the
function of each register and its respective value. The addresses
are sequential and the register size is 16 bits (2 bytes) per
address.
CONFIGURATION REGISTER
The configuration register (Table 3) controls the functionality of
the chip. ADC measurable range, converter acquisition times,
converter resolution and state machine modes are configurable
bits within this register.
RST: Reset Bit
Configuring the reset bit (Bit 15) to a 1 generates a system reset
that initializes all registers to their default values and performs a
system calibration.
BRNG: Bus Voltage Range
Bits 13 and 14 of the configuration register sets the bus
measurable voltage range. Table 4 shows the BRNG bit
configurations versus the allowable full-scale measurement
range. The shaded row is the power-up default.
PG: PGA (Shunt Voltage Only)
Bits 11 and 12 of the configuration register determines the shunt
voltage measurement range. Table 5 shows the PGA bit
configurations versus the allowable full-scale measurement
range. The shaded row is the power-up default.
TABLE 2. ISL28022 REGISTER DESCRIPTIONS
REGISTER
ADDRESS (HEX) REGISTER NAME FUNCTION
POWER-ON RESET VALUE
(HEX) ACCESS
00 Configuration Power-on reset, bus and shunt ranges, ADC
acquisition times, mode configuration
799F R/W
01 Shunt Voltage Shunt voltage measurement value 0000 R
02 Bus Voltage Bus voltage measurement value 0000 R
03 Power Power measurement value 0000 R
04 Current Current measurement value 0000 R
05 Calibration Register Register used to enable current and power
measurements.
0000 R/W
06 Shunt Voltage Threshold Min/Max shunt thresholds 7F81 R/W
07 Bus Voltage Threshold Min/Max VBUS thresholds FF00 R/W
08 DCS Interrupt Status Threshold interrupts 0000 R/W
09 Aux Control Register Register to control the interrupts and
external clock functionality
0000 R/W
TABLE 3. CONFIGURATION REGISTER
BIT D15 D14 D13 D12 D11 D10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0
NAME RST BRNG1 BRNG0 PG1 PG0 BADC3 BADC2 BADC1 BADC0 SADC3 SADC2 SADC1 SADC0 MODE2 MODE1 MODE0
TABLE 4. BRNG BIT SETTINGS
BRNG1 BRNG0
USABLE FULL
SCALE RANGE (V)
00 16
01 32
10 60
1 1 60
TABLE 5. PGA BIT SETTINGS
PG1 PG0 GAIN
RANGE
(mV)
001±40
0 1 ÷2 ±80
1 0 ÷4 ±160
1 1 ÷8 ±320
ISL28022
FN8386 Rev 8.00 Page 15 of 32
February 4, 2016
BADC: Bus ADC Resolution/Averaging
Bits [10:7] of the configuration register sets the ADC resolution/
averaging when the ADC is configured in the VBUS mode. The
ADC can be configured versus bit accuracy. The bit accuracy
selections range from 12 to 15 bits. The ADC is configurable
versus the number of averages. The selection ranges from 2 to
128 samples. Table 6 shows the breakdown of each BADC
setting. The shaded row is the default setting upon power-up.
SADC: Shunt ADC Resolution/Averaging
Bits [10:7] of the configuration register sets the ADC resolution/
averaging when the ADC is configured in the VSHUNT mode. The
ADC can be configured versus bit accuracy. The bit accuracy
selections range from 12 to 15 bits. The ADC is configurable
versus number of averages. The selection ranges from 2 to 128
samples. Table 6 shows the breakdown of each SADC setting.
The shaded row is the default setting upon power-up.
MODE: Operating Mode
Bits [2:0] of the configuration register controls the state machine
within the chip. The state machine globally controls the overall
functionality of the chip. Table 7 shows the various states the
chip can be configured to, as well as the mode bit definitions to
achieve a desired state. The shaded row is the default setting upon
power-up.
TABLE 6. ADC SETTINGS, APPLIES TO BOTH SADC AND BADC CONTROL
ADC3 ADC2 ADC1 ADC0 MODE/SAMPLES CONVERSION TIME
0 X 0 0 12-bit 72µs
0 X 0 1 13-bit 132µs
0 X 1 0 14-bit 258µs
0 X 1 1 15-bit 508µs
1 0 0 0 15-bit 508µs
100121.01ms
101042.01ms
101184.01ms
1100168.01ms
11013216.01ms
11106432.01ms
1 1 1 1 128 64.01ms
TABLE 7. OPERATING MODE SETTINGS
MODE2 MODE1 MODE0 MODE
000Power-down
0 0 1 Shunt voltage, triggered
0 1 0 Bus voltage, triggered
0 1 1 Shunt and bus, triggered
1 0 0 ADC off (disabled)
1 0 1 Shunt Voltage, continuous
1 1 0 Bus voltage, continuous
1 1 1 Shunt and bus, continuous
ISL28022
FN8386 Rev 8.00 Page 16 of 32
February 4, 2016
SHUNT VOLTAGE REGISTER 01H (READ-ONLY)
The shunt voltage register reports the measured value across the
shunt pins (VINP and VINM) into the register. The shunt register
LSB is independent of PGA range settings. The PGA setting for
the shunt register masks the unused most significant bit with a
sign bit. For lower range of PGA settings, multiple sign bits are
returned by the DPM. Only one sign bit should be used to
calculate the measured value.
Tables 8 through 11 show the weights of each bit for various PGA
ranges. The tables should be used to calculate the measured
value across the shunt pins from the binary to decimal domains.
To calculate the measured decimal value across the shunt, first
read the shunt voltage register. Assume the PGA setting is set to
the 80mV range. For this example, the reading output by the chip
is 1111 1010 0000 0101. The 80mV range has three sign bits.
Only one sign bit needs to be used to calculate the measured
decimal value. Bits 14 and 15 are omitted from the calculation.
This leaves a binary reading of 11 1010 0000 0101.
Next, multiply each bit by its respective weight. Bit0 value would
be multiplied by Bit0 weight (1), Bit1 value*Bit1 weight (2), etc.
Add all the multiplied values to equate to a single number. For
the binary reading 11 1010 0000 0101 this equates to -1531.
The LSB for a shunt register is 10µV. Multiplying the decimal
value by the LSB weight yields the measured voltage across the
shunt. A 1111 1010 0000 0101 reading equals -15.31mV
measured across the shunt pins.
TABLE 8. SHUNT VOLTAGE REGISTER, PG GAIN = /8 (RANGE = 11), FULL-SCALE = ±320mV, 15 BITS WIDE
BIT D15D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAME Sign Bit14 Bit13 Bit12 Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0
WEIGHT -32768 16384 8192 4096 2048 1024 512 256 128 64 32 16 8 4 2 1
TABLE 9. SHUNT VOLTAGE REGISTER, PG GAIN = /4 (RANGE = 10), FULL-SCALE = ±160mV, 14 BITS WIDE
BIT D15 D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAME Sign Sign Bit13 Bit12 Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0
WEIGHT -16384 8192 4096 2048 1024 512 256 128 64 32 16 8421
TABLE 10. SHUNT VOLTAGE REGISTER, PG GAIN = /2 (RANGE = 01), FULL-SCALE = ±80mV, 13 BITS WIDE
BITD15D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAME Sign Sign Sign Bit12 Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0
WEIGHT -8192 4096 2048 1024 512 256 128 64 32 16 8421
TABLE 11. SHUNT VOLTAGE REGISTER, PG GAIN = /1 (RANGE = 00), FULL-SCALE = ±40mV, 12 BITS WIDE
BITD15D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAME Sign Sign Sign Sign Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0
WEIGHT -4096 2048 1024 512 256 128 64 32 16 8 4 2 1
TABLE 12. BUS VOLTAGE REGISTER, BRNG = 10 OR 11, FULL-SCALE = 60V, 14 BITS WIDE
BITD15D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAME Bit13 Bit12 Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0 CNVR OVF
WEIGHT 8192 4096 2048 1024 512 256 128 64 32 16 8421
TABLE 13. BUS VOLTAGE REGISTER, BRNG = 01, FULL-SCALE = 32V, 13 BITS WIDE
BITD15D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAME Bit12 Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0 CNVR OVF
WEIGHT 4096 2048 1024 512 256 128 64 32 16 8 4 2 1
TABLE 14. BUS VOLTAGE REGISTER, BRNG = 00, FULL-SCALE = 16V, 12 BITS WIDE
BITD15D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAME Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0 CNVR OVF
WEIGHT 2048 1024 512 256 128 64 32 16 8 4 2 1
TABLE 15. CALIBRATION REGISTER, 05h
BITD15D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAME FS15 FS14 FS13 FS12 FS11 FS10 FS9 FS8 FS7 FS6 FS5 FS4 FS3 FS2 FS1 0
ISL28022
FN8386 Rev 8.00 Page 17 of 32
February 4, 2016
BUS VOLTAGE REGISTER 02h (READ-ONLY)
The bus voltage register is where the DPM reports the measured
value of the VBUS. There are three scale ranges possible
depending on the BRNG setting controlled from the configuration
register (00h).
Tables 12 through 14 on page 16 are the weight bits for each
BRNG setting. The binary value recorded in the Bus Voltage
register is translated to a decimal value in the same way as the
shunt voltage register is converted to a decimal value.
Equation 1 is the mathematical equation for converting the
binary VBUS value to a decimal value. N is the bit number. The
LSB value for the VBUS measurement equals 4mV across all bus
range (BRNG) settings.
CNVR: Conversion Ready (Bit 1)
The conversion ready bit indicates when the ADC has finished a
conversion and transferred the reading(s) to the appropriate
register(s). The CNVR is only operable when the DPM is set to one
of three trigger modes. The CNVR is at a low state when the
conversion is in progress. The CNVR transitions and remains at a
high state when the conversion is complete.
The CNVR bit is initialized or reinitialized in the following ways:
1. Writing to the configuration register.
2. Reading from power register.
OVF: Math Overflow Flag (Bit0)
The Math Overflow Flag (OVF) is a bit that is set to indicate the
current or power data being read from the DPM is over-ranged
and meaningless.
CALIBRATION REGISTER 05h (READ/WRITE)
To accurately read the current and power measurements from
the chip, the calibration register needs to be programmed.
The calibration register value is calculated as follows:
1. Calculate the full-scale current range that is desired. This is
calculated using Equation 2. Rshunt is the value of the shunt
resistor. VshuntFS is the full-scale setting that is desired. In
most cases, it is the PGA full-scale range (320mV, 160mV,
80mV and 40mV) that the DPM is programmed to.
2. From the current full-scale range, the current LSB is
calculated using Equation 3. Current full-scale is the outcome
from Equation 2. ADCres is the resolution of shunt voltage
reading. The value is determined by the SADC setting in
configuration register. SADC setting equal to 3 and greater
will have a 15-bit resolution. The ADCres value equals 215 or
32768.
3. From Equation 3, the calibration resister value is calculated
using Equation 4. The resolution of the math that is processed
internally in the DPM is 4096 or 12 bits of resolution. The
VshuntLSB is set to 10µV. Equation 4 yields a 16-bit binary
number that can be written to the calibration register. The
calibration value can only be 15 bits due to the ADCres value.
Bit 0 of the calibration register is fixed to a value of 0. The
calibration register format is represented in Table 15.
CURRENT REGISTER 04h (READ-ONLY)
Once the calibration register (05h) is programmed, the output
current is calculated using Equation 5:
Bit is the returned value of each bit from the current register
either 1 or a 0. The weight of each bit is represented in Table 16.
n is the bit number. The current LSB is the value calculated from
Equation 3.
POWER REGISTER 03h (READ-ONLY)
The Power register only has meaning if the calibration register
(05h) is programmed. The units for the power register are in
watts. The power is calculated using Equation 6:
Bit is the returned value of each bit from the power register either
1 or a 0. The weight of each bit is represented in Table 17. n is
the bit number. The power LSB is calculated from Equation 7:
If VBUS range, BRNG, is set to 60V, the power equation in
Equation 6 is multiplied by 2.
THRESHOLD REGISTERS
The Shunt Voltage or VBUS threshold registers are used to set the
Min/Max threshold limits that will be tested versus VSHUNT or
VBUS readings. Measurement readings exceeding the respective
VSHUNT or VBUS limits, either above or below, will set a register
flag and perhaps an external interrupt depending on the
configuration of the Interrupt Enable bit (INTREN) in register 09h.
The testing of the ADC reading versus the respective threshold
limits occurs once per ADC conversion.
(EQ. 1)
Vbus
2
15
n
BitnBit_Weight n
Vbus LSB
(EQ. 2)
Current FS
Vshunt FS
Rshunt
(EQ. 3)
Current LSB
Current FS
ADC res
(EQ. 4)
CalRegval integer
Math res VshuntLSB
CurrentLSB Rshunt
CalRegval integer 0.04096
CurrentLSB Rshunt
Current
0
15
n
BitnBit_Weight n
Current LSB
(EQ. 5)
(EQ. 6)
Power
0
15
n
BitnBit_Weightn
Power LSB
5000
Power LSB Current LSB Vbus LSB
(EQ. 7)
ISL28022
FN8386 Rev 8.00 Page 18 of 32
February 4, 2016
SHUNT VOLTAGE THRESHOLD REGISTER 06h
(READ/WRITE)
The VSHUNT minimum and maximum threshold limits are set
using one register. The shunt value readings are either positive or
negative. D15 and D7 bits of Table 18 are given to represent the
sign of the limit. SMX bits represent the upper limit threshold.
SMN represents the lower threshold limit. Equation 8 is the
calculation used to convert the VSHUNT threshold binary value to
decimal. Bit is the value of each bit set in the shunt threshold
register. The value is either 1 or a 0. The weight of each bit is
represented in Table 18. n is the bit number. The shunt voltage
threshold LSB is 2.56mV.
Vs thresh
0
7
n
BitnBit_Weightn
VsThresh LSB
(EQ. 8)
TABLE 16. CURRENT REGISTER, 04h
BITD15D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAMEBit 15Bit14Bit13Bit12Bit11Bit10Bit9Bit8Bit7Bit6Bit5Bit4Bit3Bit2Bit1Bit0
WEIGHT -32768 16384 8192 4096 2048 1024 512 256 128 64 32 16 8 4 2 1
TABLE 17. POWER REGISTER, 03h
BIT D15 D14 D13 D12 D11 D10 D9D8D7D6D5D4D3D2D1D0
NAME PD15 PD14 PD13 PD12 PD11 PD10 PD9 PD8 PD7 PD6 PD5 PD4 PD3 PD2 PD1 PD0
WEIGHT 32768 16384 8192 4096 2048 1024 512 256 128 64 32 16 8421
TABLE 18. SHUNT VOLTAGE THRESHOLD REGISTER, 06h
BITD15D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAME Sign SMX6 SMX5 SMX4 SMX3 SMX2 SMX1 SMX0 Sign SMN6 SMN5 SMN4 SMN3 SMN2 SMN1 SMN0
WEIGHT -128 64 32 16 8 4 2 1 -128 64 32 16 8 4 2 1
TABLE 19. BUS VOLTAGE THRESHOLD REGISTER, 07h
BITD15D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAME BMX7 BMX6 BMX5 BMX4 BMX3 BMX2 BMX1 BMX0 BMN7 BMN6 BMN5 BMN4 BMN3 BMN2 BMN1 BMN0
WEIGHT 128 64 32 16 8 4 2 1 128 64 32 16 8 4 2 1
TABLE 20. INTERRUPT STATUS REGISTER, 08h
BITD15D14D13D12D11D10D9D8D7D6D5D4D3D2D1D0
NAME NA NA NA NA NA NA NA NA NA NA NA NA SMXW SMNW BMXW BMNW
WEIGHT0000000000000000
TABLE 21. AUX CONTROL REGISTER, 09h
BITD15D14D13D12D11D10D9 D8 D7 D6 D5D4D3D2D1D0
NAME NA NA NA NA NA NA NA FORCEINTR INTREN ExtClkEn ExtCLKDiv[5:0]
WEIGHT0000000 0 0 0 000000
ISL28022
FN8386 Rev 8.00 Page 19 of 32
February 4, 2016
BUS VOLTAGE THRESHOLD REGISTER 07h
(READ/WRITE)
The VBUS minimum and maximum threshold limits are set using
one register. The VBUS value readings range from 0V to 60V.
Table 19 on page 18 shows the register configuration and bit
weights for the VBUS threshold register. BMX bits represent the
upper limit threshold. BMN represents the lower threshold limit.
Equation 9 is the calculation used to convert the VBUS threshold
binary value to decimal. Bit is the value of each bit set in the
VBUS threshold register. The value is either 1 or a 0. The weight of
each bit is represented in Table 19. n is the bit number. The VBUS
voltage threshold LSB is 256mV.
INTERRUPT STATUS REGISTER 08h (READ/WRITE)
The interrupt status register consists of a series of bit flags that
indicate if an ADC reading has exceeded the readings respective
limit. A 1 or high reading from a warning bit indicates the reading
has exceeded the limit. To clear a warning, write a 1 or high to
the set warning bit. Table 20 on page 18 shows the definition of
the interrupt status register.
BMNW is the Bus voltage Minimum Warning. A “1” reading for
this bit indicates the bus reading is below the bus voltage
minimum threshold limit.
BMXW is the Bus voltage Maximum Warning. A “1” reading for
this bit indicates the bus reading is above the bus voltage
maximum threshold limit.
SMNW is the Shunt voltage Minimum Warning. A “1” reading for
this bit indicates the shunt reading is below the shunt voltage
minimum threshold limit.
SMXW is the Shunt voltage Maximum Warning. A “1” reading for
this bit indicates the shunt reading is above the shunt voltage
maximum threshold limit.
AUX CONTROL REGISTER 09h (READ/WRITE)
The Aux control register controls the functionality of the
EXTCLK/INT pin of the ISL28022. Table 21 on page 18 shows the
definition of the register.
FORCEINTR is the Force Interrupt bit. Programming a 1 to the bit
will force a 0 or a low at the EXTCLK/INT pin.
INTREN is the Interrupt Enable bit. Programming a 1 to the bit
will allow for a threshold measurement violation to set the state
of the EXTCLK/INT pin. With the INTREN set, any flag set from the
interrupt status register will change the state of the EXTCLK/INT
pin from 1 to a 0.
EXCLKEN is the External Clock Enable bit. Setting the bit enables
the external clock. This also changes the EXTCLK/INT pin from an
output to an input. The internal oscillator will shut down when the
bit is enabled.
EXTCLKDIV are the External Clock Divider bits. The bits control an
internal clock divider that are useful for fast system clocks. The
internal clock frequency from pin to chip is represented in
Equation 10:
fEXTCLK is the frequency of the signal driven to the EXTCLK/INT
pin. EXTCLKDIV is the decimal value of the clock divide bits.
Serial Interface
The ISL28022 supports a bidirectional bus oriented protocol. The
protocol defines any device that sends data onto the bus as a
transmitter and the receiving device as the receiver. The device
controlling the transfer is the master and the device being
controlled is the slave. The master always initiates data transfers
and provides the clock for both transmit and receive operations.
Therefore, the ISL28022 operates as a slave device in all
applications.
The ISL28022 uses two bytes to transfer all reads and writes. All
communication over the I2C interface is conducted by sending
the MSByte of each byte of data first, followed by the LSByte.
Protocol Conventions
For normal operation, data states on the SDA line can change
only during SCL LOW periods. SDA state changes during SCL
HIGH are reserved for indicating START and STOP conditions
(see Figure 27). On power-up of the ISL28022, the SDA pin is in
the input mode.
All I2C interface operations must begin with a START condition,
which is a HIGH to LOW transition of SDA while SCL is HIGH. The
ISL28022 continuously monitors the SDA and SCL lines for the
START condition and does not respond to any command until this
condition is met (see Figure 27). A START condition is ignored
during the power-up sequence.
All I2C interface operations must be terminated by a STOP
condition, which is a LOW to HIGH transition of SDA while SCL is
HIGH (see Figure 27). A STOP condition at the end of a read
operation or at the end of a write operation places the device in
its standby mode.
SMBus Support
The ISL28022 supports SMBus protocol, which is a subset of the
global I2C protocol. SMBCLK and SMBDAT have the same pin
functionality as the SCL and SDA pins, respectively. The SMBus
operates at 100kHz.
(EQ. 9)
Vb thresh
0
7
n
BitnBit_Weightn
VbThresh LSB
freq internal
fEXTCLK
EXTCLKDIV 1()2
(EQ. 10)
ISL28022
FN8386 Rev 8.00 Page 20 of 32
February 4, 2016
FIGURE 27. VALID DATA CHANGES, START AND STOP CONDITIONS
FIGURE 28. ACKNOWLEDGE RESPONSE FROM RECEIVER
FIGURE 29. BYTE WRITE SEQUENCE (SLAVE ADDRESS INDICATED BY nnnn)
SDA
SCL
START DATA DATA STOP
STABLE CHANGE
DATA
STABLE
SDA OUTPUT FROM
TRANSMITTER
SDA OUTPUT FROM
RECEIVER
81 9
START ACK
SCL FROM
MASTER
HIGH
HIGH IMPEDANCE
S
T
A
R
T
IDENTIFICATION
BYTE
DATA
BYTE
A
C
K
SIGNALS FROM THE
MASTER
SIGNALS FROM
THE ISL28022
A
C
K
1000n
WRITE
SIGNAL AT SDA nnn
ADDRESS
BYTE
S
T
O
P
DATA
BYTE
A
C
K
N
A
C
K
ISL28022
FN8386 Rev 8.00 Page 21 of 32
February 4, 2016
Device Addressing
Following a start condition, the master must output a slave
address byte. The 7 MSBs are the device identifiers. The A0 and
A1 pins control the bus address (these bits are shown in
Table 22). There are 16 possible combinations depending on the
A0/A1 connections. The last bit of the slave address byte defines
a read or write operation to be performed. When this R/W bit is a
“1”, a read operation is selected. A “0” selects a write operation
(refer to Figure 29).
After loading the entire slave address byte from the SDA bus, the
ISL28022 compares the loaded value to the internal slave
address. Upon a correct compare, the device outputs an
acknowledge on the SDA line.
Following the slave byte is a one byte word address. The word
address is either supplied by the master device or obtained from
an internal counter. On power-up, the internal address counter is
set to address 00h, so a current address read starts at address
00h. When required, as part of a random read, the master must
supply the one word address byte, as shown in Figure 30.
In a random read operation, the slave byte in the “dummy write”
portion must match the slave byte in the “read” section. For a
random read of the registers, the slave byte must be “100nnnnx”
in both places.
TABLE 22. I2C SLAVE ADDRESSES
A1 A0 SLAVE ADDRESS
GND GND 1000 000
GND VCC 1000 001
GND SDA 1000 010
GND SCL 1000 011
VCC GND 1000 100
VCC VCC 1000 101
VCC SDA 1000 110
VCC SCL 1000 111
SDA GND 1001 000
SDA VCC 1001 001
SDA SDA 1001 010
SDA SCL 1001 011
SCL GND 1001 100
SCL VCC 1001 101
SCL SDA 1001 110
SCL SCL 1001 111
Broadcast Address 0111 111
FIGURE 30. READ SEQUENCE (SLAVE ADDRESS SHOWN AS nnnn)
SIGNALS
FROM THE
MASTER
SIGNALS FROM
THE SLAVE
SIGNAL AT
SDA
S
T
A
R
T
IDENTIFICATION
BYTE WITH
R/W = 0
ADDRESS
BYTE
A
C
K
A
C
K
0
S
T
O
P
1
IDENTIFICATION
BYTE WITH
R/W = 1
A
C
K
S
T
A
R
T
SECOND READ
DATA BYTE
FIRST READ
DATA BYTE
A
C
K
100 nnnn 100nnnn
ISL28022
FN8386 Rev 8.00 Page 22 of 32
February 4, 2016
Write Operation
A write operation requires a START condition, followed by a valid
identification byte, a valid address byte, two data bytes and a
STOP condition. The first data byte contains the MSB of the data,
the second contains the LSB. After each of the four bytes, the
ISL28022 responds with an ACK. At this time, the I2C interface
enters a standby state.
Read Operation
A read operation consists of a three byte instruction, followed by
two data bytes (see Figure 30 on page 21). The master initiates
the operation issuing the following sequence: A START, the
identification byte with the R/W bit set to “0”, an address byte, a
second START and a second identification byte with the R/W bit
set to “1”. After each of the three bytes, the ISL28022 responds
with an ACK. Then the ISL28022 transmits two data bytes as
long as the master responds with an ACK during the SCL cycle
following the eighth bit of the first byte. The master terminates
the read operation (issuing no ACK then a STOP condition)
following the last bit of the second data byte (see Figure 30 on
page 21).
The data bytes are from the memory location indicated by an
internal pointer. This pointer’s initial value is determined by the
address byte in the read operation instruction and increments by
one during transmission of each pair of data bytes. The highest
valid memory location is 09h, reads of addresses higher than
that will not return useful data.
Broadcast Addressing
The DPM has a feature that allows the user to configure the settings
of all DPM chips at once. For example, a system has 16 DPM chips
connected to an I2C bus. A user can set the range or initiate a
data acquisition in one I2C data transaction by using a slave
address of 0111 111. The broadcast feature saves time in
configuring the DPM as well as measuring signal parameters in
time synchronization. The broadcast should not be used for DPM
read backs. This will cause all devices connected to the I2C bus
to talk to the master simultaneously.
I2C Clock Speed
The device supports high-speed digital transactions up to
3.4Mbs. To access the high speed I2C feature, a master byte
code of 0000 1xxx is attached to the beginning of a standard
frequency read/write I2C protocol. The x in the master byte
signifies a do not care state. X can either equal a 0 or a 1. The
master byte code should be clocked into the chip at frequencies
equal or less than 400kHz. The master code command
configures the internal filters of the ISL28022 to permit data bit
frequencies greater than 400kHz. Once the master code has
been clocked into the device, the protocol for a standard read/
write transaction is followed. The frequency at which the
standard protocol is clocked in at can be as great as 3.4MHz. A
stop bit at the end of a standard protocol will terminate the high
speed transaction mode. Appending another standard protocol
serial transaction to the data string without a stop bit, will
resume the high speed digital transaction mode. Figure 32
illustrates the data sequence for the high speed mode.
FIGURE 31. SLAVE ADDRESS, WORD ADDRESS AND DATA BYTES
D15 D14 D13 D10D12 D11 D9 D8
A0A7 A2A4 A3 A1
DATA BYTE 1
A6 A5
100 n
nnR/W
n
WORD ADDRESS
D7 D6 D5 D2D4 D3 D1 D0
SLAVE ADDRESS
BYTE
DATA BYTE 2
FIGURE 32. BYTE TRANSACTION SEQUENCE FOR INITIATING DATA RATES ABOVE 400kbs
SIGNALS
FROM THE
MASTER
SIGNALS TO THE
ISL28022
SIGNAL AT
SDA
S
T
A
R
T
MASTER
CODE
SLAVE ADDRESS
IDENTIFICATION
BYTE
A
C
K
x
S
T
O
P
ADDRESS
BYTE
A
C
K
WRITE/READ
000 01xx
S
T
A
R
T
DATA
BYTE
DATA
BYTE
A
C
K
N
A
C
K
fclk ≤ 400kHz
fclk UP TO 3.4MHz
TERMINATES HS
MODE
x100 nnnn
N
A
C
K
ISL28022
FN8386 Rev 8.00 Page 23 of 32
February 4, 2016
Signal Integrity
The purity of the signal being measured by the ISL28022 is not
always ideal. Environmental noise or noise generated from a
regulator can degrade the measurement accuracy. The
ISL28022 maintains a high CMRR ratio from DC to
approximately 10kHz, as shown in Figure 33.
The CMRR vs Frequency graph best represents the response of
the ISL28022 when an aberrant signal is applied to the circuit.
The graph was generated by shorting the ISL28022 input without
any filtering and applying a 0V to 10V triangle wave to the Shunt
inputs, VINP and VINM. The voltage shunt measurement was
recorded for each frequency applied to the shunt input.
The CMRR can be improved by designing a filter stage before the
ISL28022. The purpose of the filter stage is to attenuate the
amplitude of the unwanted signal to the noise level of the
ISL28022. Figure 34 is a simple filter example to attenuate
unwanted signals.
CSH and RSH are single pole RC filters that differentially
attenuate unwanted signals to the ISL28022. Most power
monitoring applications require a shunt resistor to be low in value
to measure large currents. For small shunt resistors, a large
value capacitor is required to attenuate low frequency signals.
Most large value capacitors are not offered in space saving
packages. The corner frequency of the differential filter, CSH and
RSH, should be designed for higher value frequency filtering.
R1 and C1 for both inputs are single ended low pass filters. The
value of the series resistor to the ISL28022 can be a larger value
than the shunt resistor, RSH. A larger series resistor to the input
allows for a lower cutoff frequency filter design to the ISL28022.
The ISL28022 can source up to 20µA of transient current in the
measurement mode. The transient or switching offset current
can be as large as 10µA. The switching offset current combined
with the series resistance, R1, creates an error offset voltage. A
balance of the value of R1 and the shunt measurement error
should be achieved for this filter design.
The common-mode voltage of the shunt input stage ranges from
0V to 60V. The capacitor voltage rating for C1 and CSH should
comply with the nominal voltage being applied to the input.
Measurement Stability vs Acquisition Time
The BADC and SADC bits within the Configuration register
configures the conversion time and accuracy for the bus and
shunt inputs, respectively. The faster the conversion time the less
accuracy and more noise introduced into the measurement.
Figure 35 is a graph that illustrates the shunt measurement
variability versus a set SADC mode. The standard deviation of
2048 shunt VOS measurements is used to quantify the
measurement variability of each mode.
Fast Transients
A small isolation resistor placed between ISL28022 inputs and
the source is recommended. In hot swap or other fast transient
events, the amplitude of a signal can exceed the recommended
operating voltage of the part due to the line inductance. The
isolation resistor creates a low pass filter between the device and
the source. The value of the isolation resistor should not be too
large. A large value isolation resistor can effect the
measurement accuracy. The offset current for shunt input can be
as large as 10µA. The value of the isolation resistor combined
with the offset current creates an error offset voltage at the
shunt input. The input of the Bus channel is connected to the top
of a precision resistor divider. The accuracy of the resistor divider
determines the gain error of the Bus channel. The input
resistance of the Bus channel is 600kΩ. Placing an isolation
resistor of the 10Ω will change the gain error of the Bus channel
by 0.0016%.
FIGURE 33. CMRR vs FREQUENCY
FIGURE 34. SIMPLIFIED FILTER DESIGN TO IMPROVE NOISE
PERFORMANCE TO THE ISL28022
80
85
90
95
100
105
110
115
120
125
130
10 100 1k 10k 100k 1M
FREQUENCY (Hz)
CMRR (dB)
LOAD
RSH
FROM
SOURCE
ISL28022
R1
R1
CSH
C1
C1
FIGURE 35. MEASUREMENT STABILITY vs SADC MODE
0
0.02
0.04
0.06
0.08
0.10
0.12
0.14
0.16
0.18
0.20
0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16
SADC MODE
VSHUNT VOS SIGMA (mV)
ISL28022
FN8386 Rev 8.00 Page 24 of 32
February 4, 2016
External Clock
An externally controlled clock allows measurements to be
synchronized to an event that is time dependent. The event could
be application generated, such as timing a current measurement
to a charging capacitor in a switch regulator application or the
event could be environmental. A voltage or current measurement
may be susceptible to crosstalk from a controlled source. Instead
of filtering the environmental noise from the measurement,
another approach would be to synchronize the measurement to
the source. The variability and accuracy of the measurement will
improve.
The ISL28022 has the functionality to allow for synchronization
to an external clock. The speed of the external clock combined
with the choice of the internal chip frequency division value
determines the acquisition times of the ADC. The internal system
clock frequency is 500kHz. The internal system clock is also the
ADC sampling clock. The acquisition times scale linearly from
500kHz. For example, an external clock frequency of 1MHz with
a frequency divide setting of 2 results in acquisition times that
equals the internal oscillator frequency when enabled. The
internal clock frequency of the ISL28022 should not exceed
500kHz. The ADC modulator is optimized for frequencies of
500kHz and below. Operating internal clock frequencies above
500kHz result in measurement accuracy errors due to the
modulator not having enough time to settle.
Suppose an external clock frequency of 1.0MHz is applied with a
divide by 8 internal frequency setting, the system clock speed is
125kHz or 4x slower than internal system clock. The acquisition
times for this example will increase by 4. For a S(B)ADC setting
of 3, the ISL28022 will have an acquisition time of 2.032ms
instead of 508µs.
The ECLK/INT pin connects to a buffer that drives a D-flip flop.
Figure 37 illustrates a simple schematic of the ECLK/INT pin
internal connection. The series of divide by 2 configured D-flip
flops are controlled by the CLKDIV bits from the Aux Control
Register. The buffer is a Schmitt triggered buffer. The bandwidth
of the buffer is 4MHz. Figure 38 shows the bandwidth of the
ECLK/INT pin.
The VSHUNT measurement error degrades at ECLK frequencies
above 4MHz. It is recommended that the ECLK does not exceed
4MHz. At ECLK frequencies below 2.5MHz or internal clock
frequencies of 208kHz, the clock frequency to modulator is too
slow allowing the charged capacitors to discharge due to
parasitic leakages. The capacitor discharge results in a
measurement error.
Over-Ranging
It is not recommended to operate the ISL28022 outside the set
voltage range. In the event of measuring a shunt voltage beyond
the maximum set range (320mV) and lower than the clamp
voltage of the protection diode (1V), the measured output
reading may be within the accepted range but will be incorrect.
Shunt Resistor Selection
In choosing a sense resistor, the following resistor parameters
need to be considered: the resistor value, resistor temperature
coefficient and resistor power rating.
The sense resistor value is a function of the full-scale voltage
drop across the shunt resistor and the maximum current
measured for the application. The ISL28022 has 4 voltage
ranges that are controlled by programming the PGA bits within
the configuration register. The PGA bits control the voltage range
for the VSHUNT input (VINP-VINM) of the ISL28022. Once the
voltage range for the input is chosen and the maximum
measurable current is known, the sense resistor value is
calculated using Equation 11:
In choosing a sense resistor, the sense resistor power rating
should be taken into consideration. The physical size of a sense
resistor is proportional to the power rating of the resistor. The
maximum power rating for the measurement system is
calculated as the Vshunt_range multiplied by the maximum
FIGURE 36. SIMPLIFIED SCHEMATIC OF THE ISL28022
SYNCHRONIZED TO A PWM SOURCE
FIGURE 37. SIMPLIFIED INTERNAL BLOCK CONNECTION OF THE
ECLK/INT PIN
I2C
SMBUS
A1
SCL
SDA
VINP
VINM
GND
RSH
ADC
16-BIT
SW MUX
ECLK
REG
MAP
VCC
VBUS
A0
ISL28022 DPM
LOAD
TO MCU
FUNCTION
GENERATO R
3.3V
+
-
VTH
ECLK/INT
+1
÷
Fclk_Sys
2X
FIGURE 38. EXTERNAL CLOCK BANDWIDTH vs MEASUREMENT
ACCURACY
-1
0
1
2
3
4
5
1 10
EXTERNAL CLOCK FREQUENCY (MHz)
VSHUNT MEASUREMENT
ERROR NORMALIZED (%)
CLKDIV = 5 (÷ 12)
SADC = 3
(EQ. 11)
Rsense
Vshunt_range
Imeas Max
ISL28022
FN8386 Rev 8.00 Page 25 of 32
February 4, 2016
measurable current expected. The power rating equation is
represented by Equation 12:
A general rule of thumb is to multiply the power rating calculated
in Equation 12 by 2. This allows the sense resistor to survive an
event when the current passing through the shunt resistor is
greater than the measurable maximum current. The higher the
ratio between the power rating of the chosen sense resistor and
the calculated power rating of the system (Equation 12), the less
the resistor will heat up in high-current applications.
The Temperature Coefficient (TC) of the sense resistor directly
degrades the current measurement accuracy. The surrounding
temperature of the sense resistor and the power dissipated by
the resistor will cause the sense resistor value to change. The
change in resistor temperature with respect to the amount of
current that flows through the resistor, is directly proportional to
the ratio of the power rating of the resistor versus the power
being dissipated. A change in sense resistor temperature results
in a change in sense resistor value. Overall, the change in sense
resistor value contributes to the measurement accuracy for the
system. The change in a resistor value due to a temperature rise
can be calculated using Equation 13:
Temperature is the change in temperature in Celsius. RsenseTC
is the temperature coefficient rating for a sense resistor. Rsense
is the resistance value of the sense resistor at the initial
temperature.
Table 23 is a shunt resistor reference table for select full-scale
current measurement ranges (ImeasMax). The table also
provides the minimum rating for each shunt resistor.
It is often hard to readily purchase shunt resistor values for a desired
measurable current range. Either the value of the shunt resistor
does not exist or the power rating of the shunt resistor is too low. A
means of circumventing the problem is to use two or more shunt
resistors in parallel to set the desired current measurement range.
For example, an application requires a full-scale current of 50A with
a maximum voltage drop across the shunt resistor of 40mV.
Table 23 shows this requires a sense resistor of 0.8mΩ, 2W resistor.
Assume the power ratings and the shunt resistor values to choose
from are 1mΩ 1W, 2mΩ/1W, and 4mΩ/1W.
Let’s use a 1mΩ and a 4mΩ resistor in parallel to create the
shunt resistor value of 0.8mΩ. Figure 39 shows an illustration of
the shunt resistors in parallel.
The power to each shunt resistor should be calculated before
calling a solution complete. The power to each shunt resistor is
calculated using Equation 14:
TABLE 23. SHUNT RESISTOR VALUES AND POWER RATINGS FOR
SELECT MEASURABLE CURRENT RANGES
Rsense/
Prating VSHUNT RANGE (PGA SETTING)
ImeasMax
(PGA 00)
40mV
(PGA 01)
80mV
(PGA 10)
160mV
(PGA 11)
320mV
100µA 400Ω/4µW 800Ω/8µW 1.6kΩ/16µW 3.2kΩ/
32µW
1mA 40Ω/40µW 80Ω/80µW 160Ω/160µW 320Ω/
320µW
10mA 4Ω/400µW 8Ω/800µW 16Ω/1.6mW 32Ω/
3.2mW
100mA 400mΩ/4mW 800mΩ/8mW 1.6Ω/16mW 3.2Ω/
30mW
500mA 80mΩ/20mW 160mΩ/40mW 320mΩ/
80mW
640mΩ/
160mW
1A 40mΩ/40mW 80mΩ/80mW 160mΩ/
160mW
320mΩ/
320mW
5A 8mΩ/200mW 16mΩ/400mW 32mΩ/
800mW
64mΩ/
1.6W
(EQ. 12)
Pres_rating Vshunt_range Imeas Max
(EQ. 13)
Rsense Rsense Rsense TC
Temperature
10A 4mΩ/400mW 8mΩ/800mW 16mΩ/1.6W 32mΩ/
3.2W
50A 0.8mΩ/2W 1.6mΩ/4W 3.2mΩ/8W 6.4mΩ/
16W
100A 0.4mΩ/4W 0.8mΩ/8W 1.6mΩ/16W 3.2mΩ/
32W
500A 0.08mΩ/20W 0.16mΩ/40W 0.32mΩ/80W 0.64mΩ/
160W
TABLE 23. SHUNT RESISTOR VALUES AND POWER RATINGS FOR
SELECT MEASURABLE CURRENT RANGES (Continued)
Rsense/
Prating VSHUNT RANGE (PGA SETTING)
ImeasMax
(PGA 00)
40mV
(PGA 01)
80mV
(PGA 10)
160mV
(PGA 11)
320mV
0.001
0.004
FIGURE 39. A SIMPLIFIED SCHEMATIC ILLUSTRATING THE USE OF
TWO SHUNT RESISTORS TO CREATE A DESIRED SHUNT
VALUE
(EQ. 14)
PshuntRes
Vshunt_range
2
Rsense
ISL28022
FN8386 Rev 8.00 Page 26 of 32
February 4, 2016
The power dissipated by the 1mΩ resistor is 1.6W. 400mW is
dissipated by the 4mΩ resistor. 1.6W exceeds the rating limit of
1W for the 1mΩ sense resistor. Another approach would be to
use three shunt resistors in parallel as illustrated in Figure 40.
Using Equation 14 on page 25, the power dissipated to each
shunt resistor yields 0.8W for the 2mΩ shunt resistors and 0.4W
for the 4mΩ shunt resistor. All shunt resistors are within the
specified power ratings.
Lossless Current Sensing (DCR)
A DCR sense circuit is an alternative to a sense resistor. The DCR
circuit utilizes the parasitic resistance of an inductor to measure
the current to the load. A DCR circuit remotely measures the
current through an inductor. The lack of components in series
with the regulator to the load makes the circuit lossless.
A properly matched DCR circuit has an equivalent circuit seen by
the ADC equals to Rdcr in Figure 41. Before deriving the transfer
function between the inductor current and voltage seen by the
ISL28022, let’s review the definition of an inductor and capacitor
in the Laplacian domain.
Xc is the impedance of a capacitor related to the frequency and
XL is the impedance of an inductor related to frequency. ω equals
to 2*π*f. f is the chop frequency dictated by the regulator. Using
Ohms law, the voltage across the DCR circuit in terms of the
current flowing through the inductor is defined in Equation 16.
In Equation 16, Rdcr is the parasitic resistance of the inductor.
The voltage drop across the inductor (Lo) and the resistor (Rdcr)
circuit is the same as the voltage drop across the resistor (Rsen)
and the capacitor (Csen) circuit. Equation 17 defines the voltage
across the capacitor (Vcsen) in terms of the inductor current (IL).
The relationship between the inductor load current (IL) and the
voltage across capacitor simplifies if the following component
selection holds true:
If Equation 18 hold true, the numerator and denominator of the
fraction in Equation 17 cancels reducing the voltage across the
capacitor to the equation represented in Equation 19.
Most inductor datasheets will specify the average value of the
Rdcr for the inductor. Rdcr values are usually sub 1mΩ with a
tolerance averaging 8%. Common chip capacitor tolerances
average to 10%.
Inductors are constructed out of metal. Metal has a high
temperature coefficient. The temperature drift of the inductor
value could cause the DCR circuit to be untuned. An untuned
circuit results in inaccurate current measurements along with a
chop signal bleeding into the measurement. To counter the
temperature variance, a temperature sensor may be
incorporated into the design to track the change in component
values.
A DCR circuit is good for gross current measurements. As
discussed, inductors and capacitors have high tolerances and are
temperature dependent which will result in less than accurate
current measurements.
In Figure 41, there is a resistor in series with the ISL28022
negative shunt terminal, VINM, with the value of Rsen + Rdcr. The
resistor’s purpose is to counter the effects of the bias current
from creating a voltage offset at the input of the ADC.
Layout
The layout of a current measuring system is equally important as
choosing the correct sense resistor and the correct analog
converter. Poor layout techniques can result in severed traces,
signal path oscillations, magnetic contamination, which all
contribute to poor system performance.
TRACE WIDTH
Matching the current carrying density of a copper trace with the
maximum current that will pass through is critical in the
performance of the system. Neglecting the current carrying
capability of a trace will result in a large temperature rise in the
trace and the loss in system efficiency due to the increase in
resistance of the copper trace. In extreme cases, the copper
FIGURE 41. A SIMPLIFIED CIRCUIT EXAMPLE OF A DCR
0.002
0.004
0.002
FIGURE 40. INCREASING THE NUMBER OF SHUNT RESISTORS IN
PARALLEL TO CREATE A SHUNT RESISTOR VALUE
REDUCES THE POWER DISSIPATED BY EACH SHUNT
RESISTOR
VINP
VINM
Lo Rdcr
Cse n
Rse n
DCR CIRCUIT
ADC
16-BIT
FB
PHASE
Rse n + Rdcr
BUCK
REGULA TOR
LOAD
(EQ. 15)
Xcf() 1
jf()CXLf() jf()L
(EQ. 16)
Vdcr f() R
dcr jf()L
iL
(EQ. 17)
VcF()
jf()LRdcr
1jf()Csen
Rsen
IL
Rdcr
1jwf()L()
Rdcr
1jf()Csen
Rsen
IL
(EQ. 18)
L
Rdcr
Csen Rsen
(EQ. 19)
VcRdcr iL
ISL28022
FN8386 Rev 8.00 Page 27 of 32
February 4, 2016
trace could be severed because the trace could not pass the
current. The current carrying capability of a trace is calculated
using Equation 20:
Imax is the largest current expected to pass through the trace. T
is the allowable temperature rise in Celsius when the maximum
current passes through the trace. TraceThickness is the thickness
of the trace specified to the PCB fabricator in mils. A typical
thickness for general current carrying applications (<100mA) is
0.5oz copper or 0.7mils. For larger currents, the trace thickness
should be greater than 1.0oz or 1.4mils. A balance between
thickness, width and cost needs to be achieved for each design.
The coefficient k in Equation 20 changes depending on the trace
location. For external traces, the value of k equals 0.048 while
for internal traces the value of k reduces to 0.024. The k values
and Equation 20 are stated per the ANSI IPC-2221(A) standards.
TRACE ROUTING
It is always advised to make the distance between voltage
source, sense resistor and load as close as possible. The longer
the trace length between components will result in voltage drops
between components. The additional resistance will reduce the
efficiency of a system.
The bulk resistance, , of copper is 0.67µΩ/in or 1.7µΩ/cm at
+25°C. The resistance of trace can be calculated from Equation 21:
Figure 42 illustrates each dimension of a trace.
For example, assume a trace has 2oz of copper or 2.8mil
thickness, a width of 100mil and a length of 0.5in. Using
Equation 21, the resistance of the trace is approximately 2mΩ.
Assume 1A of current is passing through the trace. A 2mV
voltage drop would result from trace routing.
Current flowing through a conductor will take the path of least
resistance. When routing a trace, avoid orthogonal connections
for current bearing traces.
Orthogonal routing for high current flow traces will result in
current crowding, localized heating of the trace and a change in
trace resistance (see Figure 43).
The utilization of arcs and 45° traces in routing large current flow
traces will maintain uniform current flow throughout the trace.
Figure 44 illustrates the routing technique.
CONNECTING SENSE TRACES TO THE CURRENT SENSE
RESISTOR
Ideally, a 4 terminal current sense resistor would be used as the
sensing element. Four terminal sensor resistors can be hard to
find in specific values and in sizes. Often a two terminal sense
resistor is designed into the application.
Sense lines are high impedance by definition. The connection
point of a high impedance line reflects the voltage at the
intersection of a current bearing trace and a high impedance
trace. The high impedance trace should connect at the
intersection where the sense resistor meets the landing pad on
the PCB. The best place to make current sense line connection is
on the inner side of the sense resistor footprint. The illustration of
the connection is shown in Figure 45 on page 28. Most of the
current flow is at the outer edge of the footprint. The current
ceases at the point the sense resistor connects to the landing
pad. Assume the sense resistor connects at the middle of the
each landing pad, this leaves the inner half of the each landing
pad with little current flow. With little current flow, the inner half
of each landing pad is classified as high impedance and perfect
for a sense connection.
(EQ. 20)
Trace width
Imax
kT0.44
1
0.725
Trace Thickness
(EQ. 21)
Rtrace Trace length
Trace width Trace thickness
TRACE
WIDTH
TRACE
THICKNESS
FIGURE 42. ILLUSTRATION OF THE TRACE DIMENSIONS FOR A STRIP
LINE TRACE
CURRENT FLOW
FIGURE 43. AVOID ROUTING ORTHOGONAL CONNECTIONS FOR
TRACES THAT HAVE HIGH CURRENT FLOWS
CURRENT FLOW
CURRENT FLOW
FIGURE 44. USE ARCS AND 45 DEGREE TRACES TO SAFELY ROUTE
TRACES WITH LARGE CURRENT FLOWS
ISL28022
FN8386 Rev 8.00 Page 28 of 32
February 4, 2016
Current sense resistors are often smaller than the width of the
traces that connect to the footprint. The trace connecting to the
footprint is tapered at a 45° angle to control the uniformity of the
current flow.
MAGNETIC INTERFERENCE
The magnetic field generated from a trace is directly proportional
to the current passing through the trace and the distance from
the trace the field is being measured at. Figure 46 illustrates the
direction the magnetic field flows versus current flow.
The equation in Figure 46 determines the magnetic field, B, the
trace generates in relation to the current passing through the
trace, I, and the distance the magnetic field is being measured
from the conductor, r. The permeability of air, µo, is
4*10-7 H/m.
When routing high-current traces, avoid routing high impedance
traces in parallel with high-current bearing traces. A means of
limiting the magnetic interference from high-current traces is to
closely route the paths connected to and from the sense resistor.
The magnetic fields will cancel outside the two traces and add
between the two traces. Figure 47 illustrates a layout that is less
sensitive to magnetic field interference.
If possible, do not cross traces with high-current. If a trace
crossing cannot be avoided, cross the trace in an orthogonal
manor and the furthest layer from the current bearing trace. The
interference from the current bearing trace will be limited.
A Trace as a Sense Resistor
In previous sections, the resistance and the current carrying
capabilities of a trace were discussed. In high current sense
applications, a design may utilize the resistivity of a current
sense trace as the sense resistor. This section will discuss how to
design a sense resistor from a copper trace.
Suppose an application needs to measure current up to 200A.
The design requires the least amount of voltage drop for
maximum efficiency. The full-scale voltage range of 40mV
(PGA 00) is chosen. From Ohms law, the sense resistor is
calculated to be 200µΩ. The power rating of the resistor is
calculated to be 8W. Assume the PCB trace thickness of the
board equals 2oz/2.8mils and the maximum temperature rise of
the trace is 20°C. Using Equation 20 on page 27, the calculated
trace width is 2.192in. The trace width, thickness and the desired
sense resistor value is known. Utilizing Equation 21 on page 27,
the trace length is calculated to be 1.832in.
SENSE RESISTOR
SENSE TRACE
SENSE TRACE
LANDING PAD
LANDING PAD
CURRENT BEARING TRACE
CURRENT BEARING TRACE
FIGURE 45. CONNECTING THE SENSE LINES TO A CURRENT SENSE
RESISTOR
B
oI
2 r
FIGURE 46. THE CONDUCTOR ON THE LEFT SHOWS THE MAGNETIC
FIELD FLOWING IN A CLOCKWISE DIRECTION FOR
CURRENTS FLOWING INTO THE PAGE. A CURRENT
FLOW OUT OF THE PAGE HAS A COUNTER CLOCKWISE
MAGNETIC FLOW
FIGURE 47. CLOSELY ROUTED TRACES THAT CONNECT TO THE
SENSE RESISTOR REDUCES THE MAGNETIC
INTERFERENCE SOURCED FROM THE CURRENT
FLOWING THROUGH THE TRACES
SENSE
RESISTOR
SENSE
TRACE
SENSE
TRACE
LANDING
PAD
LANDING
PAD
CURRENT FLOW
CURRENT FLOW
TO THE SENSE
RESISTOR
FROM THE SENSE
RESISTOR
Bto Bfrom
Bto Bfrom
Bto Bfrom
TO SENSE CIRCUITRY
FN8386 Rev 8.00 Page 29 of 32
February 4, 2016
ISL28022
Intersil products are manufactured, assembled and tested utilizing ISO9001 quality systems as noted
in the quality certifications found at www.intersil.com/en/support/qualandreliability.html
Intersil products are sold by description only. Intersil may modify the circuit design and/or specifications of products at any time without notice, provided that such
modification does not, in Intersil's sole judgment, affect the form, fit or function of the product. Accordingly, the reader is cautioned to verify that datasheets are
current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its
subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
For additional products, see www.intersil.com/en/products.html
© Copyright Intersil Americas LLC 2013-2016. All Rights Reserved.
All trademarks and registered trademarks are the property of their respective owners.
Figure 48 illustrates a layout example of a current sense resistor
defined by a PCB trace. The serpentine pattern of the resistor
reduces current crowding as well as limiting the magnetic
interference caused by the current flowing through the trace.
The width of the trace in Figure 48 illustration would equal
2.192in and the length between the sense lines equals 1.832in.
The width of the resistor is long for some applications. A means
of shortening the trace width is to connect two traces in parallel.
For calculation ease, assume the resistive traces are routed on
the outside layers of a PCB. Using Equations 20 and 21 on
page 27, the width of the trace is reduced from 2.192in to
1.096in.
When using multiple layers to create a trace resistor, use
multiple vias to keep the trace potentials between the two
conductors the same. Vias are highly resistive compared to a
copper trace. Multiple vias should be employed to lower the
voltage drop due to current flowing through resistive vias.
Figure 49 illustrates a layout technique for a multiple layered
trace sense resistor.
FIGURE 48. ILLUSTRATES A LAYOUT EXAMPLE OF A CURRENT
SENSE RESISTOR MADE FROM A PCB TRACE
CURRENT FLOW IN
CURRENT FLOW OUT
SENSE NEG(-)
SENSE POS(+)
THE LENGTH OF THE
TRACE BETWEEN
THE TWO SENSE
LINES DEFINES THE
SENSE RESISTOR
VALUE.
FIGURE 49. ILLUSTRATES A LAYOUT EXAMPLE OF A MULTIPLE
LAYER TRACE RESISTOR
TOP
BOTTOM
TRACE
TRACE
VIA
VIA
TRACE
VIA
(A) CROSS SECTION VIEW (B) TOPVIEW
PCB
PCB
ISL28022
FN8386 Rev 8.00 Page 30 of 32
February 4, 2016
About Intersil
Intersil Corporation is a leading provider of innovative power management and precision analog solutions. The company's products
address some of the largest markets within the industrial and infrastructure, mobile computing and high-end consumer markets.
For the most updated datasheet, application notes, related documentation and related parts, please see the respective product
information page found at www.intersil.com.
You may report errors or suggestions for improving this datasheet by visiting www.intersil.com/ask.
Reliability reports are also available from our website at www.intersil.com/support
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to the web to make sure that
you have the latest revision.
DATE REVISION CHANGE
February 4, 2016 FN8386.8 Changed the Polarity of the CNVR bit from High to Low when the ADC is making a conversion. See section
“CNVR: Conversion Ready (Bit 1)” on page 17
Changed in “Write Operation” on page 22 the description of the first byte of data sent to the ISL28022 from
LSB to MSB. The second byte of data was changed from MSB to LSB.
In Figure 32 on page 22, added an ACK to the diagram before the repeat start.
October 2, 2015 FN8386.7 Changed in Table 15 on page 16, Bit D0 from “FS0” to “0”.
Added statement in number 3 (second to the last sentence of “Calibration Register 05h (Read/Write)” on
page 17, which reads “The calibration value...”
June 17, 2015 FN8386.6 Added Related Literature section on page 1.
Added DPM Portfolio Comparison table on page 5.
Removed Typical Applications section and made into an application note (AN1955).
February 20, 2015 FN8386.5 Electrical Specifications table on page 7- DC accuracy under Test Conditions: Updated VSENSE from ±300mV
to ±320mV.
Table 5 on page 14: Changed in range (mV) column from ±300 to ±320.
Table 8 on page 16 updated “FULL-SCALE in title from ±300mV to ±320mV.
Equation 1 on page 17: Changed n=0 to n=2.
Calibration Register 05h (Read/Write) section on page 17 above equation 2, changed range: (300mV,
160mV, 80mV and 40mV) to (320mV, 160mV, 80mV and 40mV).
Over-ranging section on page 24: Updated maximum set range from (300mV) to (320mV).
Table 22 on page 24 changed PG11 from 300mV to 320mV.
Point Of Load Power Monitor section on page 29: Changed shunt voltage from 300mV to 320mV.
Equation 22 on page 28: Changed value from 0.30 to 0.32.
page 31 updated Vshunt range from 300mV to 320mV.
June 9, 2014 FN8386.4 Equation 17 on page 26 added IL before = Rdcr
Figure 42 on page 27 changed “Of a Strip” to “For a Strip”
Figure 47 on page 28 changed “Current flow” to “A current flow”
Last sentence in paragraph following Figure 46 on page 28 and second sentence in paragraph under
Equation 28 on page 32 changed “107” to “10-7”
April 17, 2014 FN8386.3 Text revisions done in section “Signal Integrity” on page 23.
Added section “Lossless Current Sensing (DCR)” on page 26 and “Monitoring MultiCell Battery Levels Using
the ISL28022 Broadcast Command” on page 36.
Updated the Ordering Information on page 3 by removing R-spec parts.
October 10, 2013 FN8386.2 Added sections from “Shunt Resistor Selection” on page 24 to “An Efficiency Measurement Using the
ISL28022 Broadcast Feature” on page 35.
April 26, 2013 FN8386.1 Added R-spec parts to ordering information and updated verbiage in About Intersil.
April 16, 2013 FN8386.0 Initial Release
ISL28022
FN8386 Rev 8.00 Page 31 of 32
February 4, 2016
Package Outline Drawing
M10.118
10 LEAD MINI SMALL OUTLINE PLASTIC PACKAGE
Rev 1, 4/12
DETAIL "X"
SIDE VIEW 2
TYPICAL RECOMMENDED LAND PATTERN
TOP VIEW
PIN# 1 ID
0.18 - 0.27
DETAIL "X"
0.10 ± 0.05
(4.40)
(3.00)
(5.80)
H
C
1.10 MAX
0.09 - 0.20
3°±3°
GAUGE
PLANE 0.25
0.95 REF
0.55 ± 0.15
B
0.08 C A-B D
3.0±0.05
12
10
0.85±010
SEATING PLANE
A
0.50 BSC
3.0±0.05 4.9±0.15
(0.29)
(1.40)
(0.50)
D
5
5
SIDE VIEW 1
Dimensioning and tolerancing conform to JEDEC MO-187-BA
Plastic interlead protrusions of 0.15mm max per side are not
Dimensions in ( ) are for reference only.
Dimensions are measured at Datum Plane "H".
Plastic or metal protrusions of 0.15mm max per side are not
Dimensions are in millimeters.
3.
4.
5.
6.
NOTES:
1.
2.
and AMSEY14.5m-1994.
included.
included.
0.10 C
M
ISL28022
FN8386 Rev 8.00 Page 32 of 32
February 4, 2016
Package Outline Drawing
L16.3x3B
16 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 1, 4/07
located within the zone indicated. The pin #1 indentifier may be
Unless otherwise specified, tolerance : Decimal ± 0.05
Tiebar shown (if present) is a non-functional feature.
The configuration of the pin #1 identifier is optional, but must be
between 0.15mm and 0.30mm from the terminal tip.
Dimension b applies to the metallized terminal and is measured
Dimensions in ( ) for Reference Only.
Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
6.
either a mold or mark feature.
3.
5.
4.
2.
Dimensions are in millimeters.1.
NOTES:
BOTTOM VIEW
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
TOP VIEW
BOTTOM VIEW
SIDE VIEW
9
( 2. 80 TYP )
( 1. 70 )
(4X) 0.15
( 12X 0 . 5 )
( 16X 0 . 60)
( 16X 0 . 23 )
0 . 90 ± 0.1
INDEX AREA
PIN 1
6
A
3.00
B
3.00
12
4
4
5
8
16X 0.40 ± 0.10
5
0 . 2 REF
C
0 . 00 MIN.
0 . 05 MAX.
BCMA0.10
C
- 0.05
BASE PLANE
0.10 C
SEE DETAIL "X"
C
0.08
SEATING PLANE
+ 0.07
16X 0.23
16
13
12X
1.5
4X
0.50
1
6
PIN #1 INDEX AREA
1 .70
+ 0.10
- 0.15
