LM34923MMX/NOPB - NATIONAL SEMICONDUCTOR

LM34923

LM34923 80V, 600 mA Constant On-Time Buck Switching Regulator

Literature Number: SNVS695

LM34923

March 1, 2011

80V, 600 mA Constant On-Time Buck Switching Regulator

General Description

The LM34923 Step Down Switching Regulator features all of

the functions needed to implement a low cost, efficient Buck

bias regulator. This high voltage regulator contains an 80V N-

Channel MOSFET Switch and a startup regulator. The device

is easy to implement and is provided in an MSOP-10 package.

The regulator’s control scheme uses an on-time inversely

proportional to VIN. This feature results in the operating fre-

quency remaining relatively constant with line and load vari-

ations. The control scheme requires no loop compensation,

resulting in fast transient response. An intelligent current limit

is implemented with a forced off-time which is inversely pro-

portional to VOUT. This scheme ensures short circuit control

while providing minimum foldback. Other features include:

Thermal Shutdown, VCC Under Voltage Lock-out, Max Duty

Cycle Limiter, a Pre-charge Switch, and a programmable Un-

der Voltage Detector with a status flag output.

Features

■Operating input voltage range: 6V to 75V

■Integrated 80V, N-Channel Buck Switch

■Internal start-up regulator

■No Loop compensation required

■Ultra-Fast Transient Response

■Operating frequency remains constant with line and load

variations

■Adjustable output voltage from 2.5V

■Precision internal reference, ±2.5%

■Intelligent current limit reduces foldback

■Programmable Input UV detector with status flag output

■Pre-charge switch enables bootstrap gate drive with no

load

■Thermal shutdown

Typical Applications

■Non-Isolated Telecommunication Buck Regulator

■Secondary High Voltage Post Regulator

■+42V Automotive Systems

Package

■MSOP - 10

Typical Application, Basic Step-Down Regulator

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LM34923 80V, 600 mA Constant On-Time Buck Switching Regulator

Connection Diagram

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Top View

10–Lead MSOP

Ordering Information

Order Number Package Type NSC Package Drawing Supplied As

LM34923MM MSOP-10 MUB10A 1000 Units on Tape and Reel

LM34923MMX 3500 Units on Tape and Reel

Pin Descriptions

Pin Name Description Application Information

1 SW Switching Node Power switching node. Connect to the output inductor, re-circulating diode or

synchronous FET, and bootstrap capacitor.

2 BST Boost Pin An external capacitor is required between the BST and the SW pins (0.01uF

or greater ceramic). The BST pin capacitor is charged from Vcc through an

internal diode when SW is low.

3 N/C Do not connect

4 RTN Ground pin Ground for the entire circuit.

5 UV Input pin for the under voltage

indicator

A resistor divider from VIN, or some other system voltage, programs the under-

voltage detection threshold. An internal current sink is enabled when UV is

below 2.5V to provide hysteresis.

6 UVO Under voltage status indicator This open drain output is high when the UV pin voltage is below 2.5V, or when

the VCCUVLO function or the shutdown function is invoked.

7 FB Feedback Input from Regulated

Output

This pin is connected to the inverting input of the internal regulation

comparator. The regulation level is 2.5V.

8 RT/SD On-time set pin and shutdown input A resistor between this pin and Vin sets the switch on-time as a function of Vin,

and the frequency. The minimum recommended on-time is 200 ns at max input

voltage. Taking this pin to ground shuts off the regulator.

9 VCC Output from the internal high voltage

series pass regulator. Regulated at

7.5V.

The internal regulator provides bias supply for the Buck switch gate driver and

other internal circuitry. A 1uF ceramic capacitor to ground is required. The

regulator is current limited to ≈30 mA.

10 VIN Input Voltage The operating input range is 6V to 75V

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LM34923

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

VIN, UV to RTN -0.3V to 80V

BST to RTN -0.3V to 88V

SW to RTN (Steady State) -1V to VIN + 0.3V

BST to VCC 80V

BST to SW 10V

VCC, UVO, RT to RTN –0.3V to 10V

FB, RT, to RTN -0.3 to 5V

ESD Rating (Note 5)

Human Body Model 2kV

For soldering specs see:

www.national.com/ms/MS/MS-SOLDERING.pdf

Junction Temperature 150°C

Storage Temperature Range -55°C to +150°C

Operating Ratings (Note 1)

VIN 6V to 75V

Operating Junction Temperature −40°C to + 125°C

Electrical Characteristics Specifications with standard type are for TJ = 25°C only; limits in boldface type apply

over the full Operating Junction Temperature (TJ) range. Minimum and Maximum limits are guaranteed through test, design, or

statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference

purposes only. Unless otherwise stated the following conditions apply: VIN = 48V(Note 3).

Symbol Parameter Conditions Min Typ Max Units

VCC Supply

Vcc Reg Vcc Regulator Output Vin = 48V 7.1 7.5 7.9 V

Vin – Vcc VIN = 6V, ICC = 5mA 240 mV

Vcc Output Impedance Vin =6V 45 Ω

Vcc Current Limit Vin = 48V (Note 4) 20 30 mA

Vcc UVLO Vcc Increasing 4 4.8 V

Vcc UVLO hysteresis 450 mV

Iin Operating current FB = 3V, Vin = 48V 1 1.32 mA

Iin Shutdown Current RT/SD = 0V 20 70 µA

Switch Characteristics

Buckswitch Rds(on) Itest = 200 mA 0.56 1.1 Ω

Gate Drive UVLO Vbst – Vsw Rising 2.15 33.8 V

Gate Drive UVLO hysteresis 250 mV

Pre-charge switch voltage At 1 mA 0.8 V

Pre-charge switch on-time 150 ns

Current Limit

Current Limit Threshold 700 1175 1500 mA

Current Limit Response Time Iswitch = 1.24A, Time to

Switch Off

190 ns

TOFF-1 OFF time generator (test 1) FB=0V, VIN = 75V 37 µs

TOFF-2 OFF time generator (test 2) FB=2.3V, VIN = 75V 7.2 µs

TOFF-3 OFF time generator (test 3) FB=0V, VIN = 10V 5.7 µs

TOFF-4 OFF time generator (test 4) FB=2.3V, VIN = 10V 1.25 µs

On Time Generator

TON - 1 On-Time Vin = 10V

Ron = 250K

2.2 3.3 4.51 µs

TON - 2 On-Time Vin = 75V

Ron = 250K

300 450 565 ns

Remote Shutdown Threshold Voltage at RT/SD rising 0.46 0.9 1.4 V

Remote Shutdown Hysteresis 60 mV

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LM34923

Symbol Parameter Conditions Min Typ Max Units

Minimum Off Time

Minimum Off Time VIN = 6V 260 347 ns

Regulation and OV Comparators

FB Reference Threshold Internal reference

Trip point for switch ON

2.4365 2.5 2.5625 V

FB Over-Voltage Threshold Trip point for switch OFF 2.85 V

FB Bias Current 1 nA

Under Voltage Sensing

UVTH UV Threshold 2.4 2.5 2.6 V

UVHYS UV Hysteresis Current UV = 2V 2.7 57.3 uA

UVBIAS UV Bias Current UV = 3V 1 nA

UVOVOL UVO Output Low Voltage UV = 3V, IUVO = 5mA 360 600 mV

UVOIOH UVO Leakage Current UV = 2V, VUVO = 7.8V 1 nA

Thermal Shutdown

Tsd Thermal Shutdown Temp. 165 °C

Thermal Shutdown Hysteresis 20 °C

Thermal Resistance

θJA Junction to Ambient MUA Package 200 °C/W

Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the

device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.

Note 2: For detailed information on soldering plastic MSOP packages, refer to the Packaging Data Book available from National Semiconductor Corporation.

Note 3: All limits are guaranteed. All electrical characteristics having room temperature limits are tested during production with TA = TJ = 25°C. All hot and cold

limits are guaranteed by correlating the electrical characteristics to process and temperature variations and applying statistical process control.

Note 4: The VCC output is intended as a self bias for the internal gate drive power and control circuits. Device thermal limitations limit external loading.

Note 5: The human body model is a 100pF capacitor discharged through a 1.5kΩ resistor into each pin.

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LM34923

Typical Performance Characteristics

Efficiency at 300 kHz, 10V

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Efficiency Comparison at 200 kHz

0 100 200 300 400 500 600

100

EFFICIENCY (%)

LOAD CURRENT (mA)

VOUT=5V, D1=DFLS1100

7.5V, D1

24V, D1

6V, D1

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VCC vs. VIN

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VCC vs. ICC

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ICC vs. Externally Applied VCC

7.5 8.0 8.5 9.0 9.5 10.0

ICC (mA)

APPLIED VCC (V)

900 kHz, D1

200 kHz, D1

DCM

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On-Time vs. VIN and RT

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LM34923

Current Limit Off-Time vs. VFB

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Maximum Switching Frequency

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Voltage at the RT Pin

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Operating Current into VIN

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Shutdown Current into VIN

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UVO Pin Low Voltage vs. Sink Current

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LM34923

VCC UVLO vs. Temperature

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Gate Drive UVLO vs. Temperature

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VCC vs. Temperature

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VCC Dropout vs. Temperature

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VCC Output Impedance vs. Temperature

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VCC Current Limit vs. Temperature

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LM34923

Reference Voltage vs. Temperature

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On-time vs. Temperature

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Minimum Off-time vs. Temperature

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Current Limit Threshold vs. Temperature

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Current Limit Off-Time vs. Temperature

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Operating Current vs. Temperature

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LM34923

Shutdown Current vs. Temperature

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RT Pin Shutdown Threshold vs. Temperature

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UV Pin Threshold vs. Temperature

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UV Hysteresis Current vs. Temperature

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LM34923

Block Diagram

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Functional Description

The LM34923 Step Down Switching Regulator features all the

functions needed to implement a low cost, efficient, Buck bias

power converter. This high voltage regulator contains an 80

V N-Channel Buck Switch, is easy to implement and is pro-

vided in the MSOP-10 package. The regulator is based on a

control scheme using an on-time inversely proportional to

VIN. The control scheme requires no loop compensation. Cur-

rent limit is implemented with forced off-time, which is in-

versely proportional to VOUT. This scheme ensures short

circuit control while providing minimum foldback.

The LM34923 can be applied in numerous applications to ef-

ficiently regulate down higher voltages. This regulator is well

suited for 48 Volt Telecom and the new 42V Automotive pow-

er bus ranges. Features include: Thermal Shutdown, VCC

under-voltage lockout, Gate drive under-voltage lockout, Max

Duty Cycle limit timer, intelligent current limit off timer, a pre-

charge switch, and a programmable under voltage detector

with status flag.

Control Circuit Overview

The LM34923 is a Buck DC-DC regulator that uses a control

scheme in which the on-time varies inversely with line voltage

(VIN). Control is based on a comparator and the on-time one-

shot, with the output voltage feedback (FB) compared to an

internal reference (2.5V). If the FB level is below the reference

the buck switch is turned on for a fixed time determined by the

line voltage and a programming resistor (RT). Following the

ON period the switch remains off for at least the minimum off-

timer period of 260 ns. If FB is still below the reference at that

time the switch turns on again for another on-time period. This

continues until regulation is achieved.

The LM34923 operates in discontinuous conduction mode at

light load currents, and continuous conduction mode at heavy

load current. In discontinuous conduction mode, current

through the output inductor starts at zero and ramps up to a

peak during the on-time, then ramps back to zero before the

end of the off-time. The next on-time period starts when the

voltage at FB falls below the internal reference - until then the

inductor current remains zero. In this mode the operating fre-

quency is lower than in continuous conduction mode, and

varies with load current. Therefore at light loads the conver-

sion efficiency is maintained, since the switching losses re-

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LM34923

duce with the reduction in load and frequency. The discon-

tinuous operating frequency can be calculated as follows:

where RL = the load resistance

In continuous conduction mode, current flows continuously

through the inductor and never ramps down to zero. In this

mode the operating frequency is greater than the discontinu-

ous mode frequency and remains relatively constant with load

and line variations. The approximate continuous mode oper-

ating frequency can be calculated as follows:

(1)

The buck switch duty cycle is approximately equal to:

(2)

The output voltage (VOUT) is programmed by two external re-

sistors as shown in the Block Diagram. The regulation point

can be calculated as follows:

VOUT = 2.5 x (RFB1 + RFB2) / RFB1

The LM34923 regulates the output voltage based on ripple

voltage at the feedback input, requiring a minimum amount of

ESR for the output capacitor C2. A minimum of 25mV to 50mV

of ripple voltage at the feedback pin (FB) is required for the

LM34923. In cases where the capacitor ESR is too small, ad-

ditional series resistance may be required (R3 in the Block

Diagram).

For applications where lower output voltage ripple is required

the output can be taken directly from a low ESR output ca-

pacitor, as shown in Figure 1. However, R3 slightly degrades

the load regulation.

30141713

FIGURE 1. Low Ripple Output Configuration

Start-Up Regulator (VCC)

The high voltage bias regulator is integrated within the

LM34923. The input pin (VIN) can be connected directly to

line voltages between 6V and 75V, with transient capability to

80V. The VCC output is regulated at 7.5V. The VCC regulator

output current is limited at approximately 30 mA.

C3 must be located as close as possible to the VCC and RTN

pins. In applications with a relatively high input voltage, power

dissipation in the bias regulator is a concern. An auxiliary

voltage of between 7.5V and 10V can be diode connected to

the VCC pin to shut off the VCC regulator, thereby reducing

internal power dissipation. The current required into the VCC

pin depends on the voltage applied to VCC and the switching

frequency. See the graph “ICC vs. Externally Applied VCC.” In-

ternally a diode connects VCC to VIN requiring that the aux-

iliary voltage be less than VIN.

The turn-on sequence is shown in Figure 2. During the initial

delay (t1) VCC ramps up at a rate determined by its current

limit and C3 while internal circuitry stabilizes. When VCC

reaches the upper threshold of its under-voltage lock-out, the

buckswitch is enabled. The inductor current increases to the

current limit threshold (ILIM) and during t2 VOUT increases as

the output capacitor charges up. When VOUT reaches the in-

tended voltage the average inductor current decreases (t3) to

the nominal load current (IO).

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LM34923

30141714

FIGURE 2. Startup Sequence

Regulation Comparator

The feedback voltage at FB is compared to an internal 2.5V

reference. In normal operation (the output voltage is regulat-

ed), an on-time period is initiated when the voltage at FB falls

below 2.5V. The buck switch stays on for the on-time, causing

the FB voltage to rise above 2.5V. After the on-time period,

the buck switch stays off until the FB voltage again falls below

2.5V. During start-up, the FB voltage will be below 2.5V at the

end of each on-time, resulting in the minimum off-time of 260

ns.

Over-Voltage Comparator

The feedback voltage at FB is compared to an internal 2.85V

reference. If the voltage at FB rises above 2.85V the on-time

pulse is immediately terminated. This condition can occur if

the input voltage, or the output load, change suddenly. The

buck switch will not turn on again until the voltage at FB falls

below 2.5V.

On-Time Generator and Shutdown

The on-time for the LM34923 is determined by the RT resistor,

and is inversely proportional to the input voltage (Vin), result-

ing in a nearly constant frequency as Vin is varied over its

range. The on-time equation for the LM34923 is:

(3)

RT should be selected for a minimum on-time (at maximum

VIN) greater than 200 ns, for proper current limit operation.

This requirement limits the maximum frequency for each ap-

plication, depending on VIN and VOUT.

The LM34923 can be remotely disabled by taking the RT/SD

pin to ground. See Figure 3. The voltage at the RT/SD pin is

between 1.5 and 5.0 volts, depending on Vin and the value of

the RT resistor.

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LM34923

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FIGURE 3. Shutdown Implementation

Current Limit

The LM34923 contains an intelligent current limit OFF timer.

If the current in the Buck switch reaches the current limit

threshold, the present cycle is immediately terminated, and a

non-resetable OFF timer is triggered. The length of off-time is

controlled by the FB voltage and VIN (see the graph Current

Limit Off-Time vs. VFB). When FB = 0V, a maximum off-time

is required. This condition occurs when the output is shorted,

and during the initial part of start-up. This amount of time en-

sures safe short circuit operation up to the maximum input

voltage of 75V. In cases of overload where the FB voltage is

above zero volts (not a short circuit) the required current limit

off-time is less. Reducing the off-time during less severe over-

loads reduces the amount of foldback, recovery time, and the

start-up time. The off-time in microseconds is calculated from

the following equation:

The current limit sensing circuit is blanked for the first 50-70

ns of each on-time so it is not falsely tripped by the current

surge which occurs at turn-on. The current surge is required

by the re-circulating diode (D1) for its turn-off recovery.

N - Channel Buck Switch and Driver

The LM34923 integrates an N-Channel Buck switch and as-

sociated floating high voltage gate driver. The gate driver

circuit works in conjunction with an external bootstrap capac-

itor and an internal high voltage diode. A 0.01 µF ceramic

capacitor (C4) connected between the BST pin and SW pin

provides the voltage to the driver during the on-time.

During each off-time, the SW pin is at approximately 0V, and

the bootstrap capacitor charges from Vcc through the internal

diode. The minimum OFF timer, set to 260 ns, ensures a min-

imum time each cycle to recharge the bootstrap capacitor.

The internal pre-charge switch at the SW pin is turned on for

≊150 ns during the minimum off-time period, ensuring suffi-

cient voltage exists across the bootstrap capacitor for the on-

time. This feature helps prevent operating problems which

can occur during very light load conditions, involving a long

off-time, during which the voltage across the bootstrap ca-

pacitor could otherwise reduce below the Gate Drive UVLO

threshold. The pre-charge switch also helps prevent startup

problems which can occur if the output voltage is pre-charged

prior to turn-on. After current limit detection, the pre-charge

switch is turned on for the entire duration of the forced off-

time .

Under Voltage Detector

The Under Voltage Detector can be used to monitor the input

voltage, or any other system voltage as long as the voltage at

the UV pin does not exceed its maximum rating.

The Under Voltage Output indicator pin (UVO) is connected

to the drain of an internal N-channel MOSFET capable of

sustaining 10V in the off-state. An external pull-up resistor is

required at UVO to an appropriate voltage to indicate the sta-

tus to downstream circuitry. The off-state voltage at the UVO

pin can be higher or lower than the voltage at VIN, but must

not exceed 10V.

The UVO pin switches low when the voltage at the UV input

pin is above its threshold. Typically the monitored voltage

threshold is set with a resistor divider (RUV1, RUV2) as shown

in the Block Diagram. When the voltage at the UV pin is below

its threshold, the internal 5 µA current source at UV is en-

abled. As the input voltage increases, taking UV above its

threshold, the current source is disabled, raising the voltage

at UV to provide threshold hysteresis.

The UVO output is high when the VCC voltage is below its

UVLO threshold, or when the LM34923 is shutdown using the

RT/SD pin (see Figure 3), regardless of the voltage at the UV

pin.

Thermal Protection

The LM34923 should be operated so the junction temperature

does not exceed 125°C during normal operation. An internal

Thermal Shutdown circuit is provided to shutdown the

LM34923 in the event of a higher than normal junction tem-

perature. When activated, typically at 165°C, the controller is

forced into a low power reset state by disabling the buck

switch. This feature prevents catastrophic failures from acci-

dental device overheating. When the junction temperature

reduces below 145°C (typical hysteresis = 20°C) normal op-

eration is resumed.

Applications Information

SELECTION OF EXTERNAL COMPONENTS

A guide for determining the component values is illustrated

with a design example. Refer to the Block Diagram. The fol-

lowing steps will configure the LM34923 for:

•Input voltage range (Vin): 15V to 75V

•Output voltage (VOUT): 10V

•Load current (for continuous conduction mode): 100 mA

to 400 mA

•Switching Frequency: 300 kHz

RFB1, RFB2: VOUT = VFB x (RFB1 + RFB2) / RFB1, and since

VFB = 2.5V, the ratio of RFB2 to RFB1 calculates as 3:1. Stan-

dard values of 3.01 kΩ and 1.00 kΩ are chosen. Other values

could be used as long as the 3:1 ratio is maintained.

Fs and RT: Unless the application requires a specific frequen-

cy, the choice of frequency is generally a compromise. A

higher frequency allows for a smaller inductor, input capaci-

tor, and output capacitor (both in value and physical size),

while providing a lower conversion efficiency. A lower fre-

quency provides higher efficiency, but generally requires

higher values for the inductor, input capacitor and output ca-

pacitor. The maximum allowed switching frequency for the

LM34923 is limited by the minimum on-time (200 ns) at the

maximum input voltage, and by the minimum off-time (260 ns)

at the minimum input voltage. The maximum frequency limit

for each application is defined by the following two calcula-

tions:

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LM34923

The maximum allowed frequency is the lesser of the two

above calculations. See the graph “Maximum Switching Fre-

quency”. For this exercise, Fs(max)1 calculates to 667 kHz, and

Fs(max)2 calculates to 1.28 MHz. Therefore the maximum al-

lowed frequency for this example is 667 kHz, which is greater

than the 300 kHz specified for this design. Using Equation 1,

RT calculates to 258 kΩ. A standard value 261 kΩ resistor is

used. The minimum on-time calculates to 469 ns, and the

maximum on-time calculates to 2.28 µs.

L1: The main parameter affected by the inductor is the output

current ripple amplitude. The choice of inductor value there-

fore depends on both the minimum and maximum load cur-

rents, keeping in mind that the maximum ripple current occurs

at maximum Vin.

a) Minimum load current: To maintain continuous conduc-

tion at minimum Io (100 mA) if a flyback diode is used, the

ripple amplitude (IOR) must be less than 200 mA p-p so the

lower peak of the waveform does not reach zero. L1 is cal-

culated using the following equation:

At Vin = 75V, L1(min) calculates to 146µH. The next larger

standard value (150 µH) is chosen and with this value IOR

calculates to 195 mA p-p at Vin = 75V, and 75 mA p-p at Vin

= 15V.

b) Maximum load current: At a load current of 400 mA, the

peak of the ripple waveform must not reach the minimum

guaranteed value of the LM34923’s current limit threshold

(700 mA). Therefore the ripple amplitude must be less than

600 mA p-p, which is already satisfied in the above calcula-

tion. With L1 = 150 µH, at maximum Vin and Io, the peak of

the ripple is 498 mA. While L1 must carry this peak current

without saturating or exceeding its temperature rating, it also

must be capable of carrying the maximum guaranteed value

of the LM34923’s current limit threshold without saturating,

since the current limit is reached during startup.

The DC resistance of the inductor should be as low as pos-

sible. For example, if the inductor’s DCR is 0.5 ohm, the power

dissipated at maximum load current is 0.08W. While small, it

is not insignificant compared to the load power of 4W.

C3: The capacitor on the VCC output provides not only noise

filtering and stability, but its primary purpose is to prevent false

triggering of the VCC UVLO at the buck switch on/off transi-

tions. C3 should be no smaller than 1 µF.

C2 and R3: When selecting the output filter capacitor C2, the

items to consider are ripple voltage due to its ESR, ripple

voltage due to its capacitance, and the nature of the load.

A low ESR for C2 is generally desirable so as to minimize

power losses and heating within the capacitor. However, the

regulator requires a minimum amount of ripple voltage at the

feedback input for proper loop operation. For the LM34923

the minimum ripple required at pin 7 is 25 mV p-p, requiring

a minimum ripple at VOUT of 100 mV for this example. Since

the minimum ripple current (at minimum Vin) is 75 mA p-p,

the minimum ESR required at VOUT is 100 mV/75 mA =

1.33Ω. Since quality capacitors for SMPS applications have

an ESR considerably less than this, R3 is inserted as shown

in the Block Diagram. R3’s value, along with C2’s ESR, must

result in at least 25 mV p-p ripple at pin 7. See the Low Output

Ripple Configuration section for techniques to reduce the out-

put ripple voltage.

D1: A power Schottky diode is recommended. Ultra-fast re-

covery diodes are not recommended as the high speed tran-

sitions at the SW pin may inadvertently affect the IC’s

operation through external or internal EMI. The important pa-

rameters are reverse recovery time and forward voltage. The

reverse recovery time determines how long the reverse cur-

rent surge lasts with each turn-on of the internal buck switch.

The forward voltage drop affects efficiency. The diode’s re-

verse voltage rating must be at least as great as the maximum

input voltage, plus ripple and transients, and its current rating

must be at least as great as the maximum current limit spec-

ification. The diode’s average power dissipation is calculated

from:

PD1 = VF x IOUT x (1–D) (4)

Where VF is the diode’s forward voltage drop, and D is the on-

time duty cycle.

C1: This capacitor’s purpose is to supply most of the switch

current during the on-time, and limit the voltage ripple at Vin,

on the assumption that the voltage source feeding Vin has an

output impedance greater than zero. At maximum load cur-

rent, when the buck switch turns on, the current into the VIN

pin suddenly increases to the lower peak of the output current

waveform, ramp up to the peak value, then drop to zero at

turn-off. The average input current during this on-time is the

load current (400 mA). For a worst case calculation, C1 must

supply this average load current during the maximum on-time.

To keep the input voltage ripple to less than 1V (for this ex-

ercise), C1 calculates to:

Quality ceramic capacitors in this value have a low ESR which

adds only a few millivolts to the ripple. It is the capacitance

which is dominant in this case. To allow for the capacitor’s

tolerance, temperature effects, and voltage effects, a 1.0 µF,

100V, X7R capacitor is used.

C4: The recommended value is 0.01µF for C4, as this is ap-

propriate in the majority of applications. A high quality ceramic

capacitor, with low ESR is recommended as C4 supplies the

surge current to charge the buck switch gate at turn-on. A low

ESR also ensures a quick recharge during each off-time.

C5: This capacitor helps avoid supply voltage transients and

ringing due to long lead inductance at VIN. A low ESR, 0.1µF

ceramic chip capacitor is recommended, located close to the

LM34923.

UV and UVO pins: The Under Voltage Detector function is

used to monitor a system voltage, such as the input voltage

at VIN, by connecting the UV pin to two resistors (RUV1,

RUV2) as shown in the Block Diagram. When the voltage at

the UV pin increases above its threshold the UVO pin switch-

es low. The UVO pin is high when the voltage at the UV input

pin is below its threshold. Hysteresis is provided by the inter-

nal 5µA current source which is enabled when the voltage at

the UV pin is below its threshold. The resistor values are cal-

culated using the following procedure:

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LM34923

Choose the upper and lower thresholds (VUVH and VUVL) at

VIN.

As an example, assume the application requires the following

thresholds: VUVH = 15V and VUVL = 14V. Therefore VUV(HYS)

= 1V. The resistor values calculate to:

RUV2 = 200kΩ, RUV1 = 43.5kΩ(5)

Capacitor C6 is added to filter noise and ripple, which may be

present on the VIN line. Where the resistor values are known,

the threshold voltages and hysteresis are calculated from the

following:

VUV(HYS) = RUV2 x 5 µA (6)

The pull-up voltage for the UVO output can be any voltage

under 10V. The maximum continuous current into the UVO

output pin should not exceed 5 mA.

FINAL CIRCUIT

The final circuit is shown in Figure 4. The circuit was tested,

and the resulting performance is shown in Figure 6 and Figure

30141718

FIGURE 4. LM34923 Example Circuit

15 www.national.com

LM34923

30141724

FIGURE 5. Efficiency vs. Load Current and VIN

30141728

FIGURE 6. Efficiency vs. VIN

LOW OUTPUT RIPPLE CONFIGURATIONS

For applications where low output ripple is required, the fol-

lowing options can be used to reduce or nearly eliminate the

ripple.

a) Reduced ripple configuration: In Figure 7, Cff is added

across RFB2 to AC-couple the ripple at VOUT directly to the FB

pin. This allows the ripple at VOUT to be reduced to a minimum

of 25 mVp-p by reducing R3, since the ripple at VOUT is not

attenuated by the feedback resistors. The minimum value for

Cff is determined from:

where tON(max) is the maximum on-time, which occurs at the

minimum input voltage. The next larger standard value ca-

pacitor should be used for Cff.

www.national.com 16

LM34923

30141721

FIGURE 7. Reduced Ripple Configuration

b) Minimum ripple configuration: If the application requires

a lower value of ripple (<10 mVp-p), the circuit of Figure 8 can

be used. R3 is removed, and the resulting output ripple volt-

age is determined by the inductor’s ripple current and C2’s

characteristics. RA and CA are chosen to generate a saw-

tooth waveform at their junction, and that voltage is AC-

coupled to the FB pin via CB. To determine the values for RA,

CA and CB, use the following procedure:

Calculate VA = VOUT - (VSW x (1 - (VOUT/VIN(min))))

where VSW is the absolute value of the voltage at the SW pin

during the off-time. If a Schottky diode is used for the flyback

function, the off-time voltage is in the range of 0.5V to 1V,

depending on the specific diode used, and the maximum load

current. VA is the DC voltage at the RA/CA junction, and is

used in the next equation.

- Calculate RA x CA = (VIN(min) - VA) x tON/ΔV

where tON is the maximum on-time (at minimum input volt-

age), and ΔV is the desired ripple amplitude at the RA/CA

junction (typically 40-50 mV). RA and CA are then chosen

from standard value components to satisfy the above product.

Typically CA is 1000 pF to 5000 pF, and RA is 10 kΩ to 300

kΩ. CB is then chosen large compared to CA, typically 0.1 µF.

30141722

FIGURE 8. Minimum Output Ripple Using Ripple Injection

c) Alternate minimum ripple configuration: The circuit in

Figure 9 is the same as that in the Block Diagram, except the

output voltage is taken from the junction of R3 and C2. The

ripple at VOUT is determined by the inductor’s ripple current

and C2’s characteristics. However, R3 slightly degrades the

load regulation. This circuit may be suitable if the load current

is fairly constant.

30141723

FIGURE 9. Alternate Minimum Output Ripple

PC Board Layout

The LM34923 regulation, over-voltage, and current limit com-

parators are very fast, and respond to short duration noise

pulses. Layout considerations are therefore critical for opti-

mum performance. The layout must be as neat and compact

as possible, and all of the components must be as close as

possible to the associated pins. The two major current loops

have currents which switch very fast, and so the loops should

be as small as possible to minimize conducted and radiated

EMI. The first loop is formed by C1, through VIN to the SW

pin, L1, C2, and back to C1. The second loop is formed by L1,

C2, D1, and back to L1. Since a current equal to the load

current switches between these two loops with each transition

from on-time to off-time and back to on-time, it is imperative

that the ground end of C1 have a short and direct connection

to D1’s anode, without going through vias or a lengthy route.

The power dissipation in the LM34923 can be approximated

by determining the total conversion loss (PIN – POUT), and then

subtracting the power losses in D1, and in the inductor. The

power loss in the diode is approximately:

PD1 = IOUT x VF x (1–D)

where VF is the diode’s forward voltage drop, and D is the on-

time duty cycle.

PL1 = IOUT2 x RL x 1.1

where RL is the inductor’s DC resistance, and the 1.1 factor

is an approximation for the AC losses. If it is expected that the

internal dissipation of the LM34923 will produce excessive

junction temperatures during normal operation, good use of

the PC board’s ground plane can help to dissipate heat. Ad-

ditionally the use of wide PC board traces, where possible,

can help conduct heat away from the IC. Judicious positioning

of the PC board within the end product, along with the use of

any available air flow (forced or natural convection) can help

reduce the junction temperature.

17 www.national.com

LM34923

Physical Dimensions inches (millimeters) unless otherwise noted

Mini SO Molded

NS Package MUB10A

www.national.com 18

LM34923

Notes

19 www.national.com

LM34923

Notes

LM34923 80V, 600 mA Constant On-Time Buck Switching Regulator

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