LM34923 LM34923 80V, 600 mA Constant On-Time Buck Switching Regulator Literature Number: SNVS695 LM34923 80V, 600 mA Constant On-Time Buck Switching Regulator General Description Features The LM34923 Step Down Switching Regulator features all of the functions needed to implement a low cost, efficient Buck bias regulator. This high voltage regulator contains an 80V NChannel MOSFET Switch and a startup regulator. The device is easy to implement and is provided in an MSOP-10 package. The regulator's control scheme uses an on-time inversely proportional to VIN. This feature results in the operating frequency remaining relatively constant with line and load variations. The control scheme requires no loop compensation, resulting in fast transient response. An intelligent current limit is implemented with a forced off-time which is inversely proportional to VOUT. This scheme ensures short circuit control while providing minimum foldback. Other features include: Thermal Shutdown, VCC Under Voltage Lock-out, Max Duty Cycle Limiter, a Pre-charge Switch, and a programmable Under Voltage Detector with a status flag output. Operating input voltage range: 6V to 75V Integrated 80V, N-Channel Buck Switch Internal start-up regulator No Loop compensation required Ultra-Fast Transient Response Operating frequency remains constant with line and load variations Adjustable output voltage from 2.5V Precision internal reference, 2.5% Intelligent current limit reduces foldback Programmable Input UV detector with status flag output Pre-charge switch enables bootstrap gate drive with no load Thermal shutdown Typical Applications Non-Isolated Telecommunication Buck Regulator Secondary High Voltage Post Regulator +42V Automotive Systems Package MSOP - 10 Typical Application, Basic Step-Down Regulator 30141701 (c) 2011 National Semiconductor Corporation 301417 www.national.com LM34923 80V, 600 mA Constant On-Time Buck Switching Regulator March 1, 2011 LM34923 Connection Diagram 30141702 Top View 10-Lead MSOP Ordering Information Order Number Package Type LM34923MM NSC Package Drawing MSOP-10 LM34923MMX MUB10A Supplied As 1000 Units on Tape and Reel 3500 Units on Tape and Reel Pin Descriptions Pin Name 1 SW Switching Node Power switching node. Connect to the output inductor, re-circulating diode or synchronous FET, and bootstrap capacitor. 2 BST Boost Pin An external capacitor is required between the BST and the SW pins (0.01uF or greater ceramic). The BST pin capacitor is charged from Vcc through an internal diode when SW is low. 3 N/C Do not connect 4 RTN Ground pin Ground for the entire circuit. 5 UV Input pin for the under voltage indicator A resistor divider from VIN, or some other system voltage, programs the undervoltage detection threshold. An internal current sink is enabled when UV is below 2.5V to provide hysteresis. 6 UVO Under voltage status indicator This open drain output is high when the UV pin voltage is below 2.5V, or when the VCCUVLO function or the shutdown function is invoked. 7 FB Feedback Input from Regulated Output This pin is connected to the inverting input of the internal regulation comparator. The regulation level is 2.5V. 8 RT/SD On-time set pin and shutdown input A resistor between this pin and Vin sets the switch on-time as a function of Vin, and the frequency. The minimum recommended on-time is 200 ns at max input voltage. Taking this pin to ground shuts off the regulator. 9 VCC Output from the internal high voltage The internal regulator provides bias supply for the Buck switch gate driver and series pass regulator. Regulated at other internal circuitry. A 1uF ceramic capacitor to ground is required. The 7.5V. regulator is current limited to 30 mA. 10 VIN Input Voltage www.national.com Description Application Information The operating input range is 6V to 75V 2 If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN, UV to RTN BST to RTN SW to RTN (Steady State) BST to VCC BST to SW VCC, UVO, RT to RTN FB, RT, to RTN -0.3V to 80V -0.3V to 88V -1V to VIN + 0.3V 80V 10V -0.3V to 10V -0.3 to 5V Operating Ratings (Note 1) VIN Operating Junction Temperature 6V to 75V -40C to + 125C Electrical Characteristics Specifications with standard type are for TJ = 25C only; limits in boldface type apply over the full Operating Junction Temperature (TJ) range. Minimum and Maximum limits are guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25C, and are provided for reference purposes only. Unless otherwise stated the following conditions apply: VIN = 48V(Note 3). Symbol Parameter Conditions Min Typ Max 7.5 7.9 Units VCC Supply Vcc Reg Vcc Regulator Output Vin - Vcc Vcc Output Impedance Vcc Current Limit Vcc UVLO Vin = 48V 7.1 V VIN = 6V, ICC = 5mA 240 mV Vin =6V 45 mA Vin = 48V (Note 4) 20 Vcc Increasing 30 4 Vcc UVLO hysteresis 4.8 450 V mV Iin Operating current FB = 3V, Vin = 48V 1 1.32 mA Iin Shutdown Current RT/SD = 0V 20 70 A 0.56 1.1 3 3.8 V Switch Characteristics Buckswitch Rds(on) Gate Drive UVLO Itest = 200 mA Vbst - Vsw Rising 2.15 Gate Drive UVLO hysteresis Pre-charge switch voltage 250 At 1 mA Pre-charge switch on-time mV 0.8 V 150 ns Current Limit Current Limit Threshold 700 1175 1500 mA Current Limit Response Time Iswitch = 1.24A, Time to Switch Off 190 ns TOFF-1 OFF time generator (test 1) FB=0V, VIN = 75V 37 s TOFF-2 OFF time generator (test 2) FB=2.3V, VIN = 75V 7.2 s TOFF-3 OFF time generator (test 3) FB=0V, VIN = 10V 5.7 s TOFF-4 OFF time generator (test 4) FB=2.3V, VIN = 10V 1.25 s On Time Generator TON - 1 On-Time Vin = 10V Ron = 250K 2.2 3.3 4.51 s TON - 2 On-Time Vin = 75V Ron = 250K 300 450 565 ns Remote Shutdown Threshold Voltage at RT/SD rising 0.46 0.9 1.4 Remote Shutdown Hysteresis 60 3 V mV www.national.com LM34923 ESD Rating (Note 5) Human Body Model 2kV For soldering specs see: www.national.com/ms/MS/MS-SOLDERING.pdf Junction Temperature 150C Storage Temperature Range -55C to +150C Absolute Maximum Ratings (Note 1) LM34923 Symbol Parameter Conditions Min Typ Max Units 260 347 ns 2.5 2.5625 V Minimum Off Time Minimum Off Time VIN = 6V Regulation and OV Comparators FB Reference Threshold Internal reference Trip point for switch ON FB Over-Voltage Threshold Trip point for switch OFF 2.4365 FB Bias Current 2.85 V 1 nA Under Voltage Sensing UVTH UV Threshold UVHYS UV Hysteresis Current UV = 2V 2.4 2.5 2.6 V 2.7 5 7.3 uA UVBIAS UV Bias Current UV = 3V UVOVOL UVO Output Low Voltage UV = 3V, IUVO = 5mA 360 600 mV UVOIOH UVO Leakage Current UV = 2V, VUVO = 7.8V 1 nA Thermal Shutdown Temp. 165 C Thermal Shutdown Hysteresis 20 C 200 C/W 1 nA Thermal Shutdown Tsd Thermal Resistance JA Junction to Ambient MUA Package Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics. Note 2: For detailed information on soldering plastic MSOP packages, refer to the Packaging Data Book available from National Semiconductor Corporation. Note 3: All limits are guaranteed. All electrical characteristics having room temperature limits are tested during production with TA = TJ = 25C. All hot and cold limits are guaranteed by correlating the electrical characteristics to process and temperature variations and applying statistical process control. Note 4: The VCC output is intended as a self bias for the internal gate drive power and control circuits. Device thermal limitations limit external loading. Note 5: The human body model is a 100pF capacitor discharged through a 1.5k resistor into each pin. www.national.com 4 LM34923 Typical Performance Characteristics Efficiency at 300 kHz, 10V Efficiency Comparison at 200 kHz 100 EFFICIENCY (%) 90 6V, D1 80 24V, D1 70 7.5V, D1 60 VOUT=5V, D1=DFLS1100 50 0 100 200 300 400 500 LOAD CURRENT (mA) 30141724 600 30141767 VCC vs. VIN VCC vs. ICC 30141725 30141707 ICC vs. Externally Applied VCC On-Time vs. VIN and RT 5 ICC (mA) 4 3 900 kHz, D1 2 200 kHz, D1 1 DCM 0 7.5 8.0 8.5 9.0 APPLIED VCC (V) 9.5 10.0 30141765 30141727 5 www.national.com LM34923 Current Limit Off-Time vs. VFB Maximum Switching Frequency 30141739 30141738 Voltage at the RT Pin Operating Current into VIN 30141740 30141741 Shutdown Current into VIN UVO Pin Low Voltage vs. Sink Current 30141743 30141742 www.national.com 6 LM34923 VCC UVLO vs. Temperature Gate Drive UVLO vs. Temperature 30141746 30141745 VCC vs. Temperature VCC Dropout vs. Temperature 30141747 30141759 VCC Output Impedance vs. Temperature VCC Current Limit vs. Temperature 30141748 30141749 7 www.national.com LM34923 Reference Voltage vs. Temperature On-time vs. Temperature 30141750 30141751 Minimum Off-time vs. Temperature Current Limit Threshold vs. Temperature 30141753 30141752 Current Limit Off-Time vs. Temperature Operating Current vs. Temperature 30141760 www.national.com 30141754 8 RT Pin Shutdown Threshold vs. Temperature 30141755 30141756 UV Pin Threshold vs. Temperature UV Hysteresis Current vs. Temperature 30141761 30141762 9 www.national.com LM34923 Shutdown Current vs. Temperature LM34923 Block Diagram 30141710 Functional Description Control Circuit Overview The LM34923 Step Down Switching Regulator features all the functions needed to implement a low cost, efficient, Buck bias power converter. This high voltage regulator contains an 80 V N-Channel Buck Switch, is easy to implement and is provided in the MSOP-10 package. The regulator is based on a control scheme using an on-time inversely proportional to VIN. The control scheme requires no loop compensation. Current limit is implemented with forced off-time, which is inversely proportional to VOUT. This scheme ensures short circuit control while providing minimum foldback. The LM34923 can be applied in numerous applications to efficiently regulate down higher voltages. This regulator is well suited for 48 Volt Telecom and the new 42V Automotive power bus ranges. Features include: Thermal Shutdown, VCC under-voltage lockout, Gate drive under-voltage lockout, Max Duty Cycle limit timer, intelligent current limit off timer, a precharge switch, and a programmable under voltage detector with status flag. The LM34923 is a Buck DC-DC regulator that uses a control scheme in which the on-time varies inversely with line voltage (VIN). Control is based on a comparator and the on-time oneshot, with the output voltage feedback (FB) compared to an internal reference (2.5V). If the FB level is below the reference the buck switch is turned on for a fixed time determined by the line voltage and a programming resistor (RT). Following the ON period the switch remains off for at least the minimum offtimer period of 260 ns. If FB is still below the reference at that time the switch turns on again for another on-time period. This continues until regulation is achieved. The LM34923 operates in discontinuous conduction mode at light load currents, and continuous conduction mode at heavy load current. In discontinuous conduction mode, current through the output inductor starts at zero and ramps up to a peak during the on-time, then ramps back to zero before the end of the off-time. The next on-time period starts when the voltage at FB falls below the internal reference - until then the inductor current remains zero. In this mode the operating frequency is lower than in continuous conduction mode, and varies with load current. Therefore at light loads the conversion efficiency is maintained, since the switching losses re- www.national.com 10 The output voltage (VOUT) is programmed by two external resistors as shown in the Block Diagram. The regulation point can be calculated as follows: VOUT = 2.5 x (RFB1 + RFB2) / RFB1 The LM34923 regulates the output voltage based on ripple voltage at the feedback input, requiring a minimum amount of ESR for the output capacitor C2. A minimum of 25mV to 50mV of ripple voltage at the feedback pin (FB) is required for the LM34923. In cases where the capacitor ESR is too small, additional series resistance may be required (R3 in the Block Diagram). For applications where lower output voltage ripple is required the output can be taken directly from a low ESR output capacitor, as shown in Figure 1. However, R3 slightly degrades the load regulation. where RL = the load resistance In continuous conduction mode, current flows continuously through the inductor and never ramps down to zero. In this mode the operating frequency is greater than the discontinuous mode frequency and remains relatively constant with load and line variations. The approximate continuous mode operating frequency can be calculated as follows: (1) The buck switch duty cycle is approximately equal to: (2) 30141713 FIGURE 1. Low Ripple Output Configuration pin depends on the voltage applied to VCC and the switching frequency. See the graph "ICC vs. Externally Applied VCC." Internally a diode connects VCC to VIN requiring that the auxiliary voltage be less than VIN. The turn-on sequence is shown in Figure 2. During the initial delay (t1) VCC ramps up at a rate determined by its current limit and C3 while internal circuitry stabilizes. When VCC reaches the upper threshold of its under-voltage lock-out, the buckswitch is enabled. The inductor current increases to the current limit threshold (ILIM) and during t2 VOUT increases as the output capacitor charges up. When VOUT reaches the intended voltage the average inductor current decreases (t3) to the nominal load current (IO). Start-Up Regulator (VCC) The high voltage bias regulator is integrated within the LM34923. The input pin (VIN) can be connected directly to line voltages between 6V and 75V, with transient capability to 80V. The V CC output is regulated at 7.5V. The VCC regulator output current is limited at approximately 30 mA. C3 must be located as close as possible to the VCC and RTN pins. In applications with a relatively high input voltage, power dissipation in the bias regulator is a concern. An auxiliary voltage of between 7.5V and 10V can be diode connected to the VCC pin to shut off the VCC regulator, thereby reducing internal power dissipation. The current required into the VCC 11 www.national.com LM34923 duce with the reduction in load and frequency. The discontinuous operating frequency can be calculated as follows: LM34923 30141714 FIGURE 2. Startup Sequence Regulation Comparator On-Time Generator and Shutdown The feedback voltage at FB is compared to an internal 2.5V reference. In normal operation (the output voltage is regulated), an on-time period is initiated when the voltage at FB falls below 2.5V. The buck switch stays on for the on-time, causing the FB voltage to rise above 2.5V. After the on-time period, the buck switch stays off until the FB voltage again falls below 2.5V. During start-up, the FB voltage will be below 2.5V at the end of each on-time, resulting in the minimum off-time of 260 ns. The on-time for the LM34923 is determined by the RT resistor, and is inversely proportional to the input voltage (Vin), resulting in a nearly constant frequency as Vin is varied over its range. The on-time equation for the LM34923 is: (3) RT should be selected for a minimum on-time (at maximum VIN) greater than 200 ns, for proper current limit operation. This requirement limits the maximum frequency for each application, depending on VIN and VOUT. The LM34923 can be remotely disabled by taking the RT/SD pin to ground. See Figure 3. The voltage at the RT/SD pin is between 1.5 and 5.0 volts, depending on Vin and the value of the RT resistor. Over-Voltage Comparator The feedback voltage at FB is compared to an internal 2.85V reference. If the voltage at FB rises above 2.85V the on-time pulse is immediately terminated. This condition can occur if the input voltage, or the output load, change suddenly. The buck switch will not turn on again until the voltage at FB falls below 2.5V. www.national.com 12 The Under Voltage Detector can be used to monitor the input voltage, or any other system voltage as long as the voltage at the UV pin does not exceed its maximum rating. The Under Voltage Output indicator pin (UVO) is connected to the drain of an internal N-channel MOSFET capable of sustaining 10V in the off-state. An external pull-up resistor is required at UVO to an appropriate voltage to indicate the status to downstream circuitry. The off-state voltage at the UVO pin can be higher or lower than the voltage at VIN, but must not exceed 10V. The UVO pin switches low when the voltage at the UV input pin is above its threshold. Typically the monitored voltage threshold is set with a resistor divider (RUV1, RUV2) as shown in the Block Diagram. When the voltage at the UV pin is below its threshold, the internal 5 A current source at UV is enabled. As the input voltage increases, taking UV above its threshold, the current source is disabled, raising the voltage at UV to provide threshold hysteresis. The UVO output is high when the VCC voltage is below its UVLO threshold, or when the LM34923 is shutdown using the RT/SD pin (see Figure 3), regardless of the voltage at the UV pin. 30141715 FIGURE 3. Shutdown Implementation Current Limit The LM34923 contains an intelligent current limit OFF timer. If the current in the Buck switch reaches the current limit threshold, the present cycle is immediately terminated, and a non-resetable OFF timer is triggered. The length of off-time is controlled by the FB voltage and VIN (see the graph Current Limit Off-Time vs. VFB). When FB = 0V, a maximum off-time is required. This condition occurs when the output is shorted, and during the initial part of start-up. This amount of time ensures safe short circuit operation up to the maximum input voltage of 75V. In cases of overload where the FB voltage is above zero volts (not a short circuit) the required current limit off-time is less. Reducing the off-time during less severe overloads reduces the amount of foldback, recovery time, and the start-up time. The off-time in microseconds is calculated from the following equation: Thermal Protection The LM34923 should be operated so the junction temperature does not exceed 125C during normal operation. An internal Thermal Shutdown circuit is provided to shutdown the LM34923 in the event of a higher than normal junction temperature. When activated, typically at 165C, the controller is forced into a low power reset state by disabling the buck switch. This feature prevents catastrophic failures from accidental device overheating. When the junction temperature reduces below 145C (typical hysteresis = 20C) normal operation is resumed. The current limit sensing circuit is blanked for the first 50-70 ns of each on-time so it is not falsely tripped by the current surge which occurs at turn-on. The current surge is required by the re-circulating diode (D1) for its turn-off recovery. Applications Information SELECTION OF EXTERNAL COMPONENTS A guide for determining the component values is illustrated with a design example. Refer to the Block Diagram. The following steps will configure the LM34923 for: * Input voltage range (Vin): 15V to 75V * Output voltage (VOUT): 10V * Load current (for continuous conduction mode): 100 mA to 400 mA * Switching Frequency: 300 kHz RFB1, RFB2: VOUT = VFB x (RFB1 + RFB2) / RFB1, and since VFB = 2.5V, the ratio of RFB2 to RFB1 calculates as 3:1. Standard values of 3.01 k and 1.00 k are chosen. Other values could be used as long as the 3:1 ratio is maintained. Fs and RT: Unless the application requires a specific frequency, the choice of frequency is generally a compromise. A higher frequency allows for a smaller inductor, input capacitor, and output capacitor (both in value and physical size), while providing a lower conversion efficiency. A lower frequency provides higher efficiency, but generally requires higher values for the inductor, input capacitor and output capacitor. The maximum allowed switching frequency for the LM34923 is limited by the minimum on-time (200 ns) at the maximum input voltage, and by the minimum off-time (260 ns) at the minimum input voltage. The maximum frequency limit for each application is defined by the following two calculations: N - Channel Buck Switch and Driver The LM34923 integrates an N-Channel Buck switch and associated floating high voltage gate driver. The gate driver circuit works in conjunction with an external bootstrap capacitor and an internal high voltage diode. A 0.01 F ceramic capacitor (C4) connected between the BST pin and SW pin provides the voltage to the driver during the on-time. During each off-time, the SW pin is at approximately 0V, and the bootstrap capacitor charges from Vcc through the internal diode. The minimum OFF timer, set to 260 ns, ensures a minimum time each cycle to recharge the bootstrap capacitor. The internal pre-charge switch at the SW pin is turned on for 150 ns during the minimum off-time period, ensuring sufficient voltage exists across the bootstrap capacitor for the ontime. This feature helps prevent operating problems which can occur during very light load conditions, involving a long off-time, during which the voltage across the bootstrap capacitor could otherwise reduce below the Gate Drive UVLO threshold. The pre-charge switch also helps prevent startup problems which can occur if the output voltage is pre-charged prior to turn-on. After current limit detection, the pre-charge switch is turned on for the entire duration of the forced offtime . 13 www.national.com LM34923 Under Voltage Detector LM34923 the minimum ripple current (at minimum Vin) is 75 mA p-p, the minimum ESR required at VOUT is 100 mV/75 mA = 1.33. Since quality capacitors for SMPS applications have an ESR considerably less than this, R3 is inserted as shown in the Block Diagram. R3's value, along with C2's ESR, must result in at least 25 mV p-p ripple at pin 7. See the Low Output Ripple Configuration section for techniques to reduce the output ripple voltage. D1: A power Schottky diode is recommended. Ultra-fast recovery diodes are not recommended as the high speed transitions at the SW pin may inadvertently affect the IC's operation through external or internal EMI. The important parameters are reverse recovery time and forward voltage. The reverse recovery time determines how long the reverse current surge lasts with each turn-on of the internal buck switch. The forward voltage drop affects efficiency. The diode's reverse voltage rating must be at least as great as the maximum input voltage, plus ripple and transients, and its current rating must be at least as great as the maximum current limit specification. The diode's average power dissipation is calculated from: The maximum allowed frequency is the lesser of the two above calculations. See the graph "Maximum Switching Frequency". For this exercise, Fs(max)1 calculates to 667 kHz, and Fs(max)2 calculates to 1.28 MHz. Therefore the maximum allowed frequency for this example is 667 kHz, which is greater than the 300 kHz specified for this design. Using Equation 1, RT calculates to 258 k. A standard value 261 k resistor is used. The minimum on-time calculates to 469 ns, and the maximum on-time calculates to 2.28 s. L1: The main parameter affected by the inductor is the output current ripple amplitude. The choice of inductor value therefore depends on both the minimum and maximum load currents, keeping in mind that the maximum ripple current occurs at maximum Vin. a) Minimum load current: To maintain continuous conduction at minimum Io (100 mA) if a flyback diode is used, the ripple amplitude (IOR) must be less than 200 mA p-p so the lower peak of the waveform does not reach zero. L1 is calculated using the following equation: PD1 = VF x IOUT x (1-D) Where VF is the diode's forward voltage drop, and D is the ontime duty cycle. C1: This capacitor's purpose is to supply most of the switch current during the on-time, and limit the voltage ripple at Vin, on the assumption that the voltage source feeding Vin has an output impedance greater than zero. At maximum load current, when the buck switch turns on, the current into the VIN pin suddenly increases to the lower peak of the output current waveform, ramp up to the peak value, then drop to zero at turn-off. The average input current during this on-time is the load current (400 mA). For a worst case calculation, C1 must supply this average load current during the maximum on-time. To keep the input voltage ripple to less than 1V (for this exercise), C1 calculates to: At Vin = 75V, L1(min) calculates to 146H. The next larger standard value (150 H) is chosen and with this value IOR calculates to 195 mA p-p at Vin = 75V, and 75 mA p-p at Vin = 15V. b) Maximum load current: At a load current of 400 mA, the peak of the ripple waveform must not reach the minimum guaranteed value of the LM34923's current limit threshold (700 mA). Therefore the ripple amplitude must be less than 600 mA p-p, which is already satisfied in the above calculation. With L1 = 150 H, at maximum Vin and Io, the peak of the ripple is 498 mA. While L1 must carry this peak current without saturating or exceeding its temperature rating, it also must be capable of carrying the maximum guaranteed value of the LM34923's current limit threshold without saturating, since the current limit is reached during startup. The DC resistance of the inductor should be as low as possible. For example, if the inductor's DCR is 0.5 ohm, the power dissipated at maximum load current is 0.08W. While small, it is not insignificant compared to the load power of 4W. C3: The capacitor on the VCC output provides not only noise filtering and stability, but its primary purpose is to prevent false triggering of the VCC UVLO at the buck switch on/off transitions. C3 should be no smaller than 1 F. C2 and R3: When selecting the output filter capacitor C2, the items to consider are ripple voltage due to its ESR, ripple voltage due to its capacitance, and the nature of the load. A low ESR for C2 is generally desirable so as to minimize power losses and heating within the capacitor. However, the regulator requires a minimum amount of ripple voltage at the feedback input for proper loop operation. For the LM34923 the minimum ripple required at pin 7 is 25 mV p-p, requiring a minimum ripple at VOUT of 100 mV for this example. Since www.national.com (4) Quality ceramic capacitors in this value have a low ESR which adds only a few millivolts to the ripple. It is the capacitance which is dominant in this case. To allow for the capacitor's tolerance, temperature effects, and voltage effects, a 1.0 F, 100V, X7R capacitor is used. C4: The recommended value is 0.01F for C4, as this is appropriate in the majority of applications. A high quality ceramic capacitor, with low ESR is recommended as C4 supplies the surge current to charge the buck switch gate at turn-on. A low ESR also ensures a quick recharge during each off-time. C5: This capacitor helps avoid supply voltage transients and ringing due to long lead inductance at VIN. A low ESR, 0.1F ceramic chip capacitor is recommended, located close to the LM34923. UV and UVO pins: The Under Voltage Detector function is used to monitor a system voltage, such as the input voltage at VIN, by connecting the UV pin to two resistors (RUV1, RUV2) as shown in the Block Diagram. When the voltage at the UV pin increases above its threshold the UVO pin switches low. The UVO pin is high when the voltage at the UV input pin is below its threshold. Hysteresis is provided by the internal 5A current source which is enabled when the voltage at the UV pin is below its threshold. The resistor values are calculated using the following procedure: 14 LM34923 Choose the upper and lower thresholds (VUVH and VUVL) at VIN. VUV(HYS) = RUV2 x 5 A The pull-up voltage for the UVO output can be any voltage under 10V. The maximum continuous current into the UVO output pin should not exceed 5 mA. As an example, assume the application requires the following thresholds: VUVH = 15V and VUVL = 14V. Therefore VUV(HYS) = 1V. The resistor values calculate to: RUV2 = 200k, RUV1 = 43.5k (6) FINAL CIRCUIT The final circuit is shown in Figure 4. The circuit was tested, and the resulting performance is shown in Figure 6 and Figure 5. (5) Capacitor C6 is added to filter noise and ripple, which may be present on the VIN line. Where the resistor values are known, the threshold voltages and hysteresis are calculated from the following: 30141718 FIGURE 4. LM34923 Example Circuit 15 www.national.com LM34923 30141724 FIGURE 5. Efficiency vs. Load Current and VIN 30141728 FIGURE 6. Efficiency vs. VIN LOW OUTPUT RIPPLE CONFIGURATIONS For applications where low output ripple is required, the following options can be used to reduce or nearly eliminate the ripple. a) Reduced ripple configuration: In Figure 7, Cff is added across RFB2 to AC-couple the ripple at VOUT directly to the FB pin. This allows the ripple at VOUT to be reduced to a minimum of 25 mVp-p by reducing R3, since the ripple at VOUT is not attenuated by the feedback resistors. The minimum value for Cff is determined from: www.national.com where tON(max) is the maximum on-time, which occurs at the minimum input voltage. The next larger standard value capacitor should be used for Cff. 16 30141721 FIGURE 7. Reduced Ripple Configuration b) Minimum ripple configuration: If the application requires a lower value of ripple (<10 mVp-p), the circuit of Figure 8 can be used. R3 is removed, and the resulting output ripple voltage is determined by the inductor's ripple current and C2's characteristics. RA and CA are chosen to generate a sawtooth waveform at their junction, and that voltage is ACcoupled to the FB pin via CB. To determine the values for RA, CA and CB, use the following procedure: 30141723 FIGURE 9. Alternate Minimum Output Ripple PC Board Layout The LM34923 regulation, over-voltage, and current limit comparators are very fast, and respond to short duration noise pulses. Layout considerations are therefore critical for optimum performance. The layout must be as neat and compact as possible, and all of the components must be as close as possible to the associated pins. The two major current loops have currents which switch very fast, and so the loops should be as small as possible to minimize conducted and radiated EMI. The first loop is formed by C1, through VIN to the SW pin, L1, C2, and back to C1. The second loop is formed by L1, C2, D1, and back to L1. Since a current equal to the load current switches between these two loops with each transition from on-time to off-time and back to on-time, it is imperative that the ground end of C1 have a short and direct connection to D1's anode, without going through vias or a lengthy route. The power dissipation in the LM34923 can be approximated by determining the total conversion loss (PIN - POUT), and then subtracting the power losses in D1, and in the inductor. The power loss in the diode is approximately: Calculate VA = VOUT - (VSW x (1 - (VOUT/VIN(min)))) where VSW is the absolute value of the voltage at the SW pin during the off-time. If a Schottky diode is used for the flyback function, the off-time voltage is in the range of 0.5V to 1V, depending on the specific diode used, and the maximum load current. VA is the DC voltage at the RA/CA junction, and is used in the next equation. - Calculate RA x CA = (VIN(min) - VA) x tON/V where tON is the maximum on-time (at minimum input voltage), and V is the desired ripple amplitude at the RA/CA junction (typically 40-50 mV). RA and CA are then chosen from standard value components to satisfy the above product. Typically CA is 1000 pF to 5000 pF, and RA is 10 k to 300 k. CB is then chosen large compared to CA, typically 0.1 F. PD1 = IOUT x VF x (1-D) where VF is the diode's forward voltage drop, and D is the ontime duty cycle. PL1 = IOUT2 x RL x 1.1 where RL is the inductor's DC resistance, and the 1.1 factor is an approximation for the AC losses. If it is expected that the internal dissipation of the LM34923 will produce excessive junction temperatures during normal operation, good use of the PC board's ground plane can help to dissipate heat. Additionally the use of wide PC board traces, where possible, can help conduct heat away from the IC. Judicious positioning of the PC board within the end product, along with the use of any available air flow (forced or natural convection) can help reduce the junction temperature. 30141722 FIGURE 8. Minimum Output Ripple Using Ripple Injection c) Alternate minimum ripple configuration: The circuit in Figure 9 is the same as that in the Block Diagram, except the output voltage is taken from the junction of R3 and C2. The ripple at VOUT is determined by the inductor's ripple current 17 www.national.com LM34923 and C2's characteristics. However, R3 slightly degrades the load regulation. This circuit may be suitable if the load current is fairly constant. 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