LMH6611MKENOPB - NATIONAL SEMICONDUCTOR

LMH6555

LMH6555 Low Distortion 1.2 GHz Differential Driver

Literature Number: SNOSAJ1C

LMH6555

March 30, 2011

Low Distortion 1.2 GHz Differential Driver

General Description

The LMH6555 is an ultra high speed differential line driver

with 53 dB SFDR at 750 MHz. The LMH6555 features a fixed

gain of 13.7 dB. An input to the device allows the output com-

mon mode voltage to be set independent of the input common

mode voltage in order to simplify the interface to high speed

differential input ADCs. A unique architecture allows the de-

vice to operate as a fully differential driver or as a single-

ended to differential converter.

The outstanding linearity and drive capability (100Ω differen-

tial load) of this device are a perfect match for driving high

speed analog-to-digital converters. When combined with the

ADC081000/ ADC081500 (single or dual ADC), the LMH6555

forms an excellent 8-bit data acquisition system with analog

bandwidths exceeding 750 MHz.

The LMH6555 is offered in a space saving 16-pin LLP pack-

age.

Features

Typical values unless otherwise specified.

■−3 dB bandwidth (VOUT = 0.80 VPP) 1.2 GHz

■±0.5 dB gain flatness (VOUT = 0.80 VPP) 330 MHz

■Slew rate 1300 V/μs

■2nd/3rd Harmonics (750 MHz) −53/−54 dBc

■Fixed gain 13.7 dB

■Supply current 120 mA

■Single supply operation 3.3V ±10%

■Adjustable common-mode output voltage

Applications

■Differential ADC driver

■National Semiconductor ADC081500/ ADC081000

(single or dual) driver

■Single ended to differential converter

■Intermediate frequency (IF) amplifier

■Communication receivers

■Oscilloscope front end

Typical Application

20127704

Single Ended to Differential Conversion

LMH6555 Low Distortion 1.2 GHz Differential Driver

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

ESD Tolerance (Note 5)

Human Body Model 2000V

Machine Model 200V

VS4.2V

Output Short Circuit Duration

(one pin to ground) Infinite

Common Mode Input Voltage −0.4V to 3V

Maximum Junction Temperature +150°C

Storage Temperature Range −65°C to +150°C

Soldering Information

Infrared or Convection (20 sec.) 235°C

Wave Soldering (10 sec.) 260°C

Operating Ratings (Note 1)

Temperature Range (Note 4) −40°C to +85°C

Supply Voltage Range +3.3V ±10%

Package Thermal Resistance (θJA)(Note 4)

16-Pin LLP 65°C/W

3.3V Electrical Characteristics (Note 2)

Unless otherwise specified, all limits are guaranteed for TA= 25°C, VCM_REF = 1.2V, both inputs tied to 0.3V through 50Ω

(RS1 & RS2) each (Note 11), VS = 3.3V, RL = 100Ω differential, VOUT = 0.8 VPP. See the Definition of Terms and Specification section

for definition of terms used throughout the datasheet. Boldface limits apply at the temperature extremes.

Symbol Parameter Conditions Min

(Note 8)

Typ

(Note 7)

Max

(Note 8)

Units

AC/DC Performance

SSBW −3 dB Bandwidth VOUT = 0.25 VPP 1200 MHz

LSBW VOUT = 0.8 VPP 1200

Peak Peaking VOUT = 0.8 VPP 1.4 dB

GF_0.1 dB Gain Flatness ±0.1 dB 180 MHz

GF_0.5 dB ±0.5 dB 330

Ph_Delta Phase Delta Output Differential Phase Difference

f ≤ 1.2 GHz

< ±0.8 deg

Lin_Ph Linear Phase Deviation Each Output

f ≤ 2 GHz

< ±30 deg

GD Group Delay Each Output

f ≤ 2 GHz

0.75 ns

P_1 dB 1 dB Compression 1 GHz 1 VPP

TRS/TRL Rise/ Fall Time VOUT = 0.2 VPP Each Output 320 pS

OS Overshoot VOUT = 0.2 VPP Each Output 14 %

SR Slew Rate 0.8V Step, 10% to 90%,(Note 6) 1300 V/µs

tsSettling Time ±1% 2.2 ns

AV_DIFF Insertion Gain (|S21|) 13.2

13.1

13.7 14.0

14.1 dB

TC AV_DIFF Temperature Coefficient of

Insertion Gain

−0.9 mdB/°C

ΔAV_DIFF1 Insertion Gain Variation with

VCM_REF

VCM_REF Input Varied from 0.95V to

1.45, VOUT = 0.8 VPP

−0.04 ±0.50

±0.58 dB

ΔAV_DIFF2 Insertion Gain Variation with

VI_CM

−0.3 ≤ VI_CM ≤ 2.0V ±0.03 ±0.48

±0.55

Distortion And Noise Response

HD2_L 2nd Harmonic Distortion 250 MHz (Note 12) −60

dBcHD2_M 500 MHz (Note 12) −62

HD2_H 750 MHz (Note 12) −53

HD3_L 3rd Harmonic Distortion 250 MHz (Note 12) −67

dBcHD3_M 500 MHz (Note 12) −61

HD3_H 750 MHz (Note 12) −54

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LMH6555

Symbol Parameter Conditions Min

(Note 8)

Typ

(Note 7)

Max

(Note 8)

Units

OIP3 Output 3rd Order Intermodulation

Intercept

f = 1 GHz

POUT (Each Tone) ≤ –8.5 dBm

(Note 12, Note 13)

27.5 dBm

OIM3 3rd Order Intermodulation

Distortion

f = 1 GHz

POUT (Each Tone) = −6 dBm

(Note 12, Note 13)

−67 dBc

eno Output Referred Voltage Noise ≥1 MHz 19 nV/

NF Noise Figure Relative to a Differential Input

≥10 MHz

15.0 dB

Input Characteristics

RIN CM Input Resistance Each Input to Ground 45 50 55 Ω

RIN_DIFF Differential Input Resistance Differential 66 78 100 Ω

CIN Input Capacitance Each Input to GND 0.3 pF

CMRR Common Mode Rejection Ratio −0.3 ≤ CMVR ≤ 2.0V 40

68 dB

Output Characteristics

VOOS Output Offset Voltage Differential Mode 15 ±50

±55 mV

TCVOOS Output Offset Voltage

Average Drift

(Note 9) ±100 μV/°C

ROOutput Resistance RT1 and RT2 43 50 53 Ω

BAL_Error_DC Output Gain Balance Error −57 −38

BAL_Error_AC −48

BAL_Error_AC_

Phase

Output Phase Balance Error f = 750 MHz,

VOUT+ - VOUT− Phase

±0.6 deg

|ΔVO_CM/ΔVI_CM|Output Common Mode Gain DC −26 −22

−21 dB

VCM_REF Characteristics

VOS_CM Output CM Offset Voltage VOS_CM = VO_CM – VCM_REF −4 ±60

±85 mV

TC_VOS_CM CM Offset Voltage Temp

Coefficient

−0.2 mV/°C

IB_CM VCM_REF Bias Current 0.95V ≤ VCM_REF ≤ 1.45V (Note 10) −25 ±390

±415 μA

RIN_CM VCM_REF Input Resistance 3.5 5.8 kΩ

Gain_VCM_REF VCM_REF Input Gain to Output ΔVO_CM/ΔVCM_REF 0.97 0.99 1.00 V/V

Power Supply

ISSupply Current RS1 & RS2 Open (Note 3) 120 150

156 mA

PSRR Differential Power Supply

Rejection Ratio

DC, ΔVS = ±0.3V, ΔVOUT/ΔVS−27

−25

−44 dB

PSRR_CM Common Mode PSRR DC, ΔVS = ±0.3V, ΔVO_CM/ΔVS−29

−27

−39 dB

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LMH6555

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

intended to be functional, but specific performance is not guaranteed. For guaranteed specifications, see the Electrical Characteristics tables.

Note 2: Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating

of the device such that TJ = TA. No guarantee of parametric performance is indicated in the electrical tables under conditions of internal self-heating where TJ >

TA.

Note 3: Total supply current is affected by the input voltages connected through RS1 and RS2. Supply current tested with input removed.

Note 4: The maximum power dissipation is a function of TJ(MAX), θJA and TA. The maximum allowable power dissipation at any ambient temperature is

PD= (TJ(MAX) — TA)/ θJA. All numbers apply for package soldered directly into a 2 layer PC board with zero air flow. Package should be soldered unto a 6.8

mm2 copper area as shown in the “recommended land pattern” shown in the package drawing.

Note 5: Human Body Model, applicable std. MIL-STD-883, Method 3015.7. Machine Model, applicable std. JESD22-A115-A (ESD MM std. of JEDEC)

Field-Induced Charge-Device Model, applicable std. JESD22-C101-C (ESD FICDM std. of JEDEC).

Note 6: Slew Rate is the average of the rising and falling edges.

Note 7: Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary over time and will

also depend on the application and configuration. The typical values are not tested and are not guaranteed on shipped production material.

Note 8: Limits are 100% production tested at 25°C. Limits over the operating temperature range are guaranteed through correlation using Statistical Quality

Control (SQC) methods.

Note 9: Drift determined by dividing the change in parameter at temperature extremes by the total temperature change.

Note 10: Positive current is current flowing into the device.

Note 11: Quiescent device common mode input voltage is 0.3V.

Note 12: Distortion data taken under single ended input condition.

Note 13: 0 dBm = 894 mVPP across 100Ω differential load

Ordering Information

Package Part Number Package Marking Transport Media NSC Drawing

16-Pin LLP

LMH6555SQ

L6555SQ

1k Units Tape and Reel

SQA16ALMH6555SQE 250 Units Tape and Reel

LMH6555SQX 4.5k Units Tape and Reel

Connection Diagram

16-Pin LLP

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LMH6555

Definition of Terms and Specifications (Alphabetical Order)

Unless otherwise specified, VCM_REF = 1.2V

1. AV_CM (dB) Change in the differential output voltage (ΔVOUT ) with respect to the change in input common mode

voltage (ΔVI_CM)

2. AV_DIFF (dB) Insertion gain from a single ended 50Ω (or 100Ω differential) source to the differential output (ΔVOUT)

3. ΔAV_DIFF (dB) Variation in insertion gain (AV_DIFF)

4. BAL_ERR_DC &

BAL_ERR_AC

5. CM Common Mode

6. CMRR (dB) Common Mode rejection defined as: AV_DIFF (dB) - AV_CM (dB)

7. CMVR (V) Range of input common mode voltage (VI_CM)

8. Gain_VCM_REF (V/V) Variation in output common mode voltage (ΔVO_CM) with respect to change in VCM_REF input (ΔVCM_REF)

with maximum differential output

9. PSRR (dB) Differential output change (ΔVOUT) with respect to the power supply voltage change (ΔVS) with nominal

differential output

10. PSRR_CM (dB) Output common mode voltage change (ΔVO_CM) with respect to the change in the power supply voltage

(ΔVS)

11. RIN (Ω) Single ended input impedance to ground

12. RIN_DIFF (Ω) Differential input impedance

13. RL (Ω) Differential output load

14. RO (Ω) Device output impedance equivalent to RT1 & RT2

15. RS1, RS2 (Ω) Source impedance to VIN+ and VIN− respectively

16. RT1, RT2 (Ω) Output impedance looking into each output

17. VCM_REF (V) Device input pin which controls output common mode

18. ΔVCM_REF (V) Change in the VCM_REF input

19. VI_CM (V) DC average of the inputs (VIN+, VIN−) or the common mode signal at those same input pins

20. ΔVI_CM (V) Variation in input common mode voltage (VI_CM)

21. VIN+, VIN− (V) Device input pin voltages

22. ΔVIN (V) Terminated (50Ω for single ended and 100Ω for differential) generator voltage

23. VO_CM (V) Output common mode voltage (DC average of VOUT+ and VOUT−)

24. ΔVO_CM (V) Variation in output common mode voltage (VO_CM)

25. Balance Error. Measure of the output swing balance of VOUT+ and VOUT−, as reflected on the output

common mode voltage (VO_CM), relative to the differential output swing (VOUT). Calculated as output

common mode voltage change (ΔVO_CM) divided into the output differential voltage change (ΔVOUT which

is nominally around 800 mVPP)

26.

27. VOOS (V) DC Offset Voltage. Differential output voltage measured with both inputs grounded through 50Ω

28. VOS_CM (V) Difference between the output common mode voltage (VO_CM) and the voltage on the VCM_REF input, for

the allowable VCM_REF range

29. VOUT (V) Differential Output Voltage (VOUT+ - VOUT−) (Corrected for DC offset (VOOS))

30. ΔVOUT (V) Change in the differential output voltage (Corrected for DC offset (VOOS))

31. VOUT+, VOUT− (V) Device output pin voltages

32. VS (V) Supply Voltage (V+ - V−)

33. ΔVS (V) Change in VCC supply voltage

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LMH6555

Typical Performance Characteristics Unless otherwise specified, RS1 = RS2 = 50Ω, VS = 3.3V,

RL = 100Ω differential, VOUT = 0.8 VPP. See the Definition of Terms and Conditions section for definition of terms used throughout

the datasheet.

Frequency Response

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±0.5 dB Gain Flatness

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Linear Phase Deviation & Group Delay

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Bal_Error vs. Frequency

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−1 dB Compression vs. Frequency

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Step Response (VOUT+)

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LMH6555

Step Response Settling Time

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Harmonic Distortion vs. Frequency

20127742

3rd Order Intermodulation Distortion

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AV_DIFF & RIN_DIFF vs. VI_CM

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Insertion Gain Distribution

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Insertion Gain Variation vs. Input Amplitude

20127774

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LMH6555

PSRR & PSRR_CM vs. Frequency

20127756

CMRR vs. VI_CM

20127775

CMRR vs. Frequency

20127754

Noise Density & Noise Figure

20127755

S_Parameters vs. Frequency

20127763

Differential Output Offset Variation for

3 Representative Units

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LMH6555

Common Mode Offset Voltage Variation vs. VCM_REF

20127767

Supply Current vs. Temperature

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LMH6555

Application Information

See the Definition of Terms and Conditions section for defi-

nition of terms used.

GENERAL

The LMH6555 consists of three individual amplifiers: The

VOUT+ driver, VOUT− driver, and the common mode amplifier.

Being a differential amplifier, the LMH6555 will not respond

to the common mode input (as long as it is within its input

common mode range) and instead the output common mode

is forced by the built-in common mode amplifier with

VCM_REF as its input. As shown, in Figure 1 below, the VCMO

output of most differential high speed ADC’s is tied to the

VCM_REF input of the LMH6555 for direct output common

mode control. In some cases, the output drive capability of the

ADC VCMO output may need an external buffer, as shown, to

increase its current capability in order to drive the VCM_REF pin.

The LMH6555 Electrical Characteristics table shows the gain

(Gain_VCM_REF) and the offset (VOS_CM) from the VCM_REF to

the device output common mode.

20127704

FIGURE 1. Single Ended to Differential Conversion

The single ended input and output impedances of the

LMH6555 I/O pins are close to 50Ω as specified in the Elec-

trical Characteristics table (RIN and RO). With differential input

drive, the differential input impedance (RIN_DIFF) is close to

78Ω.

The device nominal input common mode voltage (VI_CM) is

close to 0.3V when RS1 and RS2 of Figure 1 are open. Thus,

the input source will experience a DC current with 0V input.

Because of this, the differential output offset voltage is influ-

enced by the matching between RS1 and RS2. So, in a single

ended input condition, if the signal source is AC coupled to

one input, the undriven input needs to also be AC coupled in

order to cancel the output offset voltage (VOOS).

In applications where low output offset is required, it is pos-

sible to inject some current to the appropriate input

(VIN+ or VIN−) as an effective method of trimming the output

offset voltage of the LMH6555. This is explained later in this

document. The nominal value of RS1 and RS2 will also affect

the insertion gain (AV_DIFF). The LMH6555 can also be used

with the input AC coupled through equal valued DC blocking

capacitors (C) in series with VIN+ and VIN−. In this case, the

coupling capacitors need to be large enough to not block the

low frequency content. The lower cutoff frequency will be

1/(πREQC)Hz with REQ= RS1+ RS2 + RIN_DIFF where RIN_DIFF

≈ 78Ω.

The single ended output impedance of the LMH6555 is 50Ω.

The LMH6555 Electrical Characteristics shows the device

performance with 100Ω differential output load, as would be

the case if a device such as the ADC081000/ ADC081500

(single/ dual ADC) were being driven.

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LMH6555

CIRCUIT ANALYSIS

Figure 2 shows the block diagram of the LMH6555.

20127708

RG1 = RG2 = RG = 39Ω

RE1 = RE2 = RE = 25Ω

RF1 = RF2 = RF = 430Ω

ICQ1 = ICQ2 = 12.6 mA

FIGURE 2. Block Diagram

The differential input stage consists of cross-coupled com-

mon base bipolar NPN stages, Q1 and Q2. These stages give

the device its differential input characteristic. The internal loop

gain from Vx and Vy internal nodes (Q1 and Q2 emitters) to

the output is large, such that these nodes act as a virtual

ground. The cross-coupling will ensure that these nodes are

at the same voltage as long as the amplifier is operating within

its normal range. Output common mode voltage is enforced

through the action of “ACM” which servos the output common

mode to the “VCM_REF” input voltage.

The discussion that follows, provides the formulas need-

ed to analyze single ended and differential input applica-

tions. For a more detailed explanation including deriva-

tions, please see the Appendix at the end of the

datasheet.

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LMH6555

SINGLE-ENDED INPUT

The following is the procedure for determining the device op-

erating conditions for single ended input applications. This

example will use the schematic shown in Figure 3.

20127710

FIGURE 3. Single-Ended Input Drive

1. Determine the driven input’s (VIN+ or VIN−) swing

knowing that each input common mode impedance to

ground (RIN) is 50Ω:

VIN+ (or VIN−) = VIN · RIN/(RIN + RS)

For Figure 3

VIN+ = 0.3 VPP · 50/(50+50) = 0.15 VPP

2. Calculate VOUT knowing the Insertion Gain (AV_DIFF):

VOUT = (VIN/2) · AV_DIFF

AV_DIFF = 2 · RF/ (2RS + RIN_DIFF)

where RF = 430Ω & RIN_DIFF = 78Ω

For Figure 3

RS = 50Ω → AV_DIFF = 4.83 V/V

VOUT = (0.3 VPP/2) · 4.83 V/V= 724.5 mVPP

3. Determine the peak-to-peak differential current

(IIN_DIFF) through the device’s differential input

impedance (RIN_DIFF) which would result in the VOUT

calculated in step 2:

IIN_DIFF = VOUT/ RF

For Figure 3

IIN_DIFF = 724.5 mVPP/ 430Ω = 1.685 mAPP

4. Determine the swing across the input terminals

(VIN_DIFF) which would give rise to the IIN_DIFF calculated

in step 3 above.

VIN_DIFF = IIN_DIFF · RIN_DIFF

For Figure 3

VIN_DIFF = 1.685 mAPP · 78Ω = 131.4 mVPP

5. Calculate the undriven input’s swing, based on VIN_DIFF

determined in step 4 and VIN+ calculated in step 1:

VIN− = VIN+ - VIN_DIFF

For Figure 3

VIN− = 150 mVPP - 131.4 mVPP = 18.6 mVPP

6. Determine the DC average of the two inputs (VI_CM) by

using the following expression:

VI_CM = 12.6 mA · RE · RS / (RS + RG + RE)

where RE = 25Ω & RG = 39Ω (both internal

to the LMH6555)

For Figure 3

RS = 50Ω → VI_CM = 15.75 / (RS + 64)

VI_CM = 15.75/ (50+64) = 138.2 mV

The values determined with the procedure outlined here are

shown in Figure 4.

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FIGURE 4. Input Voltage for Figure 3 Schematic

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LMH6555

DIFFERENTIAL INPUT

The following is the procedure for determining the device op-

erating conditions for differential input applications using the

Figure 5 schematic as an example.

20127720

Assuming transformer secondary, VIN, of 300 mVPP

FIGURE 5. Differential Input Drive

1. Calculate the swing across the input terminals

(VIN_DIFF) by considering the voltage division from the

differential source (VIN) to the LMH6555 input terminals

with differential input impedance RIN_DIFF:

VIN_DIFF = VIN · RIN_DIFF/ (2RS + RIN_DIFF)

For Figure 5

VIN_DIFF = 300 mVPP · 78 / (100 + 78) = 131.5 mVPP

2. Calculate each input pin swing to be ½ the swing

determined in step 1:

VIN+ = VIN− = VIN_DIFF/ 2

For Figure 5

VIN+ = VIN− = 131.5 mVPP/ 2 = 65.7 mVPP

3. Determine the DC average of the two inputs (VI_CM) by

using the following expression:

VI_CM = 12.6 mA · RE · RS / (RS + RG + RE)

where RE = 25Ω & RG = 39Ω (both internal

to the LMH6555)

For Figure 5

RS = 50Ω → VI_CM = 15.75 / (RS+ 64)

VI_CM = 15.75/ (50+64) = 138.2 mV

4. Calculate VOUT knowing the Insertion Gain (AV_DIFF):

VOUT = (VIN · / 2) · AV_DIFF

AV_DIFF = 2 · RF/ (2RS + RIN_DIFF)

where RF= 430Ω & RIN_DIFF = 78Ω

For Figure 5

RS = 50Ω → AV_DIFF = 4.83 V/V

VOUT = (0.3 VPP/2) · 4.83 V/V= 724.5 mVPP

The values determined with the procedure outlined here are

shown in Figure 6.

20127765

FIGURE 6. Input Voltage for Figure 5 Schematic

SOURCE IMPEDANCE(S) AND THEIR EFFECT ON GAIN

AND OFFSET

The source impedances RS1 and RS2, as shown in Figure 3

or Figure 5, affect gain and output offset. The datasheet tables

and typical performance graphs are generated with equal val-

ued source impedances RS1 and RS2, unless otherwise spec-

ified. Any mismatch between the values of these two

impedances would alter the gain and offset voltage.

OUTPUT OFFSET CONTROL AND ADJUSTMENT

There are applications which require that the LMH6555 dif-

ferential output voltage be set by the user. An example of such

an application is a unipolar signal which is converted to a dif-

ferential output by the LMH6555. In order to utilize the full

scale range of the ADC input, it is beneficial to shift the

LMH6555 outputs to the limits of the ADC analog input range

under minimal signal condition. That is, one LMH6555 output

is shifted close to the negative limit of the ADC analog input

and the other close to the positive limit of the ADC analog

input. Then, under maximum signal condition, with proper

gain, the full scale range of the ADC input can be traversed

and the ADC input dynamic range is properly utilized. If this

forced offset were not imposed, the ADC output codes would

be reduced to half of what the ADC is capable of producing,

resulting in a significant reduction in ENOB. The choice of the

direction of this shift is determined by the polarity of the ex-

pected signal.

Another scenario where it may be necessary to shift the

LMH6555 output offset voltage is in applications where it is

necessary to improve the specified Output Offset Voltage

(differential mode), “VOOS”. Some ADC’s, including the

ADC081000/ ADC081500 (and their dual counterparts), have

internal registers to correct for the driver’s (LMH6555) VOOS.

If the LMH6555 VOOS rating exceeds the maximum value al-

lowed into this register, then shifting the output is required for

maximum ADC performance.

It is possible to affect output offset voltage by manipulating

the value of one input resistance relative to the other (e.g.

RS1 relative to RS2 or vice versa). However, this will also alter

the gain. Assuming that the source is applied to the VIN+ side

through RS1, Figure 7(A) shows the effect of varying RS1 on

the overall gain and output offset voltage. Figure 7(B) shows

the same effects but this time for when the undriven side

impedance, RS2, is varied.

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LMH6555

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20127731

FIGURE 7. Gain & Output Offset Voltage vs. Source

Impedance Shift for Single Ended Input Drive

As can be seen in Figure 7, the source impedance of the input

side being driven has a bigger effect on gain than the undriven

source impedance. RS1 and RS2 affect the output offset in op-

posite directions. Manipulating the value of RS2 for offset

control has another advantage over doing the same to RS1

and that is the signal input termination is not affected by it.

This is especially important in applications where the signal

is applied to the LMH6555 through a transmission line which

needs to be terminated in its characteristic impedance for

minimum reflection.

For reference, Figure 8 shows the effect of source impedance

misbalance on overall gain and output offset voltage with dif-

ferential input drive.

20127732

FIGURE 8. Gain & Output Offset Voltage vs. Source

Impedance Shift for Differential Input Drive

It is possible to manipulate output offset with little or no effect

on source resistance balance, gain, and, cable termination.

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(a)

20127752

(b)

FIGURE 9. Differential Output Shift Circuits

RX, shown in Figure 9(a) and Figure 9(b), injects current into

the input to achieve the required output shift. For a positive

shift, positive current would need to be injected into the VIN+

terminal (Figure 9(a)) and for a negative shift, to the VIN− ter-

minal (Figure 9(b)). Figure 10 shows the effect of RX on the

output with VX = 3.3V or 5V, and RS1 = RS2 = 50Ω.

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LMH6555

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FIGURE 10. LMH6555 Differential Output Shift Due to RX

in Figure 9

To shift the LMH6555 differential output negative by about

100 mV, referring to the plot in Figure 10, RX would be chosen

to be around 3.9 kΩ in the schematic of Figure 9(b) (using

VX = VS = 3.3V).

In applications where VIN has a built-in non-zero offset volt-

age, or when RS1 and RS2 are not 50Ω, the Figure 10 plot

cannot be used to estimate the required value for RX.

Consider the case of a more general offset correction appli-

cation, shown in Figure 11(a), where RS1 = RS2 = 75Ω and

VIN has a built-in offset of −50 mV. It is necessary to shift the

differential output offset voltage of the LMH6555 to 0 mV.

Figure 11(b) is the Thevenin equivalent of the circuit in Figure

11(a) assuming RX >> RS2.

20127735

(a)

20127753

(b)

FIGURE 11. Offset Correction Example (RS = 75Ω)

From the gain expression in Equation 4 (see Appendix) (but

with opposite polarity because VTH is applied to VIN− instead):

(1)

The expression derived for VOUT in Equation 1 can be set

equal to zero to solve for RX resulting in RX = 4.95 kΩ. If the

differential output offset voltage, VOOS, is also known, VOUT

could be set to a value equal to –VOOS. For example, if the

VOOS for the particular LMH6555 is +30 mV, then the following

nulls the differential output:

(2)

RX >> RS2 confirming the assumption made in the derivation.

Note that Equation 2, which is derived based on the configu-

ration in Figure 9(b), will yield a real solution for RX if and only

if:

(3)

where VIN_OFFSET is the source offset shown as −50 mV in

Figure 11(a).

If Equation 3 were not satisfied, then Figure 9(a) offset cor-

rection, where RX is tied to the VIN+ side, should be employed

instead.

Alternatively, replace the VX and RX combination with a dis-

crete current source or current sink. Because of a current

source’s high output impedance, there will be less gain im-

balance. However, a current source might have a relatively

large output capacitance which could degrade high frequency

performance.

INTERFACE DESIGN EXAMPLE

As shown in Figure 12 below, the LMH6555 can be used to

interface an open collector output device (U1) to a high speed

ADC. In this application, the LMH6555 performs the task of

amplifying and driving the 100Ω differential input impedance

of the ADC.

20127706

VCM_REF buffer not shown

FIGURE 12. Differential Amplification and ADC Drive

15 www.national.com

LMH6555

For applications similar to the one shown in Figure 12, the

following conditions should be maintained:

1. The LMH6555 differential output voltage has to comply

with the ADC full scale voltage (800 mVPP in this case).

2. The LMH6555 input Common Mode Voltage Range is

observed. “CMVR”, as specified in the Electrical

Characteristics table, is to be between −0.3V and 2.0V

for the specified CMRR.

3. U1 collector voltage swing must to be observed so that

the U1 output transistors do not saturate. The expected

operating range of these output transistors is defined by

the specifications and operating conditions of U1.

Consider a numerical example (RL refers to RL1 & RL2, RS

refers to RS1 & RS2).

Assume:

VCC = 10V, U1 peak-to-peak collector current

(IPP) = 15 mAPP with 10 mA quiescent (IcQ), and minimum op-

erational U1 collector voltage = 6V.

Here are the series of steps to take in order to carry out this

design:

a. Select the RL value which allows compliance with the U1

collector voltage (6V in this case) with 1V extra as

margin because of LMH6555 loading.

RL = [10 - (6+1)] V / (10+ 7.5) mA = 171Ω

Choose 169Ω, 1% resistors for RL

b. Find the value of RS to get the proper swing at the output

(800 mVPP). To do so, convert the input stage into its

Norton equivalent as shown in Figure 13.

20127748

FIGURE 13. Norton Equivalent of the Input Circuitry Tied

to Q1 within the LMH6555 in Figure 12

IN = IN (common mode) + IN (differential)

IN (common mode) = (VCC – IcQ * RL) / (RL + RS + RG)

IN (differential) = IPP * RL / (RL + RS + RG)

The entirety of the Norton source differential component

will flow through the feedback resistors within the

LMH6555 and generate an output. Therefore:

IN (differential) * RF = 800 mVPP

→ RS = (RL* IPP * RF/ 0.8) – RG – RL where RF = 430Ω,

RG = 39Ω (RF and RG are internal LMH6555

resistances).

So, in this case:

RS = (169 * 15 mAPP * 430/ 0.8) – 39 – 169 = 1154Ω

Choose 1.15 kΩ, 1% resistors for RS.

c. With RL and RS defined, ensure that the U1 collector

voltage(s) minimum is not violated due to the loading

effect of the LMH6555 through RS. Also, it is important

to ensure that the LMH6555's CMVR is also not violated.

The “Vx” node voltage within the LMH6555 (see Figure

13) would need to be calculated. Use the Common

Mode component of the Norton equivalent source from

above, and write the KCL at the Vx node as follows:

Vx / RE + Vx / RN = 12.6 mA + IN (common mode); with

RE = 25Ω.

Vx / RE + Vx / RN = 12.6 mA + (VCC – IcQ RL )/ (RL + RS

+ RG)

→Vx = 0.4595V

With Vx calculated, both the input voltage range (high

and low) and the low end of the U1 collector voltage

(VC) can be derived to be within the acceptable range.

If necessary, steps “a” through “c” would have to be

repeated to readjust these values.

VC = VX RL / RN + IN (RS + RG)

IN_High = 7.05 mA, IN_Low = 5.19 mA (based on the

values derived)

→VC_High = 0.4595 * 169 / 1358 + 7.05 mA (1150 + 39)

= 8.44V

→VC_Low = 0.4595 * 169 / 1358 + 5.19 mA (1150 + 39)

= 6.22V

VIN = VX (RN – RG) / RN + IN RG

→VIN_High = 0.4595 * (1358- 39) / 1358 + 7.05 mA * 39

= 0.721V

→VIN_Low = 0.4595 * (1358- 39) / 1358 + 5.19 mA * 39

= 0.649V

Figure 14 shows the complete solution using the values de-

rived above, with the node voltages marked on the schematic

for reference.

www.national.com 16

LMH6555

20127749

FIGURE 14. Implementation #1 of Figure 12

Design Example

It is important to note that the matching of the resistors on

either input side of the LMH6555 (RS1 to RS2 and RL1 to RL2)

is very important for output offset voltage and gain balance.

This is particularly true with values of RS higher than the nom-

inal 50Ω. Therefore, in this example, 1% or better resistor

values are specified.

If the U1 collector voltage turns out to be too low due to the

loading of the LMH6555, lower RL. Lower values of RL result

in lower RS which in turn increases the LMH6555's VI_CM be-

cause of increased pull up action towards VCC. The upper limit

on VI_CM is 2V. Figure 15 shows the 2nd implementation of this

same application with lowered values of RL and RS. Notice

that the lower end of U1’s collector voltage and the upper end

of LMH6555’s VI_CM have both increased compared to the

1st implementation.

20127750

FIGURE 15. Implementation #2 of Figure 12 Design

Example

An alternative would be to AC couple the LMH6555 inputs.

With this approach, the design steps would be very similar to

the ones outlined except that there would be no common

mode interaction between the LMH6555 and U1 and this re-

sults in fewer design constraints:

Vx / RE = 12.6 mA → Vx = 0.3150V

For the component values shown in Figure 15 use:

VC_High = VCC – RL (IcQ + IPP / 2 - IN (differential) /2)

VC_Low = VCC – RL (IcQ - IPP / 2 + IN (differential) /2)

IN (differential) = IPP * RL / (RL + RS + RG) = 1.88 mA (based

on the values used.)

→VC_High = 10 – 80.6 (10 + 15 / 2 − 1.88 /2) mA = 8.67V

→VC_Low = 10 – 80.6 (10 − 15 / 2 + 1.88 /2) mA = 9.72V

VIN = VX ± RG. IN (differential) /2

→VIN_High = 0.3150 + 39 * 1.88 mA /2 = 0.3517V

→VIN_Low = 0.3150 - 39 * 1.88 mA /2 = 0.2783V

Figure 16 shows the AC coupled implementation of the

Figure 15 schematic along with the node voltages marked to

demonstrate the reduced VI_CM of the LMH6555 and the in-

crease in the U1 collector voltage minimum.

20127751

FIGURE 16. AC Coupled Version of Figure 15

Note that the lower cut-off frequency is:

f_cut-off = 1 / (πReqCS) where Req = RS1+ RS2 + RIN_DIFF

where RIN_DIFF ≈ 78Ω

So, for the component values shown (CS = 0.01 μF and RS1

= RS2 = 523Ω):

f_cut-off = 28.2 kHz

DATA ACQUISITION APPLICATIONS

Figure 17 shows the LMH6555 used as the differential driver

to the National Semiconductor ADC081500 running at 1.5G

samples/second.

20127704

FIGURE 17. Schematic of the LMH6555 Interfaced to the

ADC081500

17 www.national.com

LMH6555

In the schematic of Figure 17, the LMH6555 converts a single

ended input into a differential output for direct interface to the

ADC's 100Ω differential input. An alternative approach to us-

ing the LMH6555 for this purpose, would have been to use a

balun transformer, as shown in Figure 18.

20127777

FIGURE 18. Single Ended to Differential Conversion

(AC only) with a Balun Transformer

In the circuit of Figure 18, the ADC will see a 100Ω differential

driver which will swing the required 800 mVPP when VIN is 1.6

VPP. The source (VIN) will see an overall impedance of

200Ω for the frequency range that the transformer is specified

to operate. Note that with this scheme, the signal to the ADC

must be AC coupled, because of the transformer’s minimum

operating frequency which would prevent DC coupling. For

the transformer specified, the lower operating frequency is

around 4.5 MHz and the input high pass filter’s −3 dB band-

width is around 340 kHz for the values shown (or (1/πREQC)

Hz where REQ = 200Ω).

Table 1 compares the LMH6555 solution (Figure 17) vs. that

of the balun transformer coupling (Figure 18) for various cat-

egories.

TABLE 1. ADC Input Coupling Schemes Compared

Category

Preferred Solution

LMH6555 Balun

Transformer

Lower Power Consumption ✓

Lower Distortion ✓

Wider Dynamic Range ✓

DC Coupling & Broadband

Applications

✓

Highest Gain & Phase

Balance

✓

Input/ Output Broadband

Impedance Matching

(Highest Return Loss)

✓

Additional Gain ✓

ADC Input Protection against

Overdrive

✓

Highest SNR ✓

Ability to Control Gain

Flatness

✓

(see below)

GAIN FLATNESS

In applications where the full 1.2 GHz bandwidth of the

LMH6555 is not necessary, it is possible to improve the gain

flatness frequency at the expense of bandwidth. Figure 19

shows CO placed across the LMH6555 output terminals to

reduce the frequency response gain peaking and thereby to

increase the ±0.5 dB gain flatness frequency.

20127776

FIGURE 19. Increasing ±0.5 dB Gain Flatness using

External Output Capacitance, CO

Figures 20, 21 and Figure 22 show the FFT analysis results

with the setup shown in Figure 17.

20127743

FIGURE 20. LMH6555 FFT Result When Used as the

Differential Driver to ADC081500

20127744

FIGURE 21. LMH6555 FFT Result When Used as the

Differential Driver to ADC081500

(Lower Fs/2 Region Magnified)

www.national.com 18

LMH6555

20127745

FIGURE 22. LMH6555 FFT Result When Used as the

Differential Driver to ADC081500

(Upper Fs/2 Region Magnified)

Figures 20, 21, and Figure 22 information summary:

• Fundamental Test

Frequency

744 MHz

• LMH6555 Output 0.8 VPP

• Sampling Rate: 1.5G samples/ second

• 2nd Harmonic −59 dBc @ ∼ 12 MHz or

|1.5 GHz*1– 744 MHz*2|

• 3rd Harmonic −57 dBc @ ∼ 732 MHz or

|1.5 GHz*1- 744 MHz *3|

• 4th Harmonic −71 dBc @ ∼ 24 MHz or

|1.5 GHz*2 – 744 MHz *4|

• 5th Harmonic −68 dBc @ ∼ 720 MHz or

|1.5 GHz*2- 744 MHz*5|

• 6th Harmonic −68 dBc @ ∼ 36 MHz or

|1.5 GHz*3- 744 MHz*6|

• THD −51.8 dBc

• SNR 43.4 dB

• Spurious Free Dynamic

Range (SFDR): 57 dB

• SINAD 42.8 dB

• ENOB 6.8 bits

The LMH6555 is capable of driving a variety of National Semi-

conductor Analog to Digital Converters. This is shown in

Table 2, which offers a complete list of possible signal path

ADC+ Amplifier combinations. The use of the LMH6555 to

drive an ADC is determined by the application and the desired

sampling process (Nyquist operation, sub-sampling or over-

sampling). See application note (AN-236) for more details on

the sampling processes and application note (AN-1393) for

details on “Using High Speed Differential Amplifiers to Drive

ADCs”. For more information regarding a particular ADC, re-

fer to the particular ADC datasheet for details.

TABLE 2. Differential Input ADC’s Compatible with the

LMH6555 Driver

ADC Part Number Resolution

(bits)

Single/

Dual

Speed

(MSPS)

ADC08D500 8 S 500

ADC081000 8 S 1000

ADC08D1000 8 D 1000

ADC08D1020 8 D 1000

ADC081500 8 S 1500

ADC08D1500 8 D 1500

ADC08D1520 8 D 1500

ADC083000 8 S 3000

ADC08B3000 8 S 3000

EXPOSED PAD LLP PACKAGE

The LMH6555 is in a thermally enhanced package. The ex-

posed pad (device bottom) is connected to the GND pins. It

is recommended, but not necessary, that the exposed pad be

connected to the supply ground plane. The thermal dissipa-

tion of the device is largely dependent on the connection of

this pad. The exposed pad should be attached to as much

copper on the circuit board as possible, preferably external

copper. However, it is very important to maintain good high

speed layout practices when designing a system board.

Here is a link to more information on the National 16-pin LLP

package:

http://www.national.com/packaging/folders/sqa16a.html

EVALUATION BOARD

National Semiconductor suggests the following evaluation

board as a guide for high frequency layout and as an aid in

device testing and characterization.

Device Package Evaluation Board

Ordering ID

LMH6555 16-Pin LLP LMH6555EVAL

The evaluation board can be ordered when a device sample

request is placed with National Semiconductor.

19 www.national.com

LMH6555

Appendix

Here is a more detailed analysis of the LMH6555, including

the derivation of the expressions used throughout the Appli-

cation Information.

INPUT STAGE

Because of the input stage cross-coupling, if the instanta-

neous values of the input node voltages (VIN+ and VIN−) and

current values are required, use the circuit of Figure 23 as the

equivalent input stage for each input (VIN+ and VIN−).

20127709

FIGURE 23. Equivalent Input Stage

Using this simplified circuit, one can assume a constant col-

lector current, to simplify the analysis. This is a valid approx-

imation as the large open loop gain of the device will keep the

two collector currents relatively constant. First derive Q1 and

Q2 emitter voltages. From there, derive the voltages at VIN+

and VIN−.

With the component values shown, it is possible to analyze

the input circuits of Figure 23 in order to determine Q1 and

Q2 emitter voltages. This will result in a first order estimate of

Q1 and Q2 emitter voltages. Since Q1 and Q2 emitters are

cross-coupled, the voltages derived would have to be equal.

With the action of the common mode amplifier, “ACM”, shown

in Figure 2, these two emitters will be equalized. So, one other

iteration can be performed whereby both emitters are set to

be equal to the average of the 1st derived emitter voltages.

Using this new emitter voltage, one could recalculate VIN+ and

VIN− voltages. The values derived in this fashion will be within

±10% of the measured values.

Single Ended Input Analysis

Here is an actual example to further clarify the procedure.

Consider the case where the LMH6555 is used as a single

ended to differential converter shown in Figure 24.

20127710

FIGURE 24. Single Ended Input Drive

The first task would be to derive the internal transistor emitter

voltages based on the schematic of Figure 23 (assuming that

there is no interaction between the stages.) Here is the deriva-

tion of VX and Vy:

VX varies with VIN+ (0.213V with negative VIN swing and

0.279V with positive.) The values derived above assume that

the two halves of the input circuit do not interact with each

other. They do through the common mode amplifier and the

input stage cross-coupling. Vx and Vy are equal to the average

of Vy with either end of the swing of VX. This is calculated

below along with the derivation of VIN+ and VIN− based on this

new average emitter voltage (the average of VX and Vy.)

With 0.3 VPP VIN, VIN+ experiences 150 mVPP (213 mV - 63.2

mV) of swing and VIN− will swing by about 18.6 mVPP in the

process (147 mV – 129 mV). The input voltages are shown in

Figure 25.

20127764

FIGURE 25. Input Voltages for Figure 24 Schematic

www.national.com 20

LMH6555

Using the calculated swing on VIN+ with known VIN, one can

estimate the input impedance, RIN as follows:

Differential Input Analysis

Assume that the LMH6555 is used as a differential amplifier

with a transformer with its Center Tap at ground as shown in

Figure 26:

20127720

Assuming transformer secondary, VIN, of 300 mVPP

FIGURE 26. Differential Input Drive

The input voltages (VIN+ and VIN−) can be derived using the

technique explained previously. Assuming no transformer

output and referring to the schematic of Figure 23:

The peak VIN+ and VIN− voltages can be determined using the

transformer output voltage. Assuming there is 0.3 VPP of sig-

nal across the transformer secondary, ½ of that, or 0.15 VPP

(±75 mV peak), would appear at each input side (V1 or V2 in

Figure 26). Here is the derivation of the LMH6555 input

terminal’s peak voltages.

When V1 swings positive, V2 will go negative by the same

value, and vice versa. Therefore, the values derived above

for Vx can be used to determine the average emitter voltage,

as described earlier:

With the transformer voltage of 0.3 VPP, each input (VIN+ and

VIN−) swings from 105.3 mV to 171.0 mV or about 65.7

mVPP. The input voltages are shown in Figure 27.

20127765

FIGURE 27. Input Voltages for Figure 26 Schematic

Knowing the device input terminal voltages, one can estimate

the differential input impedance as follows:

This is comparable to RIN_DIFF found in the Electrical Charac-

teristic table.

OUTPUT STAGE AND GAIN ANALYSIS

Differential gain is determined by the differential current flow

through the feedback resistors RF1 and RF2 as shown in Fig-

ure 2. Current through RF1 (or RF2) sets the VOUT− (or VOUT+)

swing. The nominal value of these resistors is close to 430Ω.

The LMH6555 output stage consists of two bipolar common

emitter amplifiers with built in output resistances, RT1 and

RT2, of 50Ω, as shown in Figure 28.

20127725

FIGURE 28. Output Stage Including External Load RL

21 www.national.com

LMH6555

With an output differential load, RL, of 100Ω, half the differ-

ential swing between the output emitters appears at the

LMH6555 output terminals as VOUT.

With good matching between the input source impedances,

RS1 and RS2 shown in Figure 24 and Figure 26, it is possible

to infer the gain and output swing by inspection. The differ-

ential input impedance of the LMH6555, RIN_DIFF, is close to

78Ω.

In differential input drive applications, there is a balanced

swing across the input terminals of the LMH6555, VIN+ and

VIN−. So, by using the RIN_DIFF value, one determines the dif-

ferential current flow through the input terminals and from that

the output swing and gain.

(4)

For the special case where RS1 = RS2 = RS = 50Ω we have:

The following is the expression for the Insertion Gain,

AV_DIFF:

The expressions above apply equally to the single ended in-

put drive case as well, as long as RS1 = RS2 = 50Ω. For the

case of the single ended input drive:

This is comparable to AV_DIFF found in the Electrical Charac-

teristic table.

www.national.com 22

LMH6555

Physical Dimensions inches (millimeters) unless otherwise noted

16-Pin LLP

NS Package Number SQA16A

23 www.national.com

LMH6555

Notes

LMH6555 Low Distortion 1.2 GHz Differential Driver

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