QT118H-ISG - Quantum Research Group

APPLICATIONS -



Elevator buttons



Toys & games



Access systems



Pointing devices



Appliance control



Security systems



Light switches



Industrial panels

The QT118H charge-transfer (“QT’”) touch sensor is a self-contained digital IC capable of detecting near-proximity or touch. It will

project a sense field through almost any dielectric, like glass, plastic, stone, ceramic, and wood. It can also turn small metal-bearing

objects into intrinsic sensors, making them respond to proximity or touch. This capability coupled with an ability to self calibrate

continuously can lead to entirely new product concepts.

The device is designed specifically for human interfaces, like control panels, appliances, toys, lighting controls, or anywhere a

mechanical switch or button may be found; it may also be used for some material sensing and control applications provided that the

presence duration of objects does not exceed the recalibration timeout interval.

A piezo element can also be connected to create a feedback click sound.

The IC requires only a common inexpensive capacitor in order to function. Average power consumption is under 20µA in most

applications, allowing battery operation.

The QT118H employs digital signal processing techniques pioneered by Quantum, designed to make it survive real-world

challenges, such as ‘stuck sensor’ conditions and signal drift. Sensitivity is digitally determined for the highest possible stability. No

external active components are required for operation.

The device includes several user-selectable built in features. One, toggle mode, permits on/off touch control, for example for light

switch replacement. Another makes the sensor output a pulse instead of a DC level, which allows the device to 'talk' over the power

rail, permitting a simple 2-wire twisted-pair interface. Quantum’s unique HeartBeat™ signal is also included, allowing a host

controller to continuously monitor the health of the device.

By using the charge transfer principle, the IC delivers a level of performance clearly superior to older technologies in a highly

cost-effective package.

R1.08 / 0405

QProx™ QT118H

HARGE

-T

RANSFER

OUCH

ENSOR

Sns2

Vss

Sns1

GainOpt2

Opt1

Out

Vdd 1

QT118H

-QT118H-ISG-40

C to +85

QT118H-DG-0

C to +70

8-PIN DIPSOICT

AVAILABLE OPTIONS



Less expensive than many mechanical switches



Projects a ‘touch button’ through any dielectric

100% autocal for life - no adjustments required

No active external components



Piezo sounder direct drive for ‘tactile’ click feedback



LED drive for visual feedback

2.5 ~ 5V single supply operation

10µ

µµ

µA at 2.5V - very low power drain

Toggle mode for on/off control (via option pins)

10s or 60s auto-recalibration timeout (via option pins)

Pulse output mode (via option pins)



Gain settings in 3 discrete levels



Simple 2-wire operation possible



HeartBeat™ health indicator on output

Pb-Free package

1 - OVERVIEW

The QT118H is a digital burst mode charge-transfer (QT)

sensor designed specifically for touch controls; it includes all

hardware and signal processing functions necessary to

provide stable sensing under a wide variety of changing

conditions. Only a few low cost, non-critical discrete external

parts are required for operation.

Figure 1-1 shows the basic QT118H circuit using the device,

with a conventional output drive and power supply

connections. Figure 1-2 shows a second configuration using

a common power/signal rail which can be a long twisted pair

from a controller; this configuration uses the built-in pulse

mode to transmit the output state to the host controller.

1.1 BASIC OPERATION

The QT118H employs short, low duty cycle bursts of QT

cycles to acquire capacitance. Burst mode permits power

consumption in the low microamp range, dramatically

reduces RF emissions, lowers susceptibility to EMI, and yet

permits excellent response time. Internally the signals are

digitally processed to reject impulse noise, using a

'consensus' filter which requires four consecutive

confirmations of a detection before the output is activated.

The QT switches and charge measurement hardware

functions are all internal to the QT118H (Figure 1-3). A

single-slope switched capacitor ADC includes both the

required QT charge and transfer switches in a configuration

that provides direct ADC conversion. The sensitivity depends

on the values of Cs, Cx, and to a smaller degree, Vdd. Vdd is

used as the charge reference voltage.

Higher values of Cs increase gain; higher values of Cx load

reduce it. The value of Cs can thus be increased to allow

larger values of Cx to be tolerated (Figures 4-1 and 4-2, page

10).

Piezo sounder drive: The QT118H can drive a piezo

sounder after a detection for feedback. The piezo sounder

replaces or augments the Cs capacitor; this works since

piezo sounders are also capacitors, albeit with a large

thermal drift coefficient. If C

piezo

is in the proper range, no

additional capacitor. If C

piezo

is too small, it can simply be

‘topped up’ with a ceramic capacitor in parallel. The QT118H

drives a ~4kHz signal across SNS1 and SNS2 to make the

piezo (if installed) sound a short tone for 75ms immediately

after detection, to act as an audible confirmation.

Option pins allow the selection or alteration of several special

features and sensitivity.

1.2 ELECTRODE DRIVE

The internal ADC treats Cs as a floating transfer capacitor; as

a direct result, the sense electrode can in theory be

connected to either SNS1 or SNS2 with no performance

difference. However, the noise immunity of the device is

improved by connecting the electrode to SNS2, preferably via

a series resistor Re (Figure 1-1) to roll off higher harmonic

frequencies, both outbound and inbound.

In order to reduce power consumption and to assist in

discharging Cs between acquisition bursts, a 470K series

resistor Rs should always be connected across Cs (Figure

1-1).

The rule Cs >> Cx must be observed for proper operation.

Normally Cx is on the order of 10pF or so, while Cs might be

10nF (10,000pF), or a ratio of about 1:1000.

It is important to minimize the amount of unnecessary stray

capacitance Cx, for example by minimizing trace lengths and

widths and backing off adjacent ground traces and planes so

as keep gain high for a given value of Cs, and to allow for a

larger sensing electrode size if so desired.

The PCB traces, wiring, and any components associated with

or in contact with SNS1 and SNS2 will become touch

sensitive and should be treated with caution to limit the touch

area to the desired location.

2 QT118H R1.08 / 0405

Figure 1-1 Standard mode options

SENSING

ELECTRODE

Cs Rs

2nF - 500nF

+2.5 ~ +5

OUT

OPT1

OPT2

GAIN

SNS1

SNS2

Vss

Vdd

OUTPUT = DC

TIMEOUT = 10 Secs

TOGGLE = OFF

GAIN = H IGH

Figure 1-2 2-wire operation, self-powered

10µF

1N4148

n-ch Mosfet

CMOS

LOGIC

3.5 - 5.5V

1K Twisted

pair

OUT

OPT1

OPT2

GAIN

SNS1

SNS2

Vss

Vdd

SENSING

ELECTRODE

1.3 ELECTRODE DESIGN

1.3.1 E

LECTRODE

EOMETRY

AND

IZE

There is no restriction on the shape of

the electrode; in most cases common

sense and a little experimentation can

result in a good electrode design. The

QT118H will operate equally well with

long, thin electrodes as with round or

square ones; even random shapes are

acceptable. The electrode can also be

a 3-dimensional surface or object.

Sensitivity is related to electrode

surface area, orientation with respect

to the object being sensed, object

composition, and the ground coupling

quality of both the sensor circuit and

the sensed object.

1.3.2 K

IRCHOFF

’

URRENT

Like all capacitance sensors, the

QT118H relies on Kirchoff’s Current

Law (Figure 1-5) to detect the change

in capacitance of the electrode. This law as applied to

capacitive sensing requires that the sensor’s field current

must complete a loop, returning back to its source in order for

capacitance to be sensed. Although most designers relate to

Kirchoff’s law with regard to hardwired circuits, it applies

equally to capacitive field flows. By implication it requires that

the signal ground and the target object must both be coupled

together in some manner for a capacitive sensor to operate

properly. Note that there is no need to provide actual

hardwired ground connections; capacitive coupling to ground

(Cx1) is always sufficient, even if the coupling might seem

very tenuous. For example, powering the sensor via an

isolated transformer will provide ample ground coupling,

since there is capacitance between the windings and/or the

transformer core, and from the power wiring itself directly to

'local earth'. Even when battery powered, just the physical

size of the PCB and the object into which the electronics is

embedded will generally be enough to couple a few

picofarads back to local earth.

1.3.3 V

IRTUAL

APACITIVE

ROUNDS

When detecting human contact (e.g. a fingertip), grounding

of the person is never required. The human body naturally

has several hundred picofarads of ‘free space’ capacitance to

the local environment (Cx3 in Figure 1-4), which is more than

two orders of magnitude greater than that required to create

a return path to the QT118H via earth. The QT118H's PCB

however can be physically quite small, so there may be little

‘free space’ coupling (Cx1 in Figure 1-4) between it and the

environment to complete the return path. If the QT118H

circuit ground cannot be earth grounded by wire, for example

via the supply connections, then a ‘virtual capacitive ground’

may be required to increase return coupling.

A ‘virtual capacitive ground’ can be created by connecting the

QT118H’s own circuit ground to:

- A nearby piece of metal or metallized housing;

- A floating conductive ground plane;

- Another electronic device (to which its might be

connected already).

Free-floating ground planes such as metal foils should

maximize exposed surface area in a flat plane if possible. A

square of metal foil will have little effect if it is rolled up or

crumpled into a ball. Virtual ground planes are more effective

and can be made smaller if they are physically bonded to

other surfaces, for example a wall or floor.

‘Ground’ as applied to capacitive fields can also mean power

wiring or signal lines. The capacitive sensor, being an AC

device, needs only an AC ground return.

1.3.5 S

ENSITIVITY

DJUSTMENT

1.3.5.1 Gain Pin

The QT118H can be set for one of 3 gain levels using option

pin 5 (Table 1-1). This sensitivity change is made by altering

the internal numerical threshold level required for a detection.

Note that sensitivity is also a function of other things: like the

values of Cs and Cx, electrode size, shape, and orientation,

the composition and aspect of the object to be sensed, the

thickness and composition of any overlaying panel material,

and the degree of ground coupling of both sensor and object.

The Gain input should never be connected to a pullup or

pulldown resistor or tied to anything other than SNS1 or

SNS2, or left unconnected (for high gain setting).

3 QT118H R1.08 / 0405

Figure 1-3 Internal Switching & Timing

SNS2

SNS1

ELECTRODE

Single-Slope 14-bit

Switched Capacitor ADC

C ha rge

Amp

Burst Controller

Result

Do ne

Start

Figure 1-4 Kirchoff's Current Law

Sense Electrode

Surrounding environment

SENSOR

1.3.5.2 Changing Cs, Cx

The values of Cs and Cx have a dramatic effect on

sensitivity, and Cs can be easily increased in value to

improve gain. Sensitivity is directly proportional to Cs and

inversely proportional to Cx:

k$C

Where ‘k’ depends on a variety of factors including the gain

pin setting (see prior section), Vdd, etc.

Sensitivity plots are shown in Figures 4-1 and 4-2, page 10.

1.3.5.3 Electrode / Panel Adjustments

Sensitivity can often be increased by using a bigger

electrode, or reducing overlying panel thickness. Increasing

electrode size can have a diminishing effect on gain, as the

attendant higher values of Cx will start to reduce sensor gain.

Also, increasing the electrode's surface area will not

substantially increase touch sensitivity if its diameter is

already much larger in surface area than the object being

detected.

The panel or other intervening material can be made thinner,

but again there are diminishing rewards for doing so. Panel

material can also be changed to one having a higher

dielectric constant, which will help propagate the field through

to the front. Locally adding some conductive material to the

panel (conductive materials essentially have an infinite

dielectric constant) will also help; for example, adding carbon

or metal fibers to a plastic panel will greatly increase frontal

field strength, even if the fiber density is too low to make the

plastic bulk-conductive.

1.3.5.3 Ground Planes

Grounds around and under the electrode and its SNS trace

will cause high Cx loading and destroy gain. The possible

signal-to-noise ratio benefits of ground area are more than

negated by the decreased gain from the circuit, and so

ground areas around electrodes are discouraged. Keep

ground, power, and other signals traces away from the

electrodes and SNS wiring

2 - QT118H SPECIFICS

2.1 SIGNAL PROCESSING

The QT118H digitally processes all signals using

a number of algorithms pioneered by Quantum.

The algorithms are specifically designed to

provide for high survivability in the face of all

kinds of adverse environmental changes.

2.1.1 D

RIFT

OMPENSATION

LGORITHM

Signal drift can occur because of changes in Cx

and Cs over time. It is crucial that drift be

compensated for, otherwise false detections,

non-detections, and sensitivity shifts will follow.

Drift compensation (Figure 2-1) is performed by

making the reference level track the raw signal at

a slow rate, but only while there is no detection in effect. The

rate of adjustment must be performed slowly, otherwise

legitimate detections could be ignored. The QT118H drift

compensates using a slew-rate limited change to the

reference level; the threshold and hysteresis values are

slaved to this reference.

Once an object is sensed, the drift compensation mechanism

ceases since the signal is legitimately high, and therefore

should not cause the reference level to change.

The QT118H's drift compensation is 'asymmetric': the

reference level drift-compensates in one direction faster than

it does in the other. Specifically, it compensates faster for

decreasing signals than for increasing signals. Increasing

signals should not be compensated for quickly, since an

approaching finger could be compensated for partially or

entirely before even touching the sense pad. However, an

obstruction over the sense pad, for which the sensor has

already made full allowance for, could suddenly be removed

leaving the sensor with an artificially elevated reference level

and thus become insensitive to touch. In this latter case, the

sensor will compensate for the object's removal very quickly,

usually in only a few seconds.

2.1.2 T

HRESHOLD

AND

YSTERESIS

The internal signal threshold level can be set to one of three

settings (Table 1-1). These are fixed with respect to the

internal reference level, which in turn moves in accordance

with the drift compensation mechanism.

The QT118H employs a hysteresis dropout below the

threshold level of 17% of the delta between the reference and

threshold levels.

2.1.3 M

-D

URATION

If an object or material obstructs the sense pad the signal

may rise enough to create a detection, preventing further

operation. To prevent this, the sensor includes a timer which

monitors detections. If a detection exceeds the timer setting,

the timer causes the sensor to perform a full recalibration.

This is known as the Max On-Duration feature.

After the Max On-Duration interval, the sensor will once again

function normally, even if partially or fully obstructed, to the

best of its ability given electrode conditions. There are two

timeout durations available via strap option: 10 and 60

seconds.

2.1.4 D

ETECTION

NTEGRATOR

It is desirable to suppress detections generated by electrical

noise or from quick brushes with an object. To accomplish

this, the QT118H incorporates a detect integration counter

4 QT118H R1.08 / 0405

Pin 7

Low

Pin 6

Medium

Leave open

High

Tie Pin 5 to:Gain

Table 1-1 Gain Strap Options

Figure 2-1 Drift Compensation

Threshold

Signal Hysteresis

Reference

Output

that increments with each detection until a limit is reached,

after which the output is activated. If no detection is sensed

prior to the final count, the counter is reset immediately to

zero. The required count is 4.

The Detection Integrator can also be viewed as a 'consensus'

filter, that requires four detections in four successive bursts to

create an output. As the basic burst spacing is 95ms, if this

spacing was maintained through 4 consecutive bursts the

sensor would be very slow to respond. In the QT118H, after

an initial detection is sensed, the remaining three bursts are

spaced only about 2ms apart, so that the slowest reaction

time possible is the fastest possible.

2.1.5 F

ORCED

ENSOR

ECALIBRATION

The QT118H has no recalibration pin; a forced recalibration

is accomplished only when the device is powered up.

However, the supply drain is so low it is a simple matter to

treat the entire IC as a controllable load; simply driving the

QT118H's Vdd pin directly from another logic gate or a

microprocessor port (Figure 2-2) will serve as both power and

'forced recal'. The source resistance of most CMOS gates

and microprocessors is low enough to provide direct power

without any problems. Almost any CMOS logic gate can

directly power the QT118H.

A 0.01uF minimum bypass capacitor close to the device is

essential; without it the device can break into high frequency

oscillation.

Option strap configurations are read by the QT118H only on

powerup. Configurations can only be changed by powering

the QT118H down and back up again; a microcontroller can

directly alter most of the configurations and cycle power to

put them in effect.

2.2 OUTPUT FEATURES

The QT118H is designed for maximum flexibility and can

accommodate most popular sensing requirements. These

are selectable using strap options on pins OPT1 and OPT2.

All options are shown in Table 2-1.

OPT1 and OPT2 should never be left floating. If they are

floated, the device will draw excess power and the options

will not be properly read on powerup. Intentionally, there are

no pullup resistors on these lines, since pullup resistors add

to power drain if the pin(s) are tied low.

2.2.1 DC M

ODE

UTPUT

The output of the device can respond in a ‘DC mode’, where

the output is active-high upon detection. The output will

remain active for the duration of the detection, or until the

Max On-Duration expires, whichever occurs first. If the latter

occurs first, the sensor performs a full recalibration and the

output becomes inactive until the next detection.

In this mode, two nominal Max On-Duration timeouts are

available: 10 and 60 seconds.

2.2.2 T

OGGLE

ODE

UTPUT

This makes the sensor respond in an on/off mode like a flip

flop. It is most useful for controlling power loads, for example

in kitchen appliances, power tools, light switches, etc.

Max On-Duration in Toggle mode is fixed at 10 seconds.

When a timeout occurs, the sensor recalibrates but leaves

the output state unchanged.

10sVddGnd

Pulse

10sGndGnd

Toggle

60sGndVdd

DC Out

10sVddVdd

DC Out

Max On-

Duration

Tie

Pin 4 to:

Tie

Pin 3 to:

Table 2-1 Output Mode Strap Options

2.2.3 P

ULSE

ODE

UTPUT

This generates a positive pulse of 95ms duration with every

new detection. It is most useful for 2-wire operation (see

Figure 1-2), but can also be used when bussing together

several devices onto a common output line with the help of

steering diodes or logic gates, in order to control a common

load from several places.

Max On-Duration is fixed at 10 seconds if in Pulse output

mode.

The piezo beeper drive does not operate in Pulse mode.

2.2.4 H

EART

EAT

™ O

UTPUT

The output has a full-time HeartBeat™ ‘health’ indicator

superimposed on it. This operates by taking 'Out' into a

tri-state mode for 350µs once before every QT burst. This

output state can be used to determine that the sensor is

operating properly, or, it can be ignored using one of several

simple methods.

Since Out is normally low, a pullup resistor will create positive

HeartBeat pulses (Figure 2-3) when the sensor is not

detecting an object; when detecting an object, the output will

remain active for the duration of the detection, and no

HeartBeat pulse will be evident.

If the sensor is wired to a microcontroller as shown in Figure

2-4, the controller can reconfigure the load resistor to either

ground or Vcc depending on the output state of the device,

so that the pulses are evident in either state.

Electromechanical devices will usually ignore this short

pulse. The pulse also has too low a duty cycle to visibly

activate LED’s. It can be filtered completely if desired, by

adding an RC timeconstant to filter the output, or if interfacing

directly and only to a high-impedance CMOS input, by doing

nothing or at most adding a small non-critical capacitor from

Out to ground (Figure 2-5).

5 QT118H R1.08 / 0405

Figure 2-2 Powering From a CMOS Port Pin

0. 01 µ F

CMO S

m icrocontroller

OUT

PORT X.m

PORT X.n

Vdd

Vss

QT118

2.2.5 P

IEZO

COUSTIC

RIVE

A piezo drive signal is generated for use with a piezo sounder

immediately after a detection is made; the tone lasts for a

nominal 95ms to create a ‘tactile feedback’ sound.

The sensor drives the piezo using an H-bridge configuration

for the highest possible sound level. The piezo is connected

across pins SNS1 and SNS2 in place of Cs or in addition to a

parallel Cs capacitor. The piezo sounder should be selected

to have a peak acoustic output in the 3.5kHz to 4.5kHz

region.

Since piezo sounders are merely high-K ceramic capacitors,

the sounder will double as the Cs capacitor, and the piezo's

metal disc can even act as the sensing electrode. Piezo

transducer capacitances typically range from 6nF to 30nF in

value; at the lower end of this range an additional capacitor

should be added to bring the total Cs across SNS1 and

SNS2 to at least 10nF, or possibly more if Cx is above 5pF.

Piezo sounders have very high, uncharacterized thermal

coefficients and should not be used if fast temperature

swings are anticipated, especially at high gains. They are

also generally unstable at high gains; even if the total value

of Cs is largely from an added capacitor the piezo can cause

periodic false detections.

The burst acquisition process induces a small but audible

voltage step across the piezo resonator, which occurs when

SNS1 and SNS2 rapidly discharge residual voltage stored on

the resonator. The resulting slight clicking sound can be

greatly reduced by placing a 470K resistor Rs in parallel with

the resonator; this acts to slowly discharge the resonator,

attenuating of the harmonic-rich audible step (Figure 2-6).

Note that the piezo drive does not operate in Pulse mode.

2.2.6 O

UTPUT

RIVE

The QT118H’s output is active high and it can source or sink

1mA of non-inductive current.

Care should be taken when the IC and the load are both

powered from the same supply, and the supply is minimally

regulated. The device derives its internal references from the

power supply, and sensitivity shifts can occur with changes in

Vdd, as happens when loads are switched on. This can

induce detection ‘cycling’, whereby an object is detected, the

load is turned on, the supply sags, the detection is no longer

sensed, the load is turned off, the supply rises and the object

is reacquired, ad infinitum. To prevent this occurrence, the

output should only be lightly loaded if the device is operated

from an unregulated supply, e.g. batteries. Detection

‘stiction’, the opposite effect, can occur if a load is shed when

Out is active.

3 - CIRCUIT GUIDELINES

3.1 SAMPLE CAPACITOR

When used for most applications, the charge sampler Cs can

be virtually any plastic film or good quality ceramic capacitor.

The type should be relatively stable in the anticipated

6 QT118H R1.08 / 0405

Figure 2-4

Using a micro to obtain HB pulses in either output state

Figure 2-3

Getting HB pulses with a pullup resistor when not active

Figure 2-5 Eliminating HB Pulses

OUT

OPT1

OPT2

GAIN

SNS1

SNS2

CMOS

100pF

GATE O R

MICRO INPUT

Figure 2-6 Piezo Sounder Circuit

Piezo Sounder

10-30nF

OUT

OPT2

GAIN

SNS2

SNS1

Vss

Vdd

OPT1

SENSING

ELECTRODE

+2.5 ~ +5

temperature range. If fast temperature swings are expected,

especially with higher sensitivities, more stable capacitors be

required, for example PPS film. In most moderate gain

applications (ie in most cases), low-cost X7R types will work

fine.

3.2 ELECTRODE WIRING

See also Section 3.4.

The wiring of the electrode and its connecting trace is

important to achieving high signal levels and low noise.

Certain design rules should be adhered to for best results:

1. Use a ground plane under the IC itself and Cs and Rs

but NOT under Re, or under or closely around the

electrode or its connecting trace. Keep ground away

from these things to reduce stray loading (which will

dramatically reduce sensitivity).

2. Keep Cs, Rs, and Re very close to the IC.

3. Make Re as large as possible. As a test, check to be

sure that an increase of Re by 50% does not appreciably

decrease sensitivity; if it does, reduce Re until the 50%

test increase has a negligible effect on sensitivity.

4. Do not route the sense wire near other ‘live’ traces

containing repetitive switching signals; the trace will pick

up noise from external signals.

3.3 POWER SUPPLY, PCB LAYOUT

The power supply can range from 2.5 to 5.0 volts. At 2.5 volts

current drain averages less than 10µA with Cs = 10nF,

provided a 470K Rs resistor is used (Figure 1-1). Sample Idd

curves are shown in Figure 4-3.

Higher values of Cs will raise current drain. Higher Cx values

can actually decrease power drain. Operation can be from

batteries, but be cautious about loads causing supply droop

(see Output Drive, Section 2.2.6) if the batteries are

unregulated.

As battery voltage sags with use or fluctuates slowly with

temperature, the IC will track and compensate for these

changes automatically with only minor changes in sensitivity.

If the power supply is shared with another electronic system,

care should be taken to assure that the supply is free of

digital spikes, sags, and surges which can adversely affect

the device. The IC will track slow changes in Vdd, but it can

be affected by rapid voltage steps.

if desired, the supply can be regulated using a conventional

low current regulator, for example CMOS LDO regulators that

have nanoamp quiescent currents. Care should be taken that

the regulator does not have a minimum load specification,

which almost certainly will be violated by the QT118's low

current requirement. Furthermore, some LDO regulators are

unable to provide adequate transient regulation between the

quiescent and acquire states, creating Vdd disturbances that

will interfere with the acquisition process. This can usually be

solved by adding a small extra load from Vdd to ground, such

as 10K ohms, to provide a minimum load on the regulator.

Conventional non-LDO type regulators are usually more

stable than slow, low power CMOS LDO types. Consult the

regulator manufacturer for recommendations.

For proper operation a 100nF (0.1uF) ceramic bypass

capacitor must be used between Vdd and Vss; the bypass

cap should be placed very close to the device’s power pins.

Without this capacitor the part can break into high frequency

oscillation, get physically hot, stop working, or become

damaged.

PCB Cleanliness: All capacitive sensors should be treated

as highly sensitive circuits which can be influenced by stray

conductive leakage paths. QT devices have a basic

resolution in the femtofarad range; in this region, there is no

such thing as ‘no clean flux’. Flux absorbs moisture and

becomes conductive between solder joints, causing signal

drift and resultant false detections or temporary loss of

sensitivity. Conformal coatings can trap existing amounts of

moisture which will then become highly temperature

sensitive.

The designer should strongly consider ultrasonic cleaning as

part of the manufacturing process, and in more extreme

cases, the use of conformal coatings after cleaning and

baking.

3.3.1 S

UPPLY

URRENT

Measuring average power consumption is a challenging task

due to the burst nature of the device’s operation. Even a

good quality RMS DMM will have difficulty tracking the

relatively slow burst rate, and will show erratic readings.

The easiest way to measure Idd is to put a very large

capacitor, such as 2,700µF across the power pins, and put a

220 ohm resistor from there back to the power source.

Measure the voltage across the 220 resistor with a DMM and

compute the current based on Ohm’s law. This circuit will

average out current to provide a much smoother reading.

To reduce the current consumption the most, use high or low

gain pin settings only, the smallest value of Cs possible that

works, and a 470K resistor (Rs) across Cs (Figure 1-1). Rs

acts to help discharge capacitor Cs between bursts, and its

presence substantially reduces power consumption.

3.3.2 ESD P

ROTECTION

In cases where the electrode is placed behind a dielectric

panel, the IC will be protected from direct static discharge.

However even with a panel transients can still flow into the

electrode via induction, or in extreme cases via dielectric

breakdown. Porous materials may allow a spark to tunnel

right through the material. Testing is required to reveal any

problems. The device has diode protection on its terminals

which will absorb and protect the device from most ESD

events; the usefulness of the internal clamping will depending

on the dielectric properties, panel thickness, and rise time of

the ESD transients.

The best method available to suppress ESD and RFI is to

insert a series resistor Re in series with the electrode as

shown in Figure 1-1. The value should be the largest that

does not affect sensing performance. If Re is too high, the

gain of the sensor will decrease.

Because the charge and transfer times of the QT118 are

relatively long (~2µs), the circuit can tolerate a large value of

Re, often more than 10k ohms in most cases.

Diodes or semiconductor transient protection devices or

MOV's on the electrode trace are not advised; these devices

have extremely large amounts of nonlinear parasitic

capacitance which will swamp the capacitance of the

electrode and cause false detections and other forms of

instability. Diodes also act as RF detectors and will cause

serious RF immunity problems.

7 QT118H R1.08 / 0405

3.4 EMC AND RELATED NOISE ISSUES

External AC fields (EMI) due to RF transmitters or electrical

noise sources can cause false detections or unexplained

shifts in sensitivity.

The influence of external fields on the sensor is reduced by

means of the Rseries described in Section 3.2. The Cs

capacitor and Rseries (see Figure 1-1) form a natural

low-pass filter for incoming RF signals; the roll-off frequency

of this network is defined by -

2✜R

series

If for example Cs = 22nF, and Rseries = 10K ohms, the rolloff

frequency to EMI is 723Hz, vastly lower than any credible

external noise source (except for mains frequencies i.e. 50 /

60 Hz). However, Rseries and Cs must both be placed very

close to the body of the IC so that the lead lengths between

them and the IC do not form an unfiltered antenna at very

high frequencies.

PCB layout, grounding, and the structure of the input circuitry

have a great bearing on the success of a design to withstand

electromagnetic fields and be relatively noise-free.

In brief summary, the following design rules should be

adhered to for best ESD and EMC results:

1. Use only SMT components.

2. Keep Cs, Rs, Re and Vdd bypass cap close to the IC.

3. Maximize Re to the limit where sensitivity is not

noticeably affected.

4. Do not place the electrode or its connecting trace near

other traces, or near a ground plane.

5. Do use a ground plane under and around the QT118

itself, back to the regulator and power connector (but not

beyond the Cs capacitor).

6. Do not place an electrode (or its wiring) of one QT

device near the electrode or wiring of another device, to

prevent cross interference.

7. Keep the electrode (and its wiring) away from other

traces carrying AC or switched signals.

8. If there are LEDs or LED wiring near the electrode or its

wiring (ie for backlighting of the key), bypass the LED

wiring to ground on both its ends.

9. Use a voltage regulator just for the QT118 to eliminate

power noise coupling from other switching sources.

Make sure the regulator’s transient load stability provides

for stable voltage just before each burst commences.

For further tips on construction, PCB design, and EMC issues

browse the application notes and faq at www.qprox.com

8 QT118H R1.08 / 0405

4.1 ABSOLUTE MAXIMUM SPECIFICATIONS

Operating temp.............................................................. as designated by suffix

Storage temp..................................................................... -55

C to +125

.................................................................................-0.5 to +6.5V

Max continuous pin current, any control or drive pin.............................................. ±20mA

Short circuit duration to ground, any pin........................................................ infinite

Short circuit duration to V

, any pin........................................................... infinite

Voltage forced onto any pin.................................................. -0.6V to (Vdd + 0.6) Volts

4.2 RECOMMENDED OPERATING CONDITIONS

................................................................................. +2.5 to 5.0V

Short-term supply ripple+noise................................................................ ±5mV

Long-term supply stability.................................................................. ±100mV

Cs value........................................................................... 10nF to 500nF

Cx value.............................................................................. 0 to 100pF

Rs value................................................................................. 470K✡

4.3 AC SPECIFICATIONS

Vdd = 3.0, Cs = 10nF, Rs = 470K, Cx = 10pF, Gain = High, Ta = 20

C, unless otherwise noted.

kHz165Burst frequencyF

µs300Heartbeat pulse widthT

ms75Pulse output width on OutT

ms75Piezo drive durationT

kHz4.

3.6Piezo drive frequencyF

ms129Response timeT

Depends on Cs, Cxms500.5Burst lengthT

@ 5.0V Vdd

@ 3.3V Vdd

Burst spacing intervalT

µs2Charge, transfer durationT

ms550Recalibration timeT

NotesUnitsMaxTypMinDescriptionParameter

4.4 SIGNAL PROCESSING

Vdd = 3.0, Cs = 10nF, Rs = 470K, Cx = 10pF, Gain = High, Ta = 20

C, unless otherwise noted.

3, 4secs6010Post-detection recalibration timer duration (typical min/max)

ms/level75Negative drift compensation rate

ms/level750Positive drift compensation rate

samples4Detect integrator filter length

2%17Hysteresis

1counts6, 12, or 24Threshold differential

NotesUnitsMaxTypMinDescription

Note 1: Pin options

Note 2: Percentage of signal threshold

Note 3: Pin option

Note 4: Cs, Cx dependent

9 QT118H R1.08 / 0405

4.5 DC SPECIFICATIONS

Vdd = 3.0, Cs = 10nF, Rs = 470K, Cx = 10pF, Gain = High, Ta = 20

C Unless otherwise noted.

Note 2fF281,000Sensitivity rangeS

bits1

cquisition resolution

OPT1, OPT2µA±1Input leakage currentI

OUT, 1mA source

Vdd-0.7High output voltage

OUT, 4mA sinkV0.6Low output voltageV

OPT1, OPT2V2.2High input logic levelV

OPT1, OPT2

0.8Low input logic level

Required for proper startupV/s100Supply turn-on slopeV

DDS

@ Vdd = 5.0

@ Vdd = 3.3V

@ Vdd = 2.5V

µA30

Supply currentI

-E suffixV2.95 Guaranteed min VddV

DDL

-I suffix

2.45 Guaranteed min Vdd

DDL

NotesUnitsMaxTypMinDescriptionParameter

10 QT118H R1.08 / 0405

Figure 4-2 - Typical Threshold Sensitivity vs. Cx,

Medium Gain, Selected Values of Cs; Vdd = 3.0

0.01

0.10

1.00

10.00

0 10203040

Cx Load, pF

Detection Threshold, pF

10nF

20nF

50nF

100nF

200nF

500nF

Figure 4-1 - Typical Threshold Sensitivity vs. Cx,

High Gain, at Selected Values of Cs; Vdd = 3.0

0.01

0.10

1.00

10.00

0 10203040

Cx Load, pF

Detection Threshold, p

10nF

20nF

50nF

100nF

200nF

500nF

Figure 4-3 Typical Supply Current Vs Vdd

Rs = 470K, Cx = 10pF, Gain = High

2.5 3 3.5 4 4.5 5

Vdd

Idd, Microamperes

...

Cs = 20nF

Cs = 10nF

4.6 MECHANICAL

0.0150.0080.3810.203Y

0.390.329.9068.128x

BSC0.30.3BSC7.0627.62Aa

0.16-4.064-S1

0.140.123.5563.048S

-0.015-0.381r

0.150.123.813.048R

Typical0.1020.098Typical2.5912.489F

0.0650.0551.6511.397L1

0.0220.0140.5590.355L

-0.01-0.254P

-0.035-0.889Q

BSC0.30.3BSC7.627.62m

Typical0.430.355Typical10.9229.017M

0.3250.38.2557.62A

0.280.247.112 6.096a

NotesMaxMinNotesMaxMin

InchesMillimeters

SYMBOL

Package type: 8pin Dual-In-Line

8º0º8º0ºØ

0.030.2290.7620.381ß

0.010.0070.2490.19e

0.040.021.0160.508E

0.0190.0140.4830.355L

BSC0.050.050BSC1.271.27D

0.010.0040.7620.101h

0.0680.0541.7281.371H

0.1570.153.9883.81Aa

0.2440.2296.1985.816W

0.1960.1894.9794.800M

NotesMaxMinNotesMaxMin

InchesMillimeters

SYMBOL

Package type: 8pin SOIC

5 - ORDERING INFORMATION

QT1 + T + G

or QT118H-IG

SO-8

Pb-Free

-40 - 85CQT118H-ISG

QT118H-G

PDIP

Pb-Free

0 - 70CQT118H-D

MARKINGPACKAGETEMP RANGEPART

11 QT118H R1.08 / 0405

6 - SOIC MARKING DIAGRAMS

VERSION ‘A’

Pin 1 Dimple

Lot code

(last letter varies)

QPROX C

©QT1 T F

0214HB6.G

'G' ending

indicates Pb-free

package

Lot Code

VERSION ‘B’

Pin 1 Dimple

©QPROX

QT118H-IG

0214HB6C

'G' ending

indicates Pb-free

package

Lot Code

12 QT118H R1.08 / 0405

NOTES

13 QT118H R1.08 / 0405

Patented and patents pending

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1 Mitchell Point

Ensign Way, Hamble SO31 4RF

Great Britain

Tel: +44 (0)23 8056 5600 Fax: +44 (0)23 80565600

admin@qprox.com

www.qprox.com

North America

651 Holiday Drive Bldg. 5 / 300

Pittsburgh, PA 15220 USA

Tel: 412-391-7367 Fax: 412-291-1015

This device covered under one or more of the following United States and international patents: 5,730,165, 6,288,707, 6,377,009, 6,452,514,

6,457,355, 6,466,036, 6,535,200. Numerous further patents are pending which may apply to this device or the applications thereof.

The specifications set out in this document are subject to change without notice. All products sold and services supplied by QRG are subject

to our Terms and Conditions of sale and supply of services which are available online at www.qprox.com and are supplied with every order

acknowledgement. QProx, QTouch, QMatrix, QLevel, and QSlide are trademarks of QRG. QRG products are not suitable for medical

(including lifesaving equipment), safety or mission critical applications or other similar purposes. Except as expressly set out in QRG's Terms

and Conditions, no licenses to patents or other intellectual property of QRG (express or implied) are granted by QRG in connection with the

sale of QRG products or provision of QRG services. QRG will not be liable for customer product design and customers are entirely

responsible for their products and applications which incorporate QRG's products.