FAN5018BMTCX - Fairchild Semiconductor

www.fairchildsemi.com

REV. 1.0.0 Jul/15/05

Features

• Improved Noise Immunity

• Improved Load Line Accuracy

• TTL Compatible VID Inputs

• Fully Pin and Function compatible with existing

FAN5018 and ADP3180 Control lers

• Precision Multi-Phase DC-DC Core Voltage Regulation

– ±14mV Output Voltage Accuracy Over Temperature

• Differential Remote Voltage Sensing

• Selectable 2-, 3-, or 4-Phase Operation

• Selectable VRM9 or VRM10 Operation

• Up to 1MHz per Phase Operation (4MHz ripple

Frequency)

• Lossless Inductor Current Sensing for Loadli ne

Compensation

– External Temperature Compensation

• Accurate Loadline Programming (Meets Intel®

VRM/VRD10.x CPU Specifications)

• Accurate Channel-Current Balancing for Therm a l

Optimization and Layout Comp ensation

• Convenient 12V Supply Biasing

• 6-bit Voltage Identification (VID) Input

– .8375V to 1.600V in 12.5mV Steps

– Dynamic VID Capability with Fault-Bl anki ng for

glitch-less Output voltage Changes

• Adjustable Over Current Protection with Programmable

Latch-Off Delay. Latch-Off Function may be Disabled

• Over-Voltage Protection – Internal OVP Crowbar

Protection

• Package: 28L-TSSOP

Applications

• VRM/VRD 9.x and 10.x Computer DC/DC Conv erter

• High-Current, Low-Voltage DC/DC Rail

General Description

The FAN5018B is a multi-phase DC-DC cont roller for

implementing high-current, low -vo ltage, CPU core power

regulation circuits. It is part of a chipset that includes exter-

nal MOSFET drivers and power MOSFETS. The

FAN5018B drives up to four synch ro nous-recti fied buck

channels in parallel. The multi-phase buck converter archi-

tecture uses interleaved switching to multiply ripple fre-

quency by the number of phases and reduce input and output

ripple currents. Lower ripple results in fewer components,

lower component cost, reduced power dissipation, and

smaller board area.

The FAN5018B features a high-bandwidth control loop to

provide optimal response to load transients. The F AN5018B

senses current using lossless techniques: Phase current is

measured through each of the output inductors. This current

information is summed, averaged and used to set the loadline

of the output via programmable "droop." The droop is tem-

perature compensated to achieve precise loadline character-

istics over the entire operating range. Additionally,

individual phase current is measured using the RDS(ON) of

the low-side MOSFETs. This information is used to dyn am-

ically balance/steer per-phase current. The phase currents

are also summed and averaged for over-current detection.

Dynamic-VID technology allows on-the-fly VID changes

with controlled, glitch-less output. Additionally , short-circuit

protection, adjustable current limitin g, over-voltage protec-

tion and power-good circuitry combine to ensure reliable and

safe operation. FAN5018B is specified over the commercial

temperature range of 0°C to +85°C and operates from a sin-

gle +12V supply which simplifi es design. FAN5018B is

available in a 28L-TSSOP package.

Block Diagram

FAN5018B

VIN

Φ1

Φ4

VOU

Φ3

Φ2

FAN5009

FAN5018B

6-Bit VID Controller 2-4 Phase VR10.X Controller

FAN5018BPRODUCT SPECIFICATION

2REV. 1.0.0 Jul/15/05

Pin Assignments

Pin Definitions

Pin Number Pin Name Pin Function Description

1–5 VID [4:0] VID inputs. Determines the output voltage via the internal DAC. These inputs

comply to VRM10/VRD10 specifications for static and dynamic operation. All have

internal pull-ups (1.25V for VRM10 and 2.5V for VRM9) so leaving them open

results in logic high. Leaving VID[4:0] open results in a "No CPU" condition

disabling the PWM outputs.

6 VID5/SEL VID5 Input/DAC Select. Dual function pin that is either the 12.5mV DAC LSB for

VRM10 or selects the VRM9 DAC codes when forced higher than Vtblsel(VRM9)

voltage. The truth table is as follows:

VVID5/SEL held > Vtblsel(VRM9); VRM9 DAC table is selected (See Table 3)

VViD5/SEL < Vtblsel(VRM10); VRM10 DAC table is selected (See Table 2) and

VViD5/SEL pin is used as VID5 input.

7FBRTNFeedback Return. Error Amp and DAC reference point.

8FBFeedback Input. Inverting input for Error Amp this pin is used for external

compensation. This pin can also be used to introduce DC offset voltage to the

output.

9COMPError Amp output. This pin is used for external compensation.

10 PWRGD Power Good output. This is an open-drain output that asserts when the output

voltage is within the specified tolerance. It is expected to be pulled up to an external

voltage rail.

11 EN Enable. Logic signal that enables the controller when logic high.

12 DELAY Soft-start and Current Limit Delay. An external resistor and capacitor sets the

softstart ramp rate and the over-current latch off delay.

13 RT Switching Frequency Adjust. This pin adjusts the output PWM switching

frequency via an external resistor.

14 RAMPADJ PWM Current Ramp Adjust. An external resistor to Vcc will adjust the amplitude of

the internal PWM ramp.

15 ILIMIT Current Limit Adjust. An external resistor sets the current limit threshold for the

regulator circuit. This pin is internally pulled low when EN is low or the UVLO circuit

is active. It is also used to enable the drivers.

DELAY

VID4

VID3

VID2

VID1

VID0

COMP

CSCOMP

PWRGD

CSSUM

VCC

SW2

SW3

PWM3

PWM4

PWM1

GND

FBRTN

ILIMIT

CSREF

RAMPADJ

FAN5018B

TSSOP-28

SW1

PWM2

SW4

VID5/SEL

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 3

16 CSREF Current Sense Reference. Non-Inverting input of the current sense amp. Sense

point for the output voltage used for OVP and PWRGD.

17 CSSUM Current Sense Summing node. Inverting input of the current sense amp.

18 CSCOMP Current Sense Compensation node. Output of the current sense amplifier. This

pin is used, in conjunction with CSSUM to set the output droop compensation and

current loop response.

19 GND Ground. Signal ground for the device.

20–23 SW[4:1] Phase Current Sense/Balance inputs. Phase-to-phase current sense and

balancing inputs. Unused phases should be left open.

24–27 PWM[4:1] PWM outputs. CMOS outputs for driving external gate driver such as the FAN5009.

Unused phases should be connected to Ground.

28 VCC Chip Power. Bias supply for the chip. Connect directly to a +12V supply. Bypass

with a 1µF MLCC capacitor.

Pin Definitions (continued)

Pin Number Pin Name Pin Function Description

FAN5018BPRODUCT SPECIFICATION

4REV. 1.0.0 Jul/15/05

Absolute Maximum Ratings

Absolute maximum ratings are the values beyond which the device may be dam aged or have its useful life impaired. Func-

tional operation under these conditio ns is not im plied.

Thermal Information

Recommended Operating Conditions (See Figure 8)

Note:

1: θJA is defined as 1 oz. copper PCB with 1 in2 pad.

Parameter Min. Max. Units

Supply Voltage: VCC to GND -0.3 +15 V

Voltage on FBRTN pin -0.3 +0.3 V

Voltage on SW1-SW4 (<250ns duration) -5 +25 V

Voltage on SW1-SW4 (>=250ns duration) -0.3 +15 V

Voltage on RAMPADJ, CSSUM -0.3 VCC+0.3 V

Voltage on any other pin -0.3 +5.5 V

Parameter Min. Typ Max. Units

Operating Junction Temperature (TJ)0+150°C

Storage Temperature –65 +150 °C

Lead Soldering Temperature, 10 seconds +300 °C

Vapor Phase, 60 seconds +215 °C

Infrared, 15 seconds +220 °C

Power Dissipation (PD) @ TA = 25°C 2 W

Thermal Resistance (θJA) (See Note 1) 50 °C/W

Parameter Conditions Min. Typ. Max. Units

Supply Voltage VCC VCC to GND 10.2 12 13.8 V

Ambient Operating Temperature 0 +85 °C

Operating Junction Temperature (TJ) 0 +125

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 5

Electrical Specifications

(VCC = 12V, TA = 0°C to +85°C and FBRTN=GND, using circuit in Figure 1, unless otherwise noted.)

The • denotes specifications which apply over the full operating temperature range.

Parameter Symbol Conditions Min. Typ. Max. Units

Error Amplifier

Output Voltage Range VCOMP •0.5 3.5 V

Accuracy VFB Relative to DAC Setting,

referenced to FBRTN,

CSSUM = CSCOMP,

Test Circuit 3

VRM10

VRM9

•

-14

-17

+14

+17

Line Regulation ΔVFB VCC=10V to 14V 0.05 %

Input Bias Current IFB •-17 -15 -13 µA

FBRTN Current IFBRTN •150 180 µA

Output Current IO(ERR) FB forced to VOUT – 3% 300 500 µA

Gain Bandwidth Product GBW COMP = FB ( See Note 2) 20 MHz

DC Gain CCOMP = 10pF ( See Note 2) 77 dB

VID Inputs

Input Low Voltage VIL(VID) VRM10

VRM9

•

0.4

0.8

Input High Voltage VIH(VID) VRM10

VRM9

•

0.8

2.0

Input Current, VID Low IIL(VID) VID(X) = 0V •-30 -20 µA

Input Current, VID High IIH(VID) VID(X) = 1.15V •-2 2 µA

Pull-up Resistance RVID Internal •35 60 115 kΩ

Internal Pull-up Voltage VRM10

VRM9

•

1.0

2.2

1.15

2.4

1.26

2.6

VID Transition Delay

Time2

VID Code Change to FB Change •400 ns

“No CPU” Detection

Turn-off Delay Time2

VID Code Change to 11111X to PWM

going low

•400 ns

VID Table Select Vtblsel To select VRM9 table

To select VRM10 table (becomes VID5)

•

3.5

Oscillator

Frequency fOSC •200 4000 kHz

Frequency Variation fPHASE TA = +25°C, RT = 250kΩ, 4-Phase

TA = +25°C, RT = 115kΩ, 4-Phase

TA = +25°C, RT = 75kΩ, 4-Phase

•155 200

400

600

245 kHz

kHz

Output Voltage VRT RT = 100kΩ to GND •1.9 2.0 2.1 V

RAMPADJ Pin Accuracy VRAMPADJ VRAMPADJ = Vdac = +2K • (Vin–Vdac)/

(Rr+2k) @ 20µA

•-50 +50 mV

RAMPADJ Input Current IRAMPADJ Current into RAMPADJ pin 0 100 µA

Current Sense Amplifier

Offset Voltage VOS(CSA) CSSUM–CSREF, Test Circuit 1 •-3.0 +3.0 mV

Input Bias Current IBIAS(CSA) •-50 +50 nA

Gain Bandwidth Product GBW COMP = FB ( See Note 2) 10 MHz

Notes:

1. All limits at operating temperature extremes are guaranteed by design, characterization and statistical quality control

2. Guaranteed by design – NOT tested in production.

FAN5018BPRODUCT SPECIFICATION

6REV. 1.0.0 Jul/15/05

DC Gain ( See Note 2) 77 dB

Input Common Mode

Range

CSSUM and CSREF •03V

Positioning Accuracy ΔVFB COMP = FB, Test Circuit 2 •-84 -80 -76 mV

Output Voltage Range ICSCOMP = ±100µA •0.1 3.3 V

Output Current IO(CSA) FB forced to VOUT – 3% Source/Sink 300 375 µA

Current Balance Circuit

Input Operating Range VSW(X)CM •-600 +200 mV

Input Resistance RSW(X) SW(X) = 0V •20 30 40 kΩ

Input Current ISW(X) SW(X) = 0V •-10 -7 -4 µA

Input Current Matching ΔISW(X) SW(X) = 0V •-5 +5 %

Current Limit Comparator

ILIMIT Output Voltage

Normal Mode

In Shutdown

VILIMIT(NM)

VILIMIT(SD)

RILIMIT = 250kΩ

IILIMIT = -100µA

•

2.9 3 3.1

400

Output Current, Normal

Mode

IILIMIT(NM) RILIMIT = 250kΩ12 µA

Maximum Output Current VIILIMIT = 3V •60 µA

Current Limit Threshold

Voltage

VCL VCSREF–VCSCOMP

, RILIMIT = 250kΩ•105 125 150 mV

Current Limit Setting Ratio VCL/IILIMIT 10.4 mV/µA

Latch-off Delay Threshold VDELAY In Current Limit •1.7 1.8 1.9 V

Latch-off Delay Time tDELAY RDELAY = 250kΩ, CDELAY = 4.7nF 600 µs

Soft Start

Output Current,

Soft start Mode

IDELAY(SS) During Start-up, DELAY < 2.8V •-25 -20 -15 µA

Soft Start Delay Time TDELAY(SS) RDELAY = 250kΩ, CDELAY = 4.7nF,

VID = 011111 (1.475V)

400 µs

Enable Input

Input Low Voltage VIL(EN) •0.4 V

Input High Voltage VIH(EN) •0.8 V

Input Current, EN Low IIL(EN) EN = 0V •-1 1 µA

Input Current, EN High IIH(EN) EN = 1.25V •10 25 µA

Power Good Comparator

Under voltage Threshold VPWRGD(UV) Relative to DAC Output •-325 -250 -200 mV

Over voltage Threshold VPWRGD(OV) Relative to DAC Output •180 300 400 mV

Output Low Voltage VOL(PWRGD) IPWRGD(SINK) = 4mA •225 400 mV

Electrical Specifications (continued)

(VCC = 12V, TA = 0°C to +85°C and FBRTN=GND, using circuit in Figure 1, unless otherwise noted.)

The • denotes specifications which apply over the full operating temperature range.

Parameter Symbol Conditions Min. Typ. Max. Units

Notes:

1. All limits at operating temperature extremes are guaranteed by design, characterization and statistical quality control

2. Guaranteed by design – NOT tested in production.

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 7

Power Good Delay Time

Initial Start Up

VID Code Changing

VID Code Static

•

100 250

200

10 ms

µs

Crowbar Trip Point VCROWBAR Relative to Nominal DAC Output •180 300 400 mV

Crowbar Reset Point Relative to FBRTN •570 700 830 mV

Crowbar Delay Time

VID Code Changing

VID Code Static

tCROWBAR

Over voltage to PWM Going Low •100 250

400

500

600

µs

PWM Outputs

Output Voltage Low VOL(PWM) IPWM(SINK) = 400µA •160 500 mV

Output Voltage High VOH(PWM) IPWM(SOURCE) = 400µA •455.5V

Input Supply

DC Supply Current EN = Logic High •510mA

UVLO Threshold VUVLO Vcc Rising (Vcc = 12V input) •6.5 6.9 8.1 V

UVLO Hysteresis •0.7 V

UVLO Threshold Falling •5.6 7.1

Electrical Specifications (continued)

(VCC = 12V, TA = 0°C to +85°C and FBRTN=GND, using circuit in Figure 1, unless otherwise noted.)

The • denotes specifications which apply over the full operating temperature range.

Parameter Symbol Conditions Min. Typ. Max. Units

Notes:

1. All limits at operating temperature extremes are guaranteed by design, characterization and statistical quality control

2. Guaranteed by design – NOT tested in production.

FAN5018BPRODUCT SPECIFICATION

8REV. 1.0.0 Jul/15/05

Internal Block Diagram

Phase

Current

Balancing

Circuit

VID

DAC

REF

Current

Limit

Circuit

Oscillator

UVLO

Shutdown

& Bias

1314

CMP

Soft

Start

2/3/4 Phase Driver

Logic

RESET

SET

RESET

CURRENT

LIMIT

CROWBAR

CMP

CSA

Error

Amp

7 654321

DELAY

VID4 VID3 VID2 VID1 VID0 VID5

COMP

PWRGD

FBRTN

RAMPADJ

CSCOMP

CSSUM

VCC

SW2

SW3

PWM3

PWM4

PWM1

GND

ILIMIT

CSREF

SW1

PWM2

SW4

Delay

CMP

DAC

-250mV

CSREF

DAC

+150mV

3.75V

Sel VRM9 Table

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 9

Typical Characteristics

Test Circuits

TPC 1. Master Clock Frequency

TPC 2. Supply Current vs. Master Clock Frequenc

3.5

2.5

1.5

0.5

100

0 200

RESISTOR’S RT VALUE (kΩ)

MASTER CLOCK FREQUENCY (MHz)

300 400

5.2

5.3

5.1

5.0

4.9

4.8

4.7

4.6

1.0

0 2.0

MASTER CLOCK FREQUENCY (MHz)

SUPPLY CURRENT (mA)

3.0 4.00.5 1.5 2.5 3.5

TA = 25 C

4-Phase Operation

VCC

16 CSREF

CSSUM

CSCOMP

GND

CSA

100nF

39kΩ

1kΩ

+12V

CSCOMP–1V

VOS =

Test Circuit 1. Current Sense Amplifier VOS

VCC

16 CSREF

CSSUM

CSCOMP

GND

CSA

100nF200k

+12V

10k

COMP

200k

80mV

VID

FB VFB

−=

Test Circuit 2. Voltage Positioning

VID4

VID0

VID1

VID2

VID3

VID5

FAN5018B

VCC1 VID4

RAMPADJ

DELAY

PWRGD

COMP

FBRTN

VID5

VID0

VID1

VID2

VID3

ILIMIT

CSREF

CSSUM

CSCOMP

GND

SW4

SW3

SW2

SW1

PWM4

PWM3

PWM2

PWM1

250k

100nF

4.7nF 250k

20k

1μF

100nF

+12V

1kΩ

Test Circuit 3. Closed Loo

Out

ut Volta

e Accurac

FAN5018BPRODUCT SPECIFICATION

10 REV. 1.0.0 Jul/15/05

Application Circuit

Figure 1. Typical Application – 3-phase, 65A (DC), 74A (Peak) VRD/VRM10 Design

U1 FAN5009

HDRVVCC

PGND BOOT

LDRVPWM

+12V

U4 FAN5018B

VCC

1 VID4

27RAMPADJ

DELAY

PWRGD

COMP

FBFBRTN

VID5/SEL

VID0

VID1

VID2

VID3

ILIMIT

CSREF

CSSUM

CSCOMP

GND

SW4

SW3

SW2

SW1

PWM4

PWM3

PWM2

PWM1

Vcc CORE

0.8375V - 1.600V

74A DC, 93A Peak

+12V

DLY

CS1

SW1

SW3

SW2

CS2

PH1

PH2

PH3

R19

C12

HDRVVCC

PGND BOOT

LDRVPWM

+12V

R20

C13

HDRVVCC

PGND BOOT

LDRVPWM

+12V

C10

R21

C14

LIM

PWRGD

U3 FAN5009

U2 FAN5009

Note:

The design shown in this datasheet should be used as a reference only. Please contact your Fairchild sales representative for the latest information.

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 11

Bill of Materials

Table 1. FAN5018B VRM/VRD10 Application Bill of Materials for Figure 1

Note:

The design shown in this datasheet should be used as a reference only. Please contact your Fairchild sales representative

for the latest information.

Ref Qty Description Manufacturer/Number

U4 1 VR10, Multi-Phase Controller Fairchild FAN5018B

U1–3 3 Sync MOSFET Driver, 12V/12V Fairchild FAN5009

Q1–3 3 N-MOSFET, 30V, 50A, 8mΩ Fairchild FDD6696

Q4–9 6 N-MOSFET, 30V, 75A, 5mΩ Fairchild FDD6682

L1–3 3 Inductor, 650nH, 26A, 1.6mΩ Micrometals T50-8B/90, 5T, 16AWG

L4 1 Inductor, 630nH, 15A, 1.7mΩ Inter-Technical AK1418160052A-R63M

R1 1 10Ω, 5%

RR RDLY, RT3 301kΩ, 1%

R5 1 15.0kΩ, 1%

RPH1–3 3 100kΩ, 1%

RA, RCS2 2 24.9kΩ, 1%

RB1 1 10Ω, 1%

RB1 1.33kΩ, 1%

RSW1–3 3 0Ω, 5%

RCS1 1 37.4kΩ, 1%

RLIM 1 200kΩ, 1%

R19–21 3 1.5Ω, 5%

RTH 1 NTC Thermistor, 100kΩ, 5% Panasonic ERT-J1V V104J

C1–7 7 1.0µF, 25V, 10% X7R

C8–10 3 0.1µF, 50V, 10% X7R

C12–14, CCS 4 4700pF, 25V, 10% X7R

CDLY 1 0.047µF, 25V, 10% X7R

CB1 2200pF, 25V, 10% X7R

CA1 470pF, 50V, 10% X7R

CFB 1 100pF, 50V, 5% NPO

CX8 820µF, 2.5V, 20% 7mΩ, POLY Fujitsu FP-2R5RE821M

CZ22 10µF, 6.3V, 20% X5R

CIN 6 470µF, 16V, 20%, 36mΩ, Alum-Elec Rubycon 16MBZ470M

FAN5018BPRODUCT SPECIFICATION

12 REV. 1.0.0 Jul/15/05

Table 2. VRM10 VID Codes

VID4 VID3 VID2 VID1 VID0 VID5 VOUT (nominal)

11111XNo CPU

0 1 0 1 0 0 0.8375 V

0 1 0 0 1 1 0.8500 V

0 1 0 0 1 0 0.8625 V

0 1 0 0 0 1 0.8750 V

0 1 0 0 0 0 0.8875 V

0 0 1 1 1 1 0.9000 V

0 0 1 1 1 0 0.9125 V

0 0 1 1 0 1 0.9250 V

0 0 1 1 0 0 0.9375 V

0 0 1 0 1 1 0.9500 V

0 0 1 0 1 0 0.9625 V

0 0 1 0 0 1 0.9750 V

0 0 1 0 0 0 0.9875 V

0 0 0 1 1 1 1.0000 V

0 0 0 1 1 0 1.0125 V

0 0 0 1 0 1 1.0250 V

0 0 0 1 0 0 1.0375 V

0 0 0 0 1 1 1.0500 V

0 0 0 0 1 0 1.0625 V

0 0 0 0 0 1 1.0750 V

0 0 0 0 0 0 1.0875 V

1 1 1 1 0 1 1.1000 V

1 1 1 1 0 0 1.1125 V

1 1 1 0 1 1 1.1250 V

1 1 1 0 1 0 1.1375 V

1 1 1 0 0 1 1.1500 V

1 1 1 0 0 0 1.1625 V

1 1 0 1 1 1 1.1750 V

1 1 0 1 1 0 1.1875 V

1 1 0 1 0 1 1.2000 V

1 1 0 1 0 0 1.2125 V

1 1 0 0 1 1 1.2250 V

1 1 0 0 1 0 1.2375 V

1 1 0 0 0 1 1.2500 V

1 1 0 0 0 0 1.2625 V

1 0 1 1 1 1 1.2750 V

1 0 1 1 1 0 1.2875 V

1 0 1 1 0 1 1.3000 V

1 0 1 1 0 0 1.3125 V

1 0 1 0 1 1 1.3250 V

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 13

1 0 1 0 1 0 1.3375 V

1 0 1 0 0 1 1.3500 V

1 0 1 0 0 0 1.3625 V

1 0 0 1 1 1 1.3750 V

1 0 0 1 1 0 1.3875 V

1 0 0 1 0 1 1.4000 V

1 0 0 1 0 0 1.4125 V

1 0 0 0 1 1 1.4250 V

1 0 0 0 1 0 1.4375 V

1 0 0 0 0 1 1.4500 V

1 0 0 0 0 0 1.4625 V

0 1 1 1 1 1 1.4750 V

0 1 1 1 1 0 1.4875 V

0 1 1 1 0 1 1.5000 V

0 1 1 1 0 0 1.5125 V

0 1 1 0 1 1 1.5250 V

0 1 1 0 1 0 1.5375 V

0 1 1 0 0 1 1.5500 V

0 1 1 0 0 0 1.5625 V

0 1 0 1 1 1 1.5750 V

0 1 0 1 1 0 1.5875 V

0 1 0 1 0 1 1.6000 V

Table 2. VRM10 VID Codes (continued)

VID4 VID3 VID2 VID1 VID0 VID5 VOUT (nominal)

FAN5018BPRODUCT SPECIFICATION

14 REV. 1.0.0 Jul/15/05

Table 3. VRM9 VID Codes

VID4 VID3 VID2 VID1 VID0 VOUT (nominal)

11111No CPU

111101.100 V

111011.125 V

111001.150 V

110111.175 V

110101.200 V

110011.225 V

110001.250 V

101111.275 V

101101.300 V

101011.325 V

101001.350 V

100111.375 V

100101.400 V

100011.425 V

100001.450 V

011111.475 V

011101.500 V

011011.525 V

011001.550 V

010111.575 V

010101.600 V

010011.625V

010001.650V

001111.675V

001101.700V

001011.725V

001001.750V

000111.775V

000101.800V

000011.825V

000001.850V

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 15

General Description

Note: The information in this section is intended to assist

users in their design and understanding of the FAN 5018B

functionality. For clarity and ease of understanding, dev ice

parameters have been included in the text. In the event there

are discrepancies between values stated in this section and

the actual specification tables, the specification tables shall

be deemed correct.

Theory of Operation

The FAN5018B combines a multi-mode, fixed-frequency

PWM control with multi-phase logic outputs for use in 2, 3

and 4 phase synchronous buck CPU core supply power con-

verters. If VID5 is pulled up to a voltage greater than

VTBLSEL, then the DAC code corresponds to VRM9.

Multi-phase operation is important for producin g the high

currents and low voltages demanded by today’s micro pro-

cessors. Handling the high currents in a single-p hase con-

verter would place high thermal demands on the components

in the system, such as the inducto rs and MOSFETs. The

internal 6-bit VID DAC conform s to Int e l’s VRD/VRM 10

specifications.

The multi-mode control of the FAN5018B ensures a stable,

high-performance topology for:

• Balancing currents and thermals between phases

• High speed response at the lowest possib le switchi ng

frequency and output decoupling

• Minimizing thermal switching losses due to lower

frequenc y op eration

• Tight load line regulation and accuracy

• High current output from having up to 4 phase operation

• Reducing output ripple due to multi-phase cancellation

• PC board layout noise immunity

• Ease of use and design due to independent component

selection

• Flexibility in operation for tailoring desig n to low cost or

high performance

Number of Phases

The number of operational phases and their phase relation-

ship is determined by internal circuitry which monitors the

PWM outputs. Normally, the FAN5018B operates as a 4-

phase PWM controller. Grounding the PWM4 pin programs

a 3-phase operation; grounding the PWM3 and PWM4 pins

programs a 2-phase operation.

When the FAN5018B is initially enabled, the controller out-

puts a voltage on PWM3 and PWM4 of approximately

550mV. An internal comparator checks each pin’s voltage

versus a threshold of 400mV. If the pin is grounded, then it

will be below the threshold and the phase will be disabled.

The output impedance of the PWM pin is approximately

5kΩ. Any external pull-down resistance connected to the

PWM pin should not be less than 25kΩ to ensure proper

operation. The phase detection is made prior to starting nor-

mal operation. After this time, if the PWM output was not

grounded, then it will operate norm ally. If the PWM output

was grounded, then it will rem a in off.

The PWM outputs become logic-level output devices once

normal operation starts, and are intended for driving external

gate drivers. Since each phase is monitored independently,

operation approaching 100% duty cycle is possible. Also,

more than one output can be on at a time for overlappin g

phases.

Master Clock Frequency

The clock frequency of the FAN5018B is set with an exter-

nal resistor connected from the RT pin to ground. The fre-

quency follows the graph shown in TPC 1. To determine the

frequency per phase, the clock is divided by the number of

phases in use. If PWM4 is grounded, then divide the master

clock by 3 and if both PWM3 and 4 are grounded, then

divide by 2. If all phases are in use, divide by 4.

Output Voltage Differential Sensing

The FAN5018B provides a high accuracy VID DAC and

error-amplifier to maintain a ±14 mV output setpoint toler-

ance over temperature. Output voltage is differentially

sensed between the FB and FBRTN pins. FB should be con-

nected through a resistor to the regulation point, usually the

remote sense pin of the microprocessor. FBRT N should be

connected directly to the remote sense ground point. The

internal VID DAC and precision reference are referenced to

FBRTN, which has a typical curren t of 150µA, to allow

accurate remote sensing. The internal error amplifier com-

pares the output of the DAC to the FB pin to regulate the out-

put voltage.

Output Current Sensing

The FAN5018B provides a dedicated current sense amplifier

(CSA) to monitor the total ou tput current for proper voltage

positioning versus load current and for current limit detec-

tion. Sensing the load current at the output gives the total

average current being delivered to the load, which is an

inherently more accurate method than peak current detection

or sampling the current across a sense element such as the

low side MOSFET. There are several ways of configuring

this amplifier depending on the objectives of the system:

• Output inductor DCR sensing without thermistor for

lowest cost

• Output inductor DCR sensing with thermistor for

improved accuracy with tracking of inductor temperature

• Sense resistors for highest accuracy measurements

The positive input of the CSA is connected to the CSREF

pin, which is connected to the outpu t volt age. The inputs to

the amplifier are summed together through resistors from the

sensing element (such as the switch node side of the output

inductors) to the inverting input, CSSUM. The feedback

resistor between CSCOMP and CSSUM sets the gain of the

FAN5018BPRODUCT SPECIFICATION

16 REV. 1.0.0 Jul/15/05

amplifier, and a filter capacitor is placed in parallel with this

resistor. The amplifier’s gain is programmable by adjusting

the feedback resistor to set the load line required by the

microprocessor. The current information is then given as the

difference of CSREF –CSCOMP. This difference signal is

used internally to offset the VID DAC for voltage position-

ing and as a differential input for the current limit compara-

tor.

To provide the best accuracy for the current sensing, the

CSA is designed to have a low offset input voltage. Also,

external resistors determine the sensing gain so that it can be

made extremely accurate and flexible.

Active Impedance Control Mode

For controlling the output voltage droop as a fun c tion of

output current, the current sense amplifier (CSA) creates a

voltage signal proportional to the total inductor currents.

External components determine the ratio of this voltage to

the output current to allow it to be adjusted to set the

required load line. Inside the chip the CSA output voltage is

subtracted from the DAC voltage which then is used for the

reference to the error amplifier. As the output current

increases the reference to the error amp decreases causing

the output voltage to decrease accordingly.

Current Control Mode and Thermal Balance

The FAN5018B has individual inputs for each phase which

are used for monitoring the current in each phase. This infor-

mation is combined with an internal ramp to create a current

balancing feedback system that is optimized for initial cur-

rent balance accuracy and dynamic thermal balancing during

operation. This current balance information is independent

of the average output current information used for position-

ing descri bed previously.

The magnitude of the internal ram p can be set to optimize

the transient response of the system. It also monitors the sup-

ply voltage for feed-forward control for changes in the sup-

ply. A resistor connected from the power input voltage to the

RAMPADJ pin determines the slope of the internal PWM

ramp. Detailed information about programming the ramp is

given in the Application Informatio n section.

External resistors can be placed in series with individual

phases to create an intentional current imbalance if desired,

such as when one phase may have better cooling and can

support higher currents. Resistors RSW1 through RSW4

(see the typical application circuit in Figure 4) can be used

for adjusting FET thermal and current balance. Zero ohm

placeholder resistors should be provided in the initial layout

to allow the phase balance to be adjusted during design fine

tuning.

To increase the current in any given phase, make RSW for

that phase larger (make R SW = 0 for the hottest phase and do

not change during balancing). Increasing RSW to only 500Ω

will substantially increase the phase current. Increase each

RSW value by small amounts to achieve balance,

starting with the coolest phase first.

Voltage Control Mode

The voltage-mode control loop uses a high gain-bandwidth

voltage mode error amplifier. The control input voltage to

the positive input is set via the VID 6-bit logic code, accord-

ing to the voltages listed in Table 1. This voltage is also off-

set by the droop voltage for active positioning of the out put

voltage as a function of current, commonly known as active

voltage positioning. The output of the amplifier is the COMP

pin, which sets the termination voltage for the internal PWM

ramps.

The negative input (FB) is tied to the output sense locati on

with a resistor RB and is used for sensing and controlling the

output voltage at this point. A current source from the FB pin

flowing through RB is used for setting th e no-load offset

voltage from the VID voltage. The no-load voltage will be

negative with respect to the VID DAC. The main loop com-

pensation is incorporated in the feedback network between

FB and COMP.

Soft-Start

The power -on ramp up time of the output voltage is set with

a capacitor and resistor in parallel from the DELAY pin to

ground. The RC time constant also determines the current

limit latch off time as explained in the follow ing section. In

UVLO or when EN is a logic low, the DELAY pin is held at

ground. After the UVLO threshold is reached and EN is a

logic high, the DELAY cap is charged up with an internal

20µA current source. The output voltage follo ws the ram p-

ing voltage on the DELAY pin , li mit ing the inrush current.

The soft-start time depends on the value of VID DAC and

CDLY, with a secondary effect from RDLY. Refer to the Appli-

cation Information section for detailed information on set-

ting CDLY.

If EN is taken low or VCC drops below UVLO, the DELAY

cap is reset to ground to be ready for another soft start cycle.

Figure 1 shows a typical start-up sequence for the

FAN5018B.

Over Current Limit and Latch-off Protection

The FAN5018B compares a programma bl e current limit set

point to the voltage from the output of the current sense

amplifier. The level of current limit is set with the resistor

from the ILIMIT pin to ground. Duri ng normal operation,

the voltage on ILIMIT is 3V. The current through the exter-

nal resistor is internally scaled to give a current limit thresh-

old of approximately 10.4mV/µA. If the difference in

voltage between CSREF and CSCOMP rises above the cur-

rent limit threshold, the inte rnal current limit amplifier will

control the internal COMP vo ltage to maintain the average

output current at the limit.

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 17

After the limit is reached, the 3V pull-up on the DELAY pin

is disconnected, and the external delay capacitor is dis-

charged through the external resistor . A comparator monitors

the DELAY voltage and shuts of f the controller when the

voltage drops below 1.8V. Therefore, the RC time constant

discharging from 3V to 1.8V sets the current limit latch off

delay time. The Application Information section discus ses

the selection of CDLY and RDLY.

Because the controller continues to cycle the phases during

the latch-off delay time, if the short is removed before the

1.8V threshold is reached, the controller will return to nor-

mal operation. The recovery characteristic depen ds on the

state of PWRGD. If the output voltage is within the PWRGD

window, the controller resume s normal op erati on. However,

if a short circuit has caused the output voltage to drop below

the PWRGD threshold, then a soft-start cycle is initiat ed.

The latch-off function can be reset by either cycling VCC to

the FAN5018B, or by cycling the Enable pin low for a short

time. To disable the short circuit latch off function, the

external resistor to ground should be left open, and a 1MΩ

resistor should be connected from VCC to the DELAY pin .

This prevents the DELAY capacitor from discharging, so the

1.8V threshold is never reached. The resistor will have an

impact on the soft-start time because the current through it

will add to the internal 20µA current source.

During start-up when the ou tput voltage is below 200mV, a

secondary current limit is active. This is necessary because

the voltage swing of CSCOMP cann ot go below ground.

This secondary current limit controls the internal COMP

voltage to the PWM comparators to 2V. This will limit the

voltage drop across the low-side MOSFETs through the

current balance circuitry.

There is also an inherent per phase current limit that will pro-

tect individual phases if one or more phases stops function-

ing because of a faulty component. This

limit is based on the

maximum normal-mode COMP voltage.

Dynamic VID

The FAN5018B incorporat es the ability to dynami cally

change the VID input while the controller is running. This

allows the output voltage to change while the supply is run-

ning and supplying current to the load. This is commonly

referred to as VID-on-the-f ly (OTF). A VID-O TF can occur

under either light load or heavy load conditions. The proces-

sor signals the controller by changing the VID inputs in mul-

tiple steps from the start code to the finish code. This change

can be either positive or negative.

When a VID input changes state, the FAN5018B detects the

change and ignores the DAC inputs for a minimum of 400ns.

This time is to prevent a false code due to logic skew while

the six VID inputs are changing. Additionally, the first VID

change initiates the PWRGD and CROWBAR blanking

functions for a minimum of 250 µs to prevent a false

PWRGD or CROWBAR event. Each VID change will reset

the internal timer . Figure 4 shows the VID on-the-fly perfor-

mance when the output voltage is stepping up and the output

current is switching between minimum and maximum values

which is the worst-case situation.

Figure 4. VID On-the-Fly Waveforms, Circuit of Figure 5,

VID Change = 5mV, 5µs, 50 steps, IOUT Change = 5A to 65A

Figure 2. Start-Up Waveforms Figure 3. Overcurent Latch Off Waveform

Circuit of Figure 5Circuit of Figure 5

Channel 1 – Vout, Channel 2 – VccChannel 1 – Vcc, Channel 2 – Vout

Channel 3 – OD, Channel 4 – Delay pinChannel 3 – OD, Channel 4 – Delay pin

FAN5018BPRODUCT SPECIFICATION

18REV. 1.0.0 Jul/15/05

Power Good Monitoring

The Power Good comparator monitors the output voltage via

the CSREF pin. The PWRGD pin is an open drain output

whose high level (when connected to a pull-up resistor) indi-

cates that the output voltage is within the nominal limits

specified in the specifications table based on the VID voltage

setting. PWRGD will go low if the output voltage is outside

of this specified range. PWRGD is blan ked during a VID

OTF event for a period of 250µs to prevent false signals dur-

ing the time the output is changing.

Output Crowbar

As part of the protection for the load and output components

of the supply, the PWM outputs will be driven low (turning

on the low-side MOSFETs) when the output voltage exceeds

the upper Power Good threshold. This crowbar action will

stop once the output voltage has fallen below the release

threshold of approximately 550mV.

Turning on the low-side MOSFETs pulls down the output

voltage as the reverse current builds up in the inductors.

If the output overvoltage is due to a short of the high-side

MOSFET, this action will current limit the in put supply

or blow its fuse, protecting the microprocesso r from

destruction.

Output Enable and UVLO

The input supply (VCC) to the controller must be higher than

the UVLO threshold and the EN pin must be higher then its

logic threshold for the FAN5018B to be gin switching. If

UVLO is less than the threshold or the EN pin is a logic low,

the FAN5018B is disabled. This holds the PW M outputs at

ground, shorts the DELAY capacito r to ground, and holds

the ILIMIT pin at ground.

In the application circuit, the ILIMIT pin should be con-

nected to the output disable pins of the FAN5009 driv ers.

Because ILIMIT is grounded, this disables the drivers such

that both DRVH and DRVL are grounded. This feature is

important to prevent discharging of the output capacito rs

when the controller is shut of f. If the driver outputs were not

disabled, then a negativ e voltage could be generated on the

output due to the high current discharge of the output

capacitors through the inductors.

Application Information

The design parameters for a typical Intel VRD10.x-compli-

ant CPU application are as follows:

• Input voltage (VIN) = 12V

• VID setting voltage (VVID) = 1.500V

• Duty cycle (D) = 0.125

• Nominal output voltag e at no load (VONL) = 1.480V

• Nominal output voltag e at 65A load (VOFL) = 1.3955V

• Static output voltage drop based on a 1.3 mΩ load line

(RO) from no load to full load

•(V

D) = VONL – VOFL = 1.480V – 1.3955V = 84.5mV

• Maximum output current (I O) = 65A

• Maximum output current step (ΔIO) = 60A

• Number of phases (n) = 3

• Switching frequency per phase (fSW) = 228 kHz

Setting the Clock Frequency

The FAN5018B uses a fixed-frequency contro l archi tecture

with the frequency being set by an external timing resistor

(RT). The clock frequency and the number of phases deter-

mine the switching frequency per phase, which relates

directly to switching losses and the sizes of the inductors and

input and output capacitors. With n = 3 for three phases, a

clock frequency of 684kHz sets the switching frequency of

each phase, fSW, to 228kHz, which represents a practical

trade-off between the switching losses and the sizes of the

output filter components. TPC 1 shows that to achieve a

684kHz oscillator frequency, the correct value for RT is

301kΩ. Alternatively, the value for RT can be calculated

using:

where 5.0pF and 110nS are internal IC component values.

For good initial accuracy and frequency stability, it is recom-

mended to use a 1% resistor.

Soft-Start and Current Limit Latch-Off Delay

Times

Because the soft-start and current limit latch off delay

functions share the DELAY pin, these two parameters must

be considered together. The first step is to set CDLY for the

soft-start ramp. This ramp is generated with a 20µA internal

current source. The value of RDLY will have a second order

impact on the soft-start time because it sinks part of the cur-

rent source to ground. Howev er, as long as RDLY is kept

greater than 200kΩ, this effect is minor. The value for CDLY

can be approximated using:

( )

nSpFfn

T1105

−××

=(1)

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 19

where tSS is the desired soft-start time. Assuming an RDLY of

301kΩ and a desired a soft-start time of 3ms, CDLY is 35nF.

A close standard value for CDLY is 47nF. Once CDLY has

been chosen, RDLY can be calculated for the current limit

latch-off time using:

If the result for RDLY is less than 200kΩ, then a smaller soft-

start time shoul d be considered by recalculating the equation

for CDLY or a longer latch-off time should be used. RDLY

should never be less than 200kΩ. In this example, a delay

time of 8ms gives RDLY = 334kΩ. A close

standard 1% value is 301kΩ.

Inductor Selection

The choice of inductance value for the inductor det ermines

the ripple current in the in ductor. Less inductance leads to

more ripple current, which increases the output ripple volt-

age and conduction losses in the MOSFETs, but allows the

use of smaller-size inductors and, for a specified peak-to-

peak transient deviation, less total output capacitance. Con-

versely, a higher inductance means lower rip ple current and

reduced conduction losses, but requires larger inductors and

more output capacitance for the same peak-to-peak transient

deviation. In any multi-phase converter, a practical value for

the peak-to-peak inductor ripple current is less th an 50% of

the maximum DC curren t in the same inductor. Equation 4

shows the relationship between the inductance, oscillator

frequency, and peak-to-peak ripple current in the inductor.

Equation 5 can be used to determine the minimum induc-

tance based on a given output ripple voltag e:

Solving Equation 5 for a 10 mVp-p output ripple volta ge

yields:

If the ripple voltage ends up less than that designed for, the

inductor can be made smaller until the ripple value is met .

This will allow optimal transient response and minimum

output decoupling.

The smallest possible induct or shou ld be used to minimize

the number of output capacitors. Choosing a 650nH inductor

is a good choice for a starting point and gives a calculated

ripple current of 8.86A. The inductor should not saturate at

the peak current of 26.1A and shoul d be able to handle the

sum of the power dissipation caused by the average current

of 21.7A in the winding and core loss.

Another important factor in the indu ctor de si gn is the DC

Resistance (DCR), which is used for measuring the phase

currents. A large DCR will cau se excessive power losses,

while too small a value will lead to increased measurement

error. A goo d rule of thumb is to hav e the DCR be about

1 to 1 1/2 times the droop resistance (RO). For our ex ample,

we are using an inductor with a DCR of 1.6 mΩ.

Designing an Inductor

Once the inductance and Direct-Current resistance (DCR)

are known, the next step is either to design an inductor or

find a standard inductor that comes as close as possible to

meeting the overall design goals. It is also important to have

the inductance and DCR tolerance specified to keep the

accuracy of the system controlled. Using 15% for the induc-

tance and 8% for the DCR (at room temperature) are reason-

able tolerances that most manufacturers can meet.

The first decision in designing the inductor is to choose the

core material. There are several possibilities fo r providing

low core loss at high frequencies. Two examples are the

powder cores (e.g., Kool-Mm® from Magnetics, Inc. or

Micrometals) and the gapped soft ferrite cores (e.g., 3F3

or 3F4 from Philips). Low-frequency powdered iron cores

should be avoided due to their high core loss, especially

when the inductor value is relatively low and the ripple

current is high.

The best choice for a core geometry is a closed-loop type,

such as pot cores, PQ, U, and E cores, or toroids. A good

compromise between price and performance are cores with a

toroidal shape.

There are many useful references for quickly designing a

power inductor, such as:

Magnetics Design References

1. Magnetic Designer Software: Intusoft

(www.intusoft.com)

2. Designing Magn etic Components for High-Frequency

DC-DC Converters, by William T. McLyman,

Kg Magnetics, Inc. ISBN 1883107008

VID

DLY

VID

DLY V

AC ×

⎭

⎬

⎫

⎩

⎨

⎧

−= 2

20μ(2)

DLY

DELAY

DLY C

=96.1 (3)

()

−×

(4)

()()

RIPPLESW

OVID

DnRV

L×

×−××

≥1 (5)

()

mVkHz

534

10228

375.013.15.1 =

−×Ω×

≥

FAN5018BPRODUCT SPECIFICATION

20 REV. 1.0.0 Jul/15/05

Selecting a Standard Inductor

The companies listed below can provide design consultation

and deliver power inductors optimized for high power

applications upon request.

Power Inductor Manufacturers

•Coilcraft

(847)639-6400

www.coilcraft.com

• Coiltronics

(561)752-5000

www.coiltronics.com

• Sumida Electric Company

(510) 668-0660

www.sumida.com

• Vishay Intertechnology

(402) 563-6866

www.vishay.com

Output Droop Resistance

The design requires that the regulator output voltage

measured at the CPU pins drops when the output current

increases. The specified voltage drop corresponds to a DC

output resistance (RO).

The output current is measured by summing together the

voltage across each inductor and then passing the signal

through a low-pass filter. This summer-filter is the CS

amplifier configured with resistors RPH(X) (summers), and

RCS and CCS (filter). The output resistance of the regulator is

set by the following equations, where RL is the DCR of the

output inductors:

One has the flexibility of choosing either RCS or RPH(X).

It is best to select RCS equal to 100kΩ, and then solve for

RPH(X) by rearranging Equation 6.

Next, use Equation 7 to solve for CCS:

It is best to have a dual location for CCS in the layout so stan-

dard values can be used in parallel to get as close to the value

desired. For this example, choosing CCS to be 4.7nF is a

good choice. For best accuracy, CCS should be a 5% or 10%

NPO capacitor. A close standard 1% value for RPH(X) is

100kΩ.

Inductor DCR Temperature Correction

With the inductor’s DCR being used as the sense element,

and copper wire being the source of the DCR, one needs to

compensate for temperature changes of the inductor’s wind-

ing. Fortunately, copper has a well-known temperature coef-

ficient (TC) of 0.39%/°C.

If RCS is designed to have an opposite and equal percentage

change in resistance to that of the wire, it will cancel the

temperature variation of the inductor’s DCR. Due to the non-

linear nature of NTC thermistors, resist ors RCS1 and RCS2

are needed (see Figure 5) to linearize the NTC and produ ce

the desired temperature tracking.

Figure 5. Temperature Compensation Circuit

The following procedure and expressions will yield values to

use for RCS1, RCS2, and RTH (the thermistor value at 25°C)

for a given RCS value.

1. Select an NTC to be used based on type and value. Since

we do not have a value yet, start with a thermistor with a

value close to RCS. The NTC should also have an initial

tolerance of better than 5%.

2. Based on the type of NTC, find its relative resistance

value at two temperatures. The temperatures to use that

work well are 50°C and 90°C. We will call these resis-

tance values A (RTH(50°C))/RTH(25°C)) and B (RTH(90°C)/

RTH(25°C)). Note that the NTC’s relative value is always

1 at 25°C.

3. Next, find the relative value of RCS required for each of

these temperatures. This is based on the percentage

change needed, which we will initially make 0.39%/°C.

We will call these r1 and r2 where:

XPH

R×=

)(

(6)

CSL

C×

(7)

XPH R

R×=

)(

Ω=Ω×

=kk

XPH

123100

3.1

6.1

)(

CCS 06.4

1006.1

650 =

Ω×Ω

16 CSREF

CSSUM

CSCOMP

CSA

CCS

1.8nF

RCS1

RCS2

RTH

RPH1

RPH3 RPH2

Keep this path as

short as possible

and well away from

Switch Node lines

Place as close as

possible to nearest

inductor or low-side

MOSFET

To Switch Nodes

To VOUT

sense

()( )

251

1−×+

=TTC

()( )

251

2−×+

=TTC

TC = 0.0039

T1 = 50°C

T2 = 90°C

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 21

4. Compute the relative values for RCS1, RCS2, and RTH

using:

5. Calculate RTH = rTH x RCS, then select the closest value

of thermistor available. Also compute a scaling factor k

based on the ratio of the actual thermistor value used

relative to the computed one:

6. Finally, calculate values for RCS1 and RCS2 using the

following:

For this example, RCS has been chosen to be 100kΩ, so we

start with a thermistor value of 100kΩ. Looking through

available 0603 size thermistors, we find a Panasonic

ERT- J1VV104J NTC thermistor with A = 0.2954 and

B = 0.05684. From these we compute RCS1 = 0.3304,

RCS2 = 0.7426 and RTH = 1.165. Solving for RTH yields

116.5 kΩ, so we choose 100kΩ, making k = 0.8585. Finally,

we find RCS1 and RCS2 to be 28.4kΩ and 77.9kΩ. Choosing

the closest 1% resistor values yields a choice of 35.7kΩ and

73.2kΩ.

Output Offset

Intel’s specification requ ires that at no load the nominal out-

put voltage of the regulat or be offset to a lower value than

the nominal voltage corresponding to the VID code. The off-

set is set by a constant current source flowing out of the FB

pin (IFB) and flowing through RB. The value of RB can be

found using Equation 11:

The closest standard 1% resistor value is 1.33 kΩ.

COUT Selection

The required output decouplin g for the regulator is typically

recommended by Intel for various processors and platforms.

There are also some simple design guidelines to determine

what is required. These guidelines are based on having both

bulk and ceramic capacitors in the system.

The first step is to select the total amount of ceramic capaci-

tance. This is based on the number and type of capacitor to

be used. The best location for ceramics is inside the socket,

with 12 to 18 of size 1206 being the physical limit. Others

can be placed along the outer edge of the socket as well.

Combined ceramic values of 200µF–300µF are recom-

mended, usually made up of multiple 10µF or 22µF

capacitors. Select the number of ceramics and find the total

ceramic capacitance (CZ).

Next, there is an upper limit imposed on the total amount of

bulk capacitance (CX) when one considers the VID on-the-

fly voltage stepping of the output (voltage step VV in time tV

with error VERR) and a lower limit based on meeting the crit-

ical capacitance for load release for a given maximum load

step ΔIO:

To meet the conditions of these expressions and transient

response, the ESR of the bulk capacitor bank (RX) should be

less than two times the droop resistance, RO. If the CX(MIN)

is larger than CX(MAX), the system will not meet the VID

on-the-fly specification and may require the use of a smaller

inductor or more phases (and may have to increase the

switching frequency to keep the output ripple the same).

For our example, 22 10µF 1206 MLC capacitors (CZ =

220µF) were used. The VID on-the-fly step change is

250mV in 150µs with a setting error of 2.5mV. Solving for

the bulk capacitance yields:

)()1()1(

)1()1()(

1221

2BArABrBA

rABrBArrBA

CS −−×−×−×−×

×−×+×−×−××−

212

)1(

CSCS

−

= (8)

CSCS

−

)(

CALCULATEDTH

ACTUALTH

k= (9)

11 CSCSCS rkRR ××= (10)

()( )()

CSCSCS

rkkRR ×+−×=

ONLVID

R−

= (11)

Ω=

−

33.1

480.15.1

⎟

⎠

⎞

⎜

⎝

⎛−

××

Δ×

≥

VIDO

MINX

VRn

)(

(12)

VID

MAXXC

nKR

RnK

C−

⎟

⎠

⎞

⎜

⎝

⎛−

⎟

⎠

⎞

⎜

⎝

⎛×+××≤ 11

)(

(13)

where

⎟

⎠

⎞

⎜

⎝

⎛

−=

VERR

Kln

FAN5018BPRODUCT SPECIFICATION

22 REV. 1.0.0 Jul/15/05

mFF

AnH

CMINX45.6220

5.13.13

60650

)( =

⎟

⎠

⎞

⎜

⎝

⎛−

×Ω×

≥μ

( )

nHmV

mVs

mVnH

CMAXX9.232201

650250

3.16.435.1150

5.13.16.43

250650 2

)( =−

⎟

⎠

⎞

⎜

⎝

⎛−

⎟

⎠

⎞

⎜

⎝

⎛

Ω××××

+×

×Ω××

≤μ

where K=4.6

Using eight 820µF A1-Polys with a typical ESR of 8mΩ,

each yields CX = 6.56µF with an RX = 1.0mΩ. One last

check should be made to ensure that the ESL of the bulk

capacitors (LX) is low enough to limit the initial high-

frequency transient spike. This can be tested usi n g:

In this example, LX is 375pH for the eight A1-Poly capaci-

tors, which satisfies this lim itation. If the LX of the chosen

bulk capacitor bank is too large, the number of MLC capaci-

tors must be increased.

Note: For this multi-mode control technique, “all-

ceramic” designs can be used as long as the conditions of

Equations 11, 12 and 13 are satisfied.

Power MOSFETs

For this example, the N-channel power MOSFETs have been

selected for one high-side switch and two low-sid e switches

per phase. The main selection parameters for the power

MOSFETs are VGS(TH), QG, CISS, CRSS and RDS(ON).

The minimum gate drive voltage (the supply voltage to the

FAN5009) dictates whether standard threshold or logic-level

threshold MOSFETs must be used. With VGATE ~10V,

logic-level threshold MOSFETs (VGS(TH) < 2.5V) are

recommended. The maximum output current IO determines

the RDS(ON) requirement for the lo w-side (synchronous)

MOSFETs. With the FAN5018B, currents are balanced

between phases, thus the current in each low-side MOSFET

is the output current divided by the total numb er of

MOSFETs (nSF). With conduction losses being dominant,

the following expression shows the total power being

dissipated in each synchronous MOSFET in terms of the

ripple current per phase (IR) and average total output

current (IO):

Knowing the maximum output current being designed for

and the maximum allowed power dissipation, one can find

the required RDS(ON) for the MOSFET. For D-PA K

MOSFETs up to an ambient temperature of 50ºC, a safe limit

for PSF is 1W–1.5W at 125ºC junction temperature. Thus,

for our example (65A maximu m), we find RDS(SF)

(per MOSFET) < 8.7mΩ. This RDS(SF) is also at a junction

temperature of about 125ºC, so we need to make sure we

account for this when making this selection. For our exam-

ple, we selected two lower side MOSFETs at 8.6mΩ each at

room temperature, which gives 8.4mΩ at high temperature.

Another important factor for the synchro nous MOSFET is

the input capacitance and feedback capacitance. The ratio

of the feedback to input needs to be small (less than 10% is

recommended), to prevent accidental turn-on of the synchro-

nous MOSFETs when the switch node goes high.

Also, the time to switch the synch ron ous MOSFETs off

should not exceed the non-overlap dead time of the

MOSFET driver (40ns typical for the FAN5009). The

output impedance of the driver is about 2Ω and the typical

MOSFET input gate resistances are about 1Ω–2Ω, so a total

gate capacitance should be less than 6000pF. Since there are

two MOSFETs in parallel, we should limit the input capaci-

tance for each synchronous MOSFET to 3000pF.

The high-side (main) MOSFET has to be able to handle two

main power dissipation components; conduction and switch-

ing losses. The switching loss is related to the amount of

time it takes for the main MOSFET to turn on and off, and to

the current and voltage that are being switched. Basing the

switching speed on the rise and fall time of the gate driver

impedance and MOSFET input capacitance, the following

expression provides an approximate value for the switching

loss per main MOSFET, where nMF is the total number of

main MOSFETs:

Here, RG is the total gate resistance (2Ω for the FAN5009

and about 1Ω for typical high speed switching MOSFETs,

making RG = 3Ω) and CISS is the input capacitance of the

main MOSFET. Adding more main MOSFETs (nMF) does

not significantly help the switch ing loss per MOSFET since

the additional gate capacitance slows down swit ching. The

best way to reduce switching loss is to use lower gate capac-

itance devices.

The conduction loss of the main MOSFET is given by the

following, wh ere R DS(MF) is the ON-resistance of the

MOSFET:

RCL ZX ×≤ (14)

()

pHmFLX3723.1220 2=Ω×≤μ

( )

)(

SFDS

DP ×

⎥

⎦

⎤

⎢

⎣

⎡

⎟

⎠

⎞

⎜

⎝

⎛×

×+

⎟

⎠

⎞

⎜

⎝

⎛

×−=

(15)

ISS

OCC

SWMFSC

fP ×××

××= 2

)(

(16)

)(

)( 12

MFDS

MFCR

DP ×

⎥

⎦

⎤

⎢

⎣

⎡

⎟

⎠

⎞

⎜

⎝

⎛×

×+

⎟

⎠

⎞

⎜

⎝

⎛

×= (17)

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 23

Typically, for main MOSFETs, one wants the highest

speed (low CISS) device, but these usually have higher

ON-resistance. Select a device that meets the total power

dissipation (about 1.5 W for a single D-PAK) when combin-

ing the switching and conduction losses.

For our example, we have selected a Fairchild FD6696 as the

main MOSFET (three total; nMF = 3), with a Ciss = 2058 pF

(max) and RDS(MF) = 15m (max at TJ = 125ºC) and a

Fairchild FDD6682 as the synchronous MOSFET (six total;

nSF = 6), with Ciss = 2880pF (max) and RDS(SF) = 11.9mΩ

(max at TJ = 125ºC). The synchronou s MOSFET C iss is less

than 3000 pF, satisfying that requirement. Solving fo r the

power dissipation per MOSFET at IO = 65A and IR = 8.86A

yields 1.24W for each synchronous MOSFET and 1.62W for

each main MOSFET. These numbers work well considerin g

there is usually more PCB area available for each main

MOSFET versus each synchronous MOSFET.

One last item to look at is the power dissipation in the driver

for each phase. This is best described in terms of the QG for

the MOSFETs and is given by the following, where QGMF is

the total gate charge for each main MOSFET and QGSF is the

total gate charge for each synchronous MOSFET:

Also shown is the standby dissipation factor (ICC times the

VCC) for the driver . For the FAN5009, the maximum dissipa-

tion should be less than 400 mW. For our example, with

ICC = 7 mA, QGMF = 24nC (max) and QGSF = 31nC (max),

we find 202 mW in each driver, which is below the 400 mW

dissipation limit. See the FAN5009 data sheet for more

details.

Ramp Resistor Selection

The ramp resistor (RR) is used for setting the size of the

internal PWM ramp. This resistor’s value is chosen to pro-

vide the best combination of t hermal balance, stability, and

transient response. The following expression is used for

determining the optimum value:

where AR is the internal ramp amplifier gain , AD is the

current balancing amplifier gain, RDS is the total low-side

MOSFET ON-resistance, and CR is the intern al ramp

capacitor value. A close standard 1% resistor value is 301kΩ.

The internal ramp voltage magnitude can be calculated

using:

The size of the internal ramp can be made larger or smaller.

If it is made larger, stabi lit y and transient response will

improve, but thermal balance will degrade. Likewise, if the

ramp is made smaller, thermal balance wil l im prove at the

sacrifice of transient response and stability. The factor of

three in the denominator of equation 19 sets a ramp size that

gives an optimal balance for good stability, transient

response, and thermal balance.

COMP Pin Ramp

There is a ramp signal on the COMP pin due to the droop

voltage and output voltage ramps. This ramp amplitude adds

to the internal ramp to produce the following overall ramp

signal at the PWM input.

For this example, the overall ramp signal is found to be

0.974V.

Current Limit Set Point

To select the current limit set point , we need to find the

resistor value for RLIM. The current limit threshold for the

FAN5018B is set with a 3V source (VLIM) across RLIM with

a gain of 10.4mV/mA (ALIM). RLIM can be found using the

following:

For RLIM values greater than 500kΩ, the current limit may be

lower than expected, so some adjustment of RLIM may be

needed. Here, ILIM is the average current limit for the output

of the supply. For our example, choosing 120A for ILIM, we

find RLIM to be 200kΩ, for which we chose 200kΩ as the

nearest 1% value.

The per phase current limit described earlier has its limit

determined by the following:

For the FAN5018B, the maximum COMP voltage

(VCOMP(MAX)) is 3.3 V, the COMP pin bias voltage (VBIAS)

is 1.2V, and the current balancing amplifier gain (A D) is 5.

Using VR of 0.765V, and RDS(MAX) of 5.95mΩ (low-side

ON-resistance at 125°C), we find a per-phase limit of 40.44A.

()

CCCCGSFSFGMFMF

DRV VIQnQn

⎥

⎦

⎤

⎢

⎣

⎡+×+××

(18)

RDSD

RCRA

×××

(19)

Ω=

×Ω××

pFm

RR291

595.553

6502.0

()

SWRR

VIDR

RfCR

VDA

V××

×−×

=1 (20)

()

kHzpFk

VR765.0

2285301

5.1125.012.0 =

××Ω

×−×

( )

⎟

⎠

⎞

⎜

⎝

⎛

×××

×−×

−

OXSW

RCfn

V12

(21)

OLIM

LIMLIM

LIM RI

= (22)

)(

)( R

MAXDSD

BIASRMAXCOMP

PHLIM

VVV

−

−−

≅ (23)

FAN5018BPRODUCT SPECIFICATION

24 REV. 1.0.0 Jul/15/05

()

VIDOX

VID

RTL

DSDOE VRCn

VDnL

RARnR ×××

××−××

+×+×= 12 (25)

( )

OXA R

RRCT '

'−

×+−×= (26)

( )

mmmFTAμ79.4

0.1

6.03.1

3.1

375

6.03.156.6 =

Ω−Ω

+Ω−Ω×=

()

XOXB CRRRT ×−+= ' (27)

()

Ω=

×Ω××

×−××

×Ω

+Ω×+Ω×= m

VmmF

VnH

mmR

3.55

5.13.156.63

974.0375.016502

5.1

974.06.1

95.553.13

This limit can be adjusted by changing the ramp voltage VR.

Do not set the per-phase limit lower than the average per-

phase current (ILIM/n).

There is also a per phase initial duty cycle limit determ ined

by:

For this example, the maximum duty cycle is found to be

0.2696.

Feedback Loop Compensation Design

Optimized compensation of the FAN5018B allows the best

possible response of the regulator’s output to a load change.

The basis for determining the optimum compensation is to

make the regulator and output decoupling appear as an

output impedance that is entirely resistive over the widest

possible frequency range, including DC, and equal to the

droop resistance (RO). With the resistive output impedance,

the output voltage will droop in proportion with the load

current at any load current slew rate; this ensures the optimal

positioning and allows the minimization of the output

decoupling.

With the mult im ode feedback structure of the FAN5018 B,

the feedback compensation should be set to make the con-

verter’s output impedance work in conjunction with the out-

put decoupling to meet this goal. The output inductor and

decoupling capacitors (output filter) create several poles and

zeros that require compensation.

A type-III compensator on the voltage feedback is adequate

for proper compensation of the output filter . The expressions

given in Equations 25–29 are intended to yield an optimal

starting point for the design; some adjustments may be nec-

essary to account for PCB and component parasitic effects

(see the Tuning Procedure fo r the FAN5018B section).

The first step is to compute the ti me const ants for all of the

poles and zeros in the system:

where, for the FAN5018B, R' is the PCB resistance from the

bulk capacitors to the ceramics and where RDS is approxi-

mately the total low-side MOSFET ON resistance per phase

at 25ºC. For this example, AD is 5, VRT equals 0.974V, R' is

approximately 0.6mΩ (assuming a 4-layer motherboard) and

LX is 375pH for the eight Al-Poly capacitors.

The compensation values can then be solved using the fol-

lowing equations:

BIASMAXCOMP

MAX V

−

×= )( (24)

()

smFmmmTBμ97.156.63.16.00.1 =×Ω−Ω+Ω=

EVID

DSD

CRV

T×

⎟

⎠

⎞

⎜

⎝

⎛

−×

=2 (28)

kHz

nHV

TCμ86.6

3.555.1

2282

95.65

650974.0

Ω×

⎟

⎠

⎞

⎜

⎝

⎛

Ω×

−×

( )

OZOX

OZX

DRCRRC

RCC

T×+−×

××

2 (29)

()

mFmmmF

mFmF

TD500

3.12206.03.156.6

3.122056.6 2

Ω×+Ω−Ω×

Ω××

=μ

ARR

TRn

C×

××

= (30)

CA253

33.13.55

79.43.13 =

Ω×Ω

×Ω×

Ω=== k

A1.27

253

86.6

(31)

B48.1

33.1

97.1 =

== μ (32)

FB 5.18

1.27

500 =

== (33)

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 25

Choosing the closest standard va lues for these components

yields: CA = 390pF, RA = 16.9kΩ, CB = 1.5nF, and CFB =

33pF.

CIN Selection and Input Current di/dt Reduction

In continuous inductor-current mode, the source current of

the high-side MOSFET is approximately a square wave with

a duty ratio equal to n (VOUT/VIN) and an amplitude of one-

nth of the maximum output current. To prevent large voltage

transients, use a low ESR input capacitor sized for the maxi-

mum rms current. The maximum rms capacitor current is

given by:

Figure 6. Typical Transient Response for Design Example

Note that the capacitor manufacturer’s ripple current ratings

are often based on only 2000 hours of life. This makes it

advisable to furthe r derate the capacitor, or to choose a

capacitor rated at a higher temperature than required. Several

capacitors may be placed in parallel to meet size or height

requirements in the design. In this example, the input

capacitor bank is formed by three 2200µF, 16V Nichicon

capacitors with a ripple current rating of 3.5A each.

To reduce the input-current di/dt to below the recommended

maximum of 0.1A/µs, insert an additional small ind uctor (L

> 1µH @ 15A) between the converter and the supply bus.

That inductor also acts as a filter between the converter and

the primary power source.

Tuning Procedure for the FAN5018B

DC Load Line Setting

1. Build a circuit based on compensation values computed

from the design spreadsheet.

2. Hook up DC load to circuit, turn on and verify opera-

tion. Also check for jitter at no-lo a d and full-load.

3. Measure output voltage at no-load (VNL). Verify it is

within tolerance.

Figure 7. Efficiency vs. Output Current

(Circuit of Figure 5)

4. Measure output voltage at full-load cold (VFLCOLD).

Let board soak for ~10 minutes at full-load and measure

output (VFLHOT). If there is a change of more than a

couple of millivolts, adjust RCS1 and RCS2 using

Equations 35 and 37.

5. Repeat Step 4 until col d and hot voltage measurements

remain the same.

6. Measure output voltage from no-load to full-load using

5 Amp steps. Compute the loadline slope for each

change and then average to get overall loadline slope

(ROMEAS).

7. If ROMEAS is off from RO by more than 0.05 mΩ,

use the following to adjust the RPH values:

8. Repeat Steps 6 and 7 to check loadli ne and repeat

adjustments if necessary.

9. Once the DC loadline adjustment is completed, do not

change RPH, RCS1, RCS2, or RTH for the rest of proce-

dure.

1−

××= Dn

IDI OCRMS

(34)

AAICRMS 5.101

125.03

65125.0 =−

××=

()

FLHOTNL

FLCOLDNL

OLDCSNEWCS VV

−

×= )(2)(2

(35)

100

0 40

OUTPUT CURRENT (A)

EFFICIENCY (%)

6010 30 50

OMEAS

OLDPHNEWPH R

×= )()( (36)

FAN5018BPRODUCT SPECIFICATION

26 REV. 1.0.0 Jul/15/05

(37)

()

( )

)25()25(

)(1)(2)(2

)25(

)(1

)25(

)(

CTHCTH

OLDCSNEWCSOLDCS

CTH

OLDCS

CTH

OLDCS

NEWCS

ooo

RRRRRRR

−

−×−+×

10. Measure the output ripple at no-load and full-load with a

scope and make sure it is within spec.

AC Loadline Setting

11. Remove DC load from circuit and hook up dynamic

load.

12. Hook up scope to output voltage and set to DC coupling

with a time scale at 100µs/div.

13. Set dynamic load for a transient step of approximately

40A at 1kHz with 50% duty cycle.

14. Measure the output waveform (may have to use DC off-

set on scope to see waveform). Try to use a verti cal

scale of 100 mV/div or finer.

15. You will see a waveform that looks something like

Figure 8. Use the horizontal cursors to measure VACDRP

and VDCDRP as shown.

NOTE: DO NOT MEASURE THE UNDERSHOOT OR

OVERSHOOT THAT HAPPENS IMMEDIATELY AFTER

THE STEP.

Figure 8. AC Loadline Waveform

16. If the VACDRP and VDCDRP are dif ferent by more than a

couple of millivolts, use Equation 38 to adjust CCS. You

may need to parallel different values to get the right one

since there are limited standard capacitor values avail-

able (it is a good idea to have locations for two capaci-

tors in the layout for this).

17. Repeat Steps 11 to 13 and repeat adjustments if

necessary. Once complete, do not change CCS for the

rest of the procedure.

18. Set the dynamic load step to the maximum step size (do

not use a step size larger than needed) and verify that the

output waveform is square (which means VACDRP and

VDCDRP are equal). NOTE: MAKE SURE LOAD

STEP SLEW RATE AND TURN-ON ARE SET FOR A

SLEW RATE OF ~150–250A/µs (for example, a load

step of 50A should take 200ns–300ns) WITH NO

OVERSHOOT. Some dyn amic lo ads w ill have an

excessive turn-on overshoot if a minimum current is not

set properly (this is an issue if using a VTT tool).

Initial Transient Setting

19. With dynamic load still set at maximum step size,

expand the scope time scale to see 2µs/div to 5µs/div.

You will see a waveform that may have two oversho ots

and one minor undershoot (see Figure 9). Here, VDROOP

is the final desired value.

Figure 9. Transient Setting Waveform

20. If both overshoots are larger than desired, try making

the following adjustments in this order. (NOTE: If these

adjustments do not change the response, you are limited

by the output decoupling.) Check the output response

each time you make a change as well as the switching

nodes (to make sure it is still stable).

a. Make ramp resistor larger by 25% (RRAMP).

b. For VTRAN1, increase CB or increase switching fre-

quency.

c. For VTRAN2, increase RA and decrease CA by 25%.

21. For load release (see Figure 10), if VTRANREL is larger

than VTRAN1 (see Figure 9), you do not have enou gh

output capacitance. You will either need more capaci-

tance or need to make the inductor values smaller (if

you change inductors, you need to start the design over

using the spreadsheet and this tuning procedure).

100

0 40

OUTPUT CURRENT (A)

EFFICIENCY (%)

6010 30 50

DCDRP

ACDRP

OLDCSNEWCS V

CC ×= )()(

(38)

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 27

Figure 10. Transient Setting Waveform

Since the FAN5018B turns off all of the phases (switches

inductors to ground), there is no ripple voltag e present

during load release. Thus, you do not have to add headroom

for ripple, allowing your load release VTRANREL to be lar ger

than VTRAN1 by that amoun t and still be meeting spec. If

VTRAN1 and VTRANREL are less than the desired final droop,

this implies that capacitors can be removed. When removing

capacitors, check the output ripple voltage as well to ensure

it is still within spec.

Layout and Component Placement

The following guidelines are recommended for optimal

performance of a switching regu lator in a PC system.

Key layout issues are illustrated in Figure 11.

Figure 11. Layout Recommendations

General Recommendations

• For good results, at least a four-layer PCB is

recommended. This should allow the needed versatility

for control circuitry interconnections with optimal

placement, power planes for ground, input, and output

power, and wide interconnection traces in the rest of the

power delivery current paths. Keep in mind that each

square unit of 1 ounce copper trace has a resistance of

~0.53 mΩ at room temperature.

• Whenever high currents must be routed between PCB

layers, vias should be used liberally to create several

parallel current paths so that the resistance and inductance

introduced by these current paths is minimized and the via

current rating is not exceeded.

• If critical signal lines (including the output voltage sense

lines of the FAN5018B) must cross through power

circuitry, it is best if a signal ground plane can be

interposed between those signal lines and the traces of the

power circuitry to serves as a shield to minimize noise

injection into the signals..

• Use an analog ground plane both around and under the

FAN5018B for ground connections to the components

associated with the controller. Tie this plane to the nearest

output decoupling capacitor gro und and not to any other

power circuitry to prevent power currents from flowing in

it.

• Connect the components around th e FAN5018B close to

the controller with short traces. The most important traces

to keep short and away from other traces are the FB and

CSSUM pins.

• Connect the output capacitors as close as possible to the

load (or connector) that receives the power (e.g., a

microprocessor core). If the load is distributed, the

capacitors should also be distributed, and generally in

proportion to where the load tends to be m ore dynamic.

Power Circuitry

• Route the switching power path on the PCB to encompass

the shortest possible length in order to minimize radiated

switching noise energy (i.e., EMI) and conduction losses

in the board. Failure to take proper precautions often

results in EMI problems for the entire PC system as well

as noise related operational problems in the power

converter control circuitry. The switching power path is

the loop formed by the current path th rough the input

capacitors and the power MOSFETs including all

interconnecting PCB traces and planes. The use of short

and wide interconnection traces is especially critical in

this path for two reasons: it mini mizes the inductance in

the switching loop, which can cause high-energy ringing,

and it accommodates the high current demand with

minimal voltage loss. A void crossing any signal lines over

the switching power path loop, describe d below.

FAN5018BPRODUCT SPECIFICATION

28REV. 1.0.0 Jul/15/05

• Whenever a power dissipating component (e.g., a power

MOSFET) is soldered to a PCB, the liberal use of vias,

both directly on the mounting pad and imm e diat ely

surrounding it, is recommended. Two important reasons

for this are: improved current rating through the vias, and

improved thermal perform a nce from vias extended to the

opposite side of the PCB where a plane can more readily

transfer the heat to the air. To achieve the best thermal

dissipation to the air around the board, make a mirror

image of any pad being used to heatsink the MOSFETs on

the opposite side of the PCB . To further improve thermal

performance, use the largest possible pad area.

• Route the output power path to encompass a short

distance. The output power path is formed by the current

path through the inductor, the output capacitors, and the

load.

• For best EMI containment, use a solid power ground

plane as one of the inner layers extending fully under all

the power components.

Signal Circuitry

• The output voltage is sensed and regulated between th e

FB pin and the FBRTN pin (which connect s to the signal

ground at the load). To avoid differential mode noise

pickup in the sensed signal, the loop area should be small.

Thus the FB and FBRTN traces should be routed adjacent

to each other atop the power ground plane back to the

controller.

• Connect the feedback traces from the switch nodes as

close as possible to the inductor. Connect the CSREF

signal to the output voltage at the nearest inductor to the

controller.

PRODUCT SPECIFICATION FAN5018B

REV. 1.0.0 Jul/15/05 29

Mechanical Dimensions

28-Pin TSSOP

9.7 ± 0.1

– B –

0.1 C

PIN # 1 IDENT

14 ALL Lead Tips

0.2

LAND PATTERN RECOMMENDATION

0.65 0.42

– A –

4.4 ± 0.1

1.78

4.16

7.72

0.51 TYP

3.2

6.4

1.2 MAX

ALL LEAD TIPS

0.65 0.19–0.30

0.13

0.90See Detail A

0.09–0.20

0.10 ± 0.05

0°–8°

R0.31

R0.16

.025

GAGE PLANE

SEATING PLANE

DETAIL A

0.61 ± 0.1

DIMENSIONS ARE IN MILLIMETERS

NOTES:

A. Conforms to JEDEC registration MO-153, variation AB,

Ref. Note 6, dated 7/93.

B. Dimensions are in millimeters.

C. Dimensions are exclusive of burrs, mold flash, and tie bar extensions.

D Dimensions and Tolerances per ANsI Y14.5M, 1982

1.00

12.00° Top & Botom

+0.15

–0.10

BCA

– C –

FAN5018BPRODUCT SPECIFICATION

Jul/15/05 0.0m 005

Stock#DS30005019

LIFE SUPPORT POLICY

FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES

OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR

CORPORATION. As used herein:

1. Life support devices or systems are devices or systems

which, (a) are intended for surgical implant into the body,

or (b) support or sustain life, and (c) whose failure to

perform when properly used in accordance with

instructions for use provided in the labeling, can be

reasonably expected to result in a significant injury of the

user.

2. A critical component in any component of a life support

device or system whose failure to perform can be

reasonably expected to cause the failure of the life support

device or system, or to affect its safety or effectiveness.

www.fairchildsemi.com

DISCLAIMER

FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO

ANY PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION OR DESIGN. FAIRCHILD DOES NOT ASSUME

ANY LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN;

NEITHER DOES IT CONVEY ANY LICENSE UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS.

Ordering Information

Part Number Temperature Range Pb Free Package Packing

FAN5018BMTCX 0°C to +85°C Yes TSSOP-28 Tape and Reel