QT118H-IS - Quantum Research Group

APPLICATIONS -



Elevator buttons



Toys & games



Access systems



Pointing devices



Appliance control



Security systems



Light switches



Industrial panels

The QT118H charge-transfer (“QT’”) touch sensor is a self-contained digital IC capable of detecting near-proximity or touch. It will

project a sense field through almost any dielectric, like glass, plastic, stone, ceramic, and most kinds of wood. It can also turn

small metal-bearing objects into intrinsic sensors, making them respond to proximity or touch. This capability coupled with its

ability to self calibrate continuously can lead to entirely new product concepts.

It is designed specifically for human interfaces, like control panels, appliances, toys, lighting controls, or anywhere a mechanical

switch or button may be found; it may also be used for some material sensing and control applications provided that the presence

duration of objects does not exceed the recalibration timeout interval.

The IC requires only a common inexpensive capacitor in order to function. A bare piezo beeper can be connected to create a

‘tactile’ feedback clicking sound; the beeper itself then doubles as the required external capacitor, and it can also become the

sensing electrode. An LED can also be added to provide visual sensing indication. With a second inexpensive capacitor the device

can operated in 2-wire mode, where both power and signal traverse the same wire pair to a host. This mode allows the sensor to

be wired to a controller with only a twisted pair over a long distances.

Power consumption is under 20µA in most applications, allowing operation from Lithium cells for many years. In most cases the

power supply need only be minimally regulated.

The IC’s RISC core employs signal processing techniques pioneered by Quantum; these are specifically designed to make the

device survive real-world challenges, such as ‘stuck sensor’ conditions and signal drift. Even sensitivity is digitally determined and

remains constant in the face of large variations in sample capacitor Cs and electrode Cx. No external switches, opamps, or other

analog components aside from Cs are usually required.

The device includes several user-selectable built in features. One, toggle mode, permits on/off touch control, for example for light

switch replacement. Another makes the sensor output a pulse instead of a DC level, which allows the device to 'talk' over the

power rail, permitting a simple 2-wire interface. The Quantum-pioneered HeartBeat™ signal is also included, allowing a host

controller to monitor the health of the QT118H continuously if desired. By using the charge transfer principle, the IC delivers a

level of performance clearly superior to older technologies in a highly cost-effective package.

R1.03 / 0302

QProx™ QT118H

HARGE

-T

RANSFER

OUCH

ENSOR

Sns2

Vss

Sns1

GainOpt2

Opt1

Out

Vdd 1

QT118H

-QT118H-IS-40

C to +85

QT118H-DQT118H-S0

C to +70

8-PIN DIPSOICT

AVAILABLE OPTIONS



Less expensive than many mechanical switches



Projects a ‘touch button’ through any dielectric



Turns small objects into intrinsic touch sensors



100% autocal for life - no adjustments required



Only one external part required - a 1¢ capacitor



Piezo sounder direct drive for ‘tactile’ click feedback



LED drive for visual feedback



3V 20µ

µµ

µA single supply operation



Toggle mode for on/off control (strap option)



10s or 60s auto-recalibration timeout (strap option)



Pulse output mode (strap option)



Gain settings in 3 discrete levels



Simple 2-wire operation possible



HeartBeat™ health indicator on output

1 - OVERVIEW

The QT118H is a digital burst mode charge-transfer (QT)

sensor designed specifically for touch controls; it includes all

hardware and signal processing functions necessary to

provide stable sensing under a wide variety of changing

conditions. Only a single low cost, non-critical capacitor is

required for operation.

Figure 1-1 shows the basic QT118H circuit using the device,

with a conventional output drive and power supply

connections. Figure 1-2 shows a second configuration using

a common power/signal rail which can be a long twisted pair

from a controller; this configuration uses the built-in pulse

mode to transmit the output state to the host controller.

1.1 BASIC OPERATION

The QT118H employs short, ultra-low duty cycle bursts of

charge-transfer cycles to acquire its signal. Burst mode

permits power consumption in the low microamp range,

dramatically reduces RF emissions, lowers susceptibility to

EMI, and yet permits excellent response time. Internally the

signals are digitally processed to reject impulse noise, using

a 'consensus' filter which requires four consecutive

confirmations of a detection before the output is activated.

The QT switches and charge measurement hardware

functions are all internal to the QT118H (Figure 1-3). A 14-bit

single-slope switched capacitor ADC includes both the

required QT charge and transfer switches in a configuration

that provides direct ADC conversion. The ADC is designed to

dynamically optimize the QT burst length according to the

rate of charge buildup on Cs, which in turn depends on the

values of Cs, Cx, and Vdd. Vdd is used as the charge

reference voltage. Larger values of Cx cause the charge

transferred into Cs to rise more rapidly, reducing available

resolution; as a minimum resolution is required for proper

operation, this can result in dramatically reduced apparent

gain. Conversely, larger values of Cs reduce the rise of

differential voltage across it, increasing available resolution

by permitting longer QT bursts. The value of Cs can thus be

increased to allow larger values of Cx to be tolerated

(Figures 4-1, 4-2, 4-3 in Specifications, rear).

The IC is highly tolerant of changes in Cs since it computes

the threshold level ratiometrically with respect to absolute

load, and does so dynamically at all times.

Cs is thus non-critical; as it drifts with temperature, the

threshold algorithm compensates for the drift automatically.

A simple circuit variation is to replace Cs with a bare piezo

sounder (Section 2), which is merely another type of

capacitor, albeit with a large thermal drift coefficient. If C

piezo

is in the proper range, no other external component is

required. If C

piezo

is too small, it can simply be ‘topped up’

with an inexpensive ceramic capacitor connected in parallel

with it. The QT118H drives a 4kHz signal across SNS1 and

SNS2 to make the piezo (if installed) sound a short tone for

75ms immediately after detection, to act as an audible

confirmation.

Option pins allow the selection or alteration of several

special features and sensitivity.

1.2 ELECTRODE DRIVE

The internal ADC treats Cs as a floating transfer capacitor;

as a direct result, the sense electrode can be connected to

either SNS1 or SNS2 with no performance difference. In both

cases the rule Cs >> Cx must be observed for proper

operation. The polarity of the charge buildup across Cs

during a burst is the same in either case.

It is possible to connect separate Cx and Cx’ loads to SNS1

and SNS2 simultaneously, although the result is no different

than if the loads were connected together at SNS1 (or

SNS2). It is important to limit the

amount of stray capacitance on

both terminals, especially if the load

Cx is already large, for example by

minimizing trace lengths and widths

so as not to exceed the Cx load

specification and to allow for a

larger sensing electrode size if so

desired.

The PCB traces, wiring, and any

components associated with or in

contact with SNS1 and SNS2 will

become touch sensitive and should

be treated with caution to limit the

touch area to the desired location.

Multiple touch electrodes can be

used, for example to create a

control button on both sides of an

Figure 1-1 Standard mode options

SENSING

ELECTRODE

2nF - 500nF

+2.5 to 5

OUT

OPT1

OPT2

GAIN

SNS1

SNS2

Vss

Vdd

OUTPUT = DC

TIMEOUT = 10 Secs

TOGGLE = OFF

GAIN = HIGH

Figure 1-2 2-wire operation, self-powered

object, however it is impossible for the

sensor to distinguish between the two

touch areas.

1.3 ELECTRODE DESIGN

1.3.1 E

LECTRODE

EOMETRY

AND

IZE

There is no restriction on the shape of

the electrode; in most cases common

sense and a little experimentation can

result in a good electrode design. The

QT118H will operate equally well with

long, thin electrodes as with round or

square ones; even random shapes are

acceptable. The electrode can also be

a 3-dimensional surface or object.

Sensitivity is related to electrode

surface area, orientation with respect

to the object being sensed, object

composition, and the ground coupling

quality of both the sensor circuit and

the sensed object.

If a relatively large electrode surface is desired, and if tests

show that the electrode has more capacitance than the

QT118H can tolerate, the electrode can be made into a

sparse mesh (Figure 1-4) having lower Cx than a solid plane.

Sensitivity may even remain the same, as the sensor will be

operating in a lower region of the gain curves.

1.3.2 K

IRCHOFF

’

URRENT

Like all capacitance sensors, the QT118H relies on Kirchoff’s

Current Law (Figure 1-5) to detect the change in capacitance

of the electrode. This law as applied to capacitive sensing

requires that the sensor’s field current must complete a loop,

returning back to its source in order for capacitance to be

sensed. Although most designers relate to Kirchoff’s law with

regard to hardwired circuits, it applies equally to capacitive

field flows. By implication it requires that the signal ground

and the target object must both be coupled together in some

manner for a capacitive sensor to operate properly. Note that

there is no need to provide actual hardwired ground

connections; capacitive coupling to ground (Cx1) is always

sufficient, even if the coupling might seem very tenuous. For

example, powering the sensor via an isolated transformer

will provide ample ground coupling, since there is

capacitance between the windings and/or the transformer

core, and from the power wiring itself directly to 'local earth'.

Even when battery powered, just the physical size of the

PCB and the object into which the electronics is embedded

will generally be enough to couple a few picofarads back to

local earth.

1.3.3 V

IRTUAL

APACITIVE

ROUNDS

When detecting human contact (e.g. a fingertip), grounding

of the person is never required. The human body naturally

has several hundred picofarads of ‘free space’ capacitance

to the local environment (Cx3 in Figure 1-5), which is more

than two orders of magnitude greater than that required to

create a return path to the QT118H via earth. The QT118H's

PCB however can be physically quite small, so there may be

little ‘free space’ coupling (Cx1 in Figure 1-5) between it and

the environment to complete the return path. If the QT118H

circuit ground cannot be earth grounded by wire, for example

via the supply connections, then a ‘virtual capacitive ground’

may be required to increase return coupling.

A ‘virtual capacitive ground’ can be created by connecting

the QT118H’s own circuit ground to:

(1) A nearby piece of metal or metallized housing;

Figure 1-3 Internal Switching & Timing

SNS2

SNS1

ELECTRODE

Single-Slope 14-bit

Switched Capacitor ADC

C harge

Amp

Burst Controller

Result

Do ne

Start

Figure 1-5 Kirchoff's Current Law

Sense Electrode

Surrounding environment

SENSOR

(2) A floating conductive ground plane;

(3) A nail driven into a wall when used with small

electrodes;

(4) A larger electronic device (to which its output might be

connected anyway).

Free-floating ground planes such as metal foils should

maximize exposed surface area in a flat plane if possible. A

square of metal foil will have little effect if it is rolled up or

crumpled into a ball. Virtual ground planes are more

effective and can be made smaller if they are physically

bonded to other surfaces, for example a wall or floor.

1.3.4 F

IELD

HAPING

The electrode can be prevented from sensing in undesired

directions with the assistance of metal shielding connected

to circuit ground (Figure 1-6). For example, on flat surfaces,

the field can spread laterally and create a larger touch area

than desired. To stop field spreading, it is only necessary to

surround the touch electrode on all sides with a ring of metal

connected to circuit ground; the ring can be on the same or

opposite side from the electrode. The ring will kill field

spreading from that point outwards.

If one side of the panel to which the electrode is fixed has

moving traffic near it, these objects can cause

inadvertent detections. This is called ‘walk-by’

and is caused by the fact that the fields radiate

from either surface of the electrode equally well.

Again, shielding in the form of a metal sheet or

foil connected to circuit ground will prevent

walk-by; putting a small air gap between the

grounded shield and the electrode will keep the

value of Cx lower and is encouraged. In the case

of the QT118H, the sensitivity is low enough that

'walk-by' should not be a concern if the product

has more than a few millimeters of internal air

gap; if the product is very thin and contact with

the product's back is a concern, then some form

of rear shielding may be required.

1.3.5 S

ENSITIVITY

The QT118H can be set for one of 3 gain levels using option

pin 5 (Table 1-1). This sensitivity change is made by altering

the internal numerical threshold level required for a

detection. Note that sensitivity is also a function of other

things: like the value of Cs, electrode size, shape, and

orientation, the composition and aspect of the object to be

sensed, the thickness and composition of any overlaying

panel material, and the degree of ground coupling of both

sensor and object.

1.3.5.1 Increasing Sensitivity

In some cases it may be desirable to increase sensitivity

further, for example when using the sensor with very thick

panels having a low dielectric constant.

Sensitivity can often be increased by using a bigger

electrode, reducing panel thickness, or altering panel

composition. Increasing electrode size can have diminishing

returns, as high values of Cx will reduce sensor gain

(Figures 4-1, 4-2). The value of Cs also has a dramatic

effect on sensitivity, and this can be increased in value (up to

a limit). Also, increasing the electrode's surface area will not

substantially increase touch sensitivity if its diameter is

already much larger in surface area than the object being

detected. The panel or other intervening material can be

made thinner, but again there are diminishing rewards for

doing so. Panel material can also be changed to one having

a higher dielectric constant, which will help propagate the

field through to the front. Locally adding some conductive

material to the panel (conductive materials essentially have

an infinite dielectric constant) will also help; for example,

adding carbon or metal fibers to a plastic panel will greatly

increase frontal field strength, even if the fiber density is too

low to make the plastic bulk-conductive.

Pin 7Low

Pin 6Medium

FloatingHigh

Tie Pin 5 to:Gain

Table 1-1 Gain Setting Strap Options

1.3.5.2 Decreasing Sensitivity

In some cases the QT118H may be too sensitive, even on

low gain. In this case gain can be lowered further by a

number of strategies: making the electrode smaller, making

the electrode into a sparse mesh using a high

space-to-conductor ratio (Figure 1-4), or by decreasing Cs.

Figure 1-6

Shielding Against Fringe Fields

Sense

wire

Sense

wire

Unshielded

Electrode

Shielded

Electrode

Figure 2-1 Drift Compensation

Threshold

Signal Hysteresis

Reference

Output

2 - QT118H SPECIFICS

2.1 SIGNAL PROCESSING

The QT118H processes all signals using 16 bit precision,

using a number of algorithms pioneered by Quantum. The

algorithms are specifically designed to provide for high

survivability in the face of all kinds of adverse environmental

changes.

2.1.1 D

RIFT

OMPENSATION

LGORITHM

Signal drift can occur because of changes in Cx and Cs over

time. It is crucial that drift be compensated for, otherwise

false detections, non-detections, and sensitivity shifts will

follow.

Drift compensation (Figure 2-1) is performed by making the

reference level track the raw signal at a slow rate, but only

while there is no detection in effect. The rate of adjustment

must be performed slowly, otherwise legitimate detections

could be ignored. The QT118H drift compensates using a

slew-rate limited change to the reference level; the threshold

and hysteresis values are slaved to this reference.

Once an object is sensed, the drift compensation

mechanism ceases since the signal is legitimately high, and

therefore should not cause the reference level to change.

The QT118H's drift compensation is 'asymmetric': the

reference level drift-compensates in one direction faster than

it does in the other. Specifically, it compensates faster for

decreasing signals than for increasing signals. Increasing

signals should not be compensated for quickly, since an

approaching finger could be compensated for partially or

entirely before even touching the sense pad. However, an

obstruction over the sense pad, for which the sensor has

already made full allowance for, could suddenly be removed

leaving the sensor with an artificially elevated reference level

and thus become insensitive to touch. In this latter case, the

sensor will compensate for the object's removal very quickly,

usually in only a few seconds.

2.1.2 T

HRESHOLD

ALCULATION

Unlike the QT110 device, the internal threshold level is fixed

at one of two setting as determined by Table 1-1. These

setting are fixed with respect to the internal reference level,

which in turn can move in accordance with the drift

compensation mechanism..

The QT118H employs a hysteresis dropout below the

threshold level of 17% of the delta between the reference

and threshold levels.

2.1.3 M

-D

URATION

If an object or material obstructs the sense pad the signal

may rise enough to create a detection, preventing further

operation. To prevent this, the sensor includes a timer which

monitors detections. If a detection exceeds the timer setting,

the timer causes the sensor to perform a full recalibration.

This is known as the Max On-Duration feature.

After the Max On-Duration interval, the sensor will once

again function normally, even if partially or fully obstructed,

to the best of its ability given electrode conditions. There are

two timeout durations available via strap option: 10 and 60

seconds.

10sVddGndPulse

10sGndGndToggle

60sGndVddDC Out

10sVddVddDC Out

Max On-

Duration

Tie

Pin 4 to:

Tie

Pin 3 to:

Table 2-1 Output Mode Strap Options

2.1.4 D

ETECTION

NTEGRATOR

It is desirable to suppress detections generated by electrical

noise or from quick brushes with an object. To accomplish

this, the QT118H incorporates a detect integration counter

that increments with each detection until a limit is reached,

after which the output is activated. If no detection is sensed

prior to the final count, the counter is reset immediately to

zero. The required count is 4.

The Detection Integrator can also be viewed as a 'consensus'

filter, that requires four detections in four successive bursts

to create an output. As the basic burst spacing is 75ms, if

this spacing was maintained throughout all 4 counts the

sensor would react very slowly. In the QT118H, after an

initial detection is sensed, the remaining three bursts are

spaced about 18ms apart, so that the slowest reaction time

possible is 75+18+18+18 or 129ms and the fastest possible

is 54ms, depending on where in the initial burst interval the

contact first occurred. The response time will thus average

92ms.

2.1.5 F

ORCED

ENSOR

ECALIBRATION

The QT118H has no recalibration pin; a forced recalibration

is accomplished only when the device is powered up.

However, the supply drain is so low it is a simple matter to

treat the entire IC as a controllable load; simply driving the

QT118H's Vdd pin directly from another logic gate or a

microprocessor port (Figure 2-2) will serve as both power

and 'forced recal'. The source resistance of most CMOS

gates and microprocessors is low enough to provide direct

power without any problems. Note that most 8051-based

microcontrollers have only a weak pullup drive capability

and will require true CMOS buffering. Any 74HC or 74AC

series gate can directly power the QT118H, as can most

other microcontrollers. A 0.01uF minimum bypass capacitor

close to the device is essential; without it the device can

Figure 2-2 Powering From a CMOS Port Pin

0. 01 µ F

CMOS

m ic rocontroller

OUT

PORT X.m

PORT X.n

Vdd

Vss

QT118

break into high frequency oscillation, get hot, and stop

working.

Option strap configurations are read by the QT118H only on

powerup. Configurations can only be changed by powering

the QT118H down and back up again; a microcontroller can

directly alter most of the configurations and cycle power to

put them in effect.

2.2 OUTPUT FEATURES

The QT118H is designed for maximum flexibility and can

accommodate most popular sensing requirements. These

are selectable using strap options on pins OPT1 and OPT2.

All options are shown in Table 2-1.

2.2.1 DC M

ODE

UTPUT

The output of the device can respond in a DC mode, where

the output is active-high upon detection. The output will

remain active for the duration of the detection, or until the

Max On-Duration expires, whichever occurs first. If the latter

occurs first, the sensor performs a full recalibration and the

output becomes inactive until the next detection.

In this mode, two Max On-Duration timeouts are available:

10 and 60 seconds.

2.2.2 T

OGGLE

ODE

UTPUT

This makes the sensor respond in an on/off mode like a flip

flop. It is most useful for controlling power loads, for

example in kitchen appliances, power tools, light switches,

etc.

Max On-Duration in Toggle mode is fixed at 10 seconds.

When a timeout occurs, the sensor recalibrates but leaves

the output state unchanged.

2.2.3 P

ULSE

ODE

UTPUT

This generates a positive pulse of 75ms duration with every

new detection. It is most useful for 2-wire operation (see

Figure 1-2), but can also be used when bussing together

several devices onto a common output line with the help of

steering diodes or logic gates, in order to control a common

load from several places.

Max On-Duration is fixed at 10 seconds if in Pulse output

mode.

2.2.4 H

EART

EAT

™ O

UTPUT

The output has a full-time HeartBeat™ ‘health’ indicator

superimposed on it. This operates by taking 'Out' into a

3-state mode for 350µs once before every QT burst. This

output state can be used to determine that the sensor is

operating properly, or, it can be ignored using one of several

simple methods.

Since Out is normally low, a pullup resistor will create

positive HeartBeat pulses (Figure 2-3) when the sensor is

not detecting an object; when detecting an object, the output

will remain active for the duration of the detection, and no

HeartBeat pulse will be evident.

If the sensor is wired to a microcontroller as shown in Figure

2-4, the controller can reconfigure the load resistor to either

ground or Vcc depending on the output state of the device,

so that the pulses are evident in either state.

Electromechanical devices will usually ignore this short

pulse. The pulse also has too low a duty cycle to visibly

activate LED’s. It can be filtered completely if desired, by

adding an RC timeconstant to filter the output, or if

interfacing directly and only to a high-impedance CMOS

input, by doing nothing or at most adding a small non-critical

capacitor from Out to ground (Figure 2-5).

2.2.5 P

IEZO

COUSTIC

RIVE

A piezo drive signal is generated for use with a bare piezo

sounder immediately after a detection is made; the tone lasts

for a nominal 75ms to create a ‘tactile feedback’ sound.

The sensor will drive most common bare piezo ‘beepers’

directly using an H-bridge drive configuration for the highest

possible sound level at all supply voltages; H-bridge drive

effectively doubles the supply voltage across the piezo. The

piezo is connected across pins SNS1 and SNS2. This drive

operates at a nominal 4kHz frequency, a common resonance

point for enclosed piezo sounders. Other frequencies can be

obtained upon special request.

If desired a bare piezo sounder can be directly adhered to

the rear of a control panel, provided that an acoustically

resonant cavity is also incorporated to give the desired

sound level.

Since piezo sounders are merely high-K ceramic capacitors,

the sounder will double as the Cs capacitor, and the piezo's

Figure 2-4

Using a micro to obtain HB pulses in either output state

Figure 2-3

Getting HB pulses with a pulup resistor when not active

metal disc will act as the sensing electrode. Piezo transducer

capacitances typically range from 6nF to 30nF (0.006µF to

0.03µF) in value; at the lower end of this range an additional

capacitor should be added to bring the total Cs across SNS1

and SNS2 to at least 10nF, or more if Cx is large.

The burst acquisition process induces a small but audible

voltage step across the piezo resonator, which occurs when

SNS1 and SNS2 rapidly discharge residual voltage stored on

the resonator. The resulting slight clicking sound can be

used to provide an audible confirmation of functionality if

desired, or, it can be suppressed by placing a non-critical 1M

to 2M ohm bleed resistor in parallel with the resonator. The

resistor acts to slowly discharge the resonator, preempting

the occurrence of the harmonic-rich step (Figure 2-6).

With the resistor in place, an almost inaudible clicking sound

may still be heard, which is caused by the small charge

buildup across the piezo device during each burst.

2.2.6 O

UTPUT

RIVE

The QT118H’s `output is active high and it can source 1mA

or sink 5mA of non-inductive current.

Care should be taken when the IC and the load are both

powered from the same supply, and the supply is minimally

regulated. The device derives its internal references from the

power supply, and sensitivity shifts can occur with changes

in Vdd, as happens when loads are switched on. This can

induce detection ‘cycling’, whereby an object is detected, the

load is turned on, the supply sags, the detection is no longer

sensed, the load is turned off, the supply rises and the object

is reacquired, ad infinitum. To prevent this occurrence, the

output should only be lightly loaded if the device is operated

from an unregulated supply, e.g. batteries. Detection

‘stiction’, the opposite effect, can occur if a load is shed

when Out is active.

3 - CIRCUIT GUIDELINES

3.1 SAMPLE CAPACITOR

When used for most applications, the charge sampler Cs

can be virtually any plastic film or ceramic capacitor. The

type should be relatively stable in the anticipated

temperature range. If fast temperature swings are expected,

especially with higher sensitivities, more stable capacitors be

required, for example PPS film, X7R, or NPO/C0G ceramic.

3.2 PIEZO SOUNDER

The use of a piezo sounder in place of Cs is described in the

previous section. Piezo sounders have very high,

uncharacterized thermal coefficients and should not be used

if fast temperature swings are anticipated, especially at high

gains.

3.3 OPTION STRAPPING

The option pins Opt1 and Opt2 should never be left floating.

If they are floated, the device will draw excess power and the

options will not be properly read on powerup. Intentionally,

there are no pullup resistors on these lines, since pullup

resistors add to power drain if tied low.

The Gain input is designed to be floated for sensing one of

the three gain settings. It should never be connected to a

pullup resistor or tied to anything other than Sns1 or Sns2.

Table 2-1 shows the option strap configurations available.

3.4 POWER SUPPLY, PCB LAYOUT

The power supply can range from 2.5 to 5.0 volts. At 3 volts

current drain averages less than 20µA in most cases, but

can be higher if Cs is large. Large Cx values will actually

decrease power drain. Operation can be from batteries, but

be cautious about loads causing supply droop (see Output

Drive, previous section).

As battery voltage sags with use or fluctuates slowly with

temperature, the IC will track and compensate for these

changes automatically with only minor changes in sensitivity.

If the power supply is shared with another electronic system,

care should be taken to assure that the supply is free of

digital spikes, sags, and surges which can adversely affect

the device. The IC will track slow changes in Vdd, but it can

be affected by rapid voltage steps.

if desired, the supply can be regulated using a conventional

low current regulator, for example CMOS regulators that

have nanoamp quiescent currents. Care should be taken that

the regulator does not have a minimum load specification,

which almost certainly will be violated by the QT118H's low

current requirement.

Figure 2-6 Damping Piezo Clicks with R

SENS ING

ELECTRODE

Piezo Sounder

10-30nF

+2.5 to 5

OUT

OPT2

GA IN

SNS2

SNS1

Vss

Vdd

OPT1

Figure 2-5 Eliminating HB Pulses

OUT

OPT1

OPT2

GAIN

SNS1

SNS2

CMOS

100pF

GATE O R

MICRO INPUT

Since the IC operates in a burst mode, almost all

the power is consumed during the course of each

burst. During the time between bursts the sensor

is quiescent.

For proper operation a 100nF (0.1uF) ceramic

bypass capacitor should be used between Vdd and

Vss; the bypass cap should be placed very close

to the device’s power pins. Without this capacitor

the part can break into high frequency oscillation,

get physically hot, and stop working.

3.4.1 M

EASURING

UPPLY

URRENT

Measuring average power consumption is a

challenging task due to the burst nature of the

device’s operation. Even a good quality RMS DMM

will have difficulty tracking the relatively slow burst

rate.

The simplest method for measuring average current is to

replace the power supply with a large value low-leakage

electrolytic capacitor, for example 2,700µF. 'Soak' the

capacitor by connecting it to a bench supply at the desired

operating voltage for 24 hours to form the electrolyte and

reduce leakage to a minimum. Connect the capacitor to the

circuit at T=0, making sure there will be no detections during

the measurement interval; at T=30 seconds measure the

capacitor's voltage with a DMM. Repeat the test without a

load to measure the capacitor's internal leakage, and

subtract the internal leakage result from the voltage droop

measured during the QT118H load test. Be sure the DMM is

connected only at the end of each test, to prevent the DMM's

impedance from contributing to the capacitor's discharge.

Supply drain can be calculated from the adjusted voltage

droop using the basic charge equation:

✁VC

where C is the large supply cap value, t is the elapsed

measurement time in seconds, and ∆V is the adjusted

voltage droop on C.

A good approximation can be made to this method by using

a 2,700µF cap across the circuit, and inserting a 220 ohm

resistor in series with a current meter in the power wire.

3.4.2 ESD, RFI

ROTECTION

ESD protection. In cases where the electrode is placed

behind a dielectric panel, the IC will be protected from direct

static discharge. However even with a panel transients can

still flow into the electrode via induction, or in extreme cases

via dielectric breakdown. Porous materials may allow a

spark to tunnel right through the material. Testing is required

to reveal any problems. The device does have diode

protection on its terminals which can absorb and protect the

device from most induced discharges, up to 20mA; the

usefulness of the internal clamping will depending on the

dielectric properties, panel thickness, and rise time of the

ESD transients.

ESD dissipation can be aided further with an added diode

protection network as shown in Figure 2-7, in extreme cases.

Because the charge and transfer times of the QT118H are

relatively long, the circuit can tolerate very large values of

Re1 and Re2, more than 100k ohms combined in most

cases where the electrode load is small. The added diodes

shown (1N4150, BAV99 or equivalent low-C

high-conductance diodes) will shunt the ESD transients

away from the part, and Re1 will current limit the rest into

the QT118H's own internal clamp diodes. C1 should be

around 10µF if it is to absorb positive transients from a

human body model standpoint without rising in value by

more than 1 volt. If desired C1 can be replaced with an

appropriate zener diode. Directly placing semiconductor

transient protection devices or MOV's on the sense lead is

not advised; these devices have extremely large amounts of

nonlinear parasitic C which will swamp the capacitance of

the electrode.

Re1 and Re2 should be as large as possible given the load

value of Cx, Cf, and the diode capacitances of D1 and D2.

Re1 and Re2 should be low enough to permit at least 6 RC

time-constants to occur during the charge and transfer

phases. Re1 should be about 20% of Re2. Cf is used for RFI

suppression; see below.

Re3 functions to isolate the transient from the Vdd pin;

values of around 1K ohms are reasonable.

As with all protection networks, it is crucial that transients be

led away from the circuit. PCB ground layout is crucial; the

ground connections to D1, D2, and C1 should all go back to

the power supply ground or preferably, if available, a chassis

ground connected to earth. The currents should not be

allowed to traverse the area directly under the IC.

If the electrode operates behind glass or insulating plastics

thicker than 2mm, D1 and D2 can be safely deleted.

However it is still wise to use Re1, of a value as large as can

be tolerated. Values up to 100K and sometimes well beyond

can usually be tolerated quite well.

If the device is connected to an external circuit via a cable or

long twisted pair, it is possible for ground-bounce to cause

damage to the Out pin; even though the transients are led

away from the IC itself, the connected signal or power

ground line will act as an inductor causing a high differential

voltage to build up on the Out wire with respect to ground. If

this is a possibility the Out pin should have a resistance Re4

in series with it to limit current; this resistor should be as

large as can be tolerated by the load.

RFI Suppression. PCB layout, grounding, and the structure

of the input circuitry have a great bearing on the success of

a design to withstand RF fields.

Figure 2-7 ESD Protection

The circuit is remarkably immune to HF RFI provided that

certain design rules be adhered to:

1. Use SMT components.

2. Always use a ground plane under the circuit.

3. Use a 0.1uF bypass cap very close to the supply pins.

4. If ESD diodes are used, always use Re1, Re2, and Cf.

Make Re1 / Re2 as large as possible without

compromising gain (depends on Cf and Cx).

5. Keep all ESD components close to the IC.

6. Do not route the sense wire near other traces or wires

7. The sense electrode should be kept away from other

conductors, even ground, which can re-radiate in RF

currents.

8. If the ESD diodes are not used, use Re1 in the electrode

trace anyway, with a value as large as possible without

compromising gain.

Cf acts to shunt aside RF from entering the two diodes, thus

preventing their conduction due to RF currents. This form of

conduction will lead to false or erratic operation. Cf also acts

to lower sensitivity, and in many cases Cs will need to be

increased to compensate for this loss.

4.1 ABSOLUTE MAXIMUM SPECIFICATIONS

Operating temp ............................................................ as designated by suffix

Storage temp ................................................................... -55

C to +125

...............................................................................-0.5 to +6.5V

Max continuous pin current, any control or drive pin ........................................... ±20mA

Short circuit duration to ground, any pin ...................................................... infinite

Short circuit duration to V

, any pin ......................................................... infinite

Voltage forced onto any pin ................................................ -0.6V to (Vdd + 0.6) Volts

4.2 RECOMMENDED OPERATING CONDITIONS

............................................................................... +2.5 to 5.5V

Short-term supply ripple+noise ...............................................................±5mV

Long-term supply stability ................................................................ ±100mV

Cs value ......................................................................... 10nF to 500nF

Cx value ............................................................................ 0 to 100pF

4.3 AC SPECIFICATIONS

Vdd = 3.0, Ta = recommended operating range

µs300Heartbeat pulse widthT

ms75Pulse output width on OutT

ms75Piezo drive durationT

kHz4Piezo drive frequencyF

with minimal Csms129Response timeT

depends heavily on Cs, Cxms500.5Burst lengthT

ms75Burst spacing intervalT

µs2Transfer durationT

µs2Charge durationT

ms550Recalibration timeT

NotesUnitsMaxTypMinDescriptionParameter

4.4 SIGNAL PROCESSING

3, 4secs6010Post-detection recalibration timer duration (typical)

4ms/level75Negative drift compensation rate

4ms/level750Positive drift compensation rate

samples4Consensus filter length

2%17Hysteresis

1counts6, 12, or 24Threshold differential

NotesUnitsMaxTypMinDescription

Note 1: Pin options

Note 2: Of signal threshold

Note 3: Pin option

Note 4: Cs, Cx dependent

4.5 DC SPECIFICATIONS

Vdd = 3.0V, Cs = 10nF, Cx = 5pF, T

= recommended range, unless otherwise noted

Note 2fF281,000Sensitivity rangeS

bits14Acquisition resolutionA

Resistance from SNS1 to SNS2

✡

500KMin shunt resistanceI

pF1000Load capacitance rangeC

OPT1, OPT2µA±1Input leakage currentI

OUT, 1mA sourceVVdd-0.7High output voltageV

OUT, 4mA sinkV0.6Low output voltageV

OPT1, OPT2V2.2High input logic levelV

OPT1, OPT2V0.8Low input logic levelV

Required for proper startupV/s100Supply turn-on slopeV

DDS

µA20Supply currentI

V5.252.45Supply voltageV

NotesUnitsMaxTypMinDescriptionParameter

Preliminary Data: All specifications subject to change.

Figure 4-2 - Typical Threshold Sensitivity vs. Cx,

Medium Gain, Selected Values of Cs; Vdd = 3.0

0.01

0.10

1.00

10.00

0 10203040

Cx Load, pF

Detection Threshold, pF

10nF

20nF

50nF

100nF

200nF

500nF

Figure 4-1 - Typical Threshold Sensitivity vs. Cx,

High Gain, at Selected Values of Cs; Vdd = 3.0

0.01

0.10

1.00

10.00

0 10203040

Cx Load, pF

Detection Threshold, pF

10nF

20nF

50nF

100nF

200nF

500nF

0.0150.0080.3810.203Y

0.390.329.9068.128x

BSC0.30.3BSC7.0627.62Aa

0.16-4.064-S1

0.140.123.5563.048S

-0.015-0.381r

0.150.123.813.048R

Typical0.1020.098Typical2.5912.489F

0.0650.0551.6511.397L1

0.0220.0140.5590.355L

-0.01-0.254P

-0.035-0.889Q

BSC0.30.3BSC7.627.62m

Typical0.430.355Typical10.9229.017M

0.3250.38.2557.62A

0.280.247.112 6.096a

NotesMaxMinNotesMaxMin

InchesMillimeters

SYMBOL

Package type: 8pin Dual-In-Line

8º0º8º0ºØ

0.030.2290.7620.381ß

0.010.0070.2490.19e

0.040.021.0160.508E

0.0190.0140.4830.355L

BSC0.050.050BSC1.271.27D

0.010.0040.7620.101h

0.0680.0541.7281.371H

0.1570.153.9883.81Aa

0.2440.2296.1985.816W

0.1960.1894.9794.800M

NotesMaxMinNotesMaxMin

InchesMillimeters

SYMBOL

Package type: 8pin SOIC

5 - ORDERING INFORMATION

QT1 + TSOIC-8-40 - 85CQT118H-IS

QT1 + 8SOIC-80 - 70CQT118H-S

QT1 + 118PDIP0 - 70CQT118H-D

MARKINGPACKAGETEMP RANGEPART

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