LT3798
1
3798fa
n Isolated PFC Flyback with Minimum Number of
External Components
n VIN and VOUT Limited Only by External Components
n Active Power Factor Correction
n Low Harmonic Distortion
n No Opto-Coupler Required
n Constant-Current and Constant-Voltage Regulation
n Accurate Regulated Voltage and Current (±5% Typical)
n Energy Star Compliant (<0.5W No Load Operation)
n Thermally Enhanced 16-lead MSOP Package
TYPICAL APPLICATION
FEATURES DESCRIPTION
Isolated No Opto-Coupler
Flyback Controller
with Active PFC
The LT
®
3798 is a constant-voltage/constant-current iso-
lated flyback controller that combines active power factor
correction (PFC) with no opto-coupler required for output
voltage feedback into a single-stage converter. A LT3798
based design can achieve a power factor of greater than 0.97
by actively modulating the input current, allowing compli-
ance with most Harmonic Current Emission requirements.
The LT3798 is well suited for a wide variety of off-line
applications. The input range can be scaled up or down,
depending mainly on the choice of external components.
Efficiencies higher than 86% can be achieved with output
power levels up to 100W. In addition, the LT3798 can easily
be designed into high DC input applications.
Universal Input 24W PFC Bus Converter VOUT vs IOUT
APPLICATIONS
n Offline 5W to 100W+ Applications
n High DC VIN Isolated Applications
n Offline Bus Converter (12V, 24V or 48V Outputs) L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. All other trademarks are the property of their respective owners.
Protected by U.S. Patents, including 5438499 and 7471522.
3798 TA01a
VIN_SENSE
EN/UVLO
VIN DCM
FB
VREF
CTRL2
CTRL1
OVP
GATE
SENSE
INTVCC
GND
LT3798
COMP+
VCCOMP–
560µF
× 2
4.7pF
10µF
2.2nF
4:1:1
2.2µF 0.1µF
20Ω
20Ω
0.05Ω
499k
499k
95.3k
1M
D2
Z1
Z2
D1
D4
22pF
D3
100k
100k
4.99k
90.9k
CTRL3
90V
TO 265V
AC
0.1µF
40.2k
100k
16.5k
221k
4.7µF
2k
24V
1A
IOUT (A)
0
23.50
VOUT (V)
24.25
24.00
23.75
24.50
0.6 0.8 10.2 0.4
3798 TA01b
VAC = 265V
VAC = 220V
VAC = 120V
VAC = 90V
LT3798
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3798fa
PIN CONFIGURATIONABSOLUTE MAXIMUM RATINGS
EN/UVLO ...................................................................30V
VIN ............................................................................42V
INTVCC ...................................................................... 12V
CTRL1, CTRL2, CTRL3 ................................................ 4V
FB, VREF, COMP+ ........................................................3V
VC, OVP, COMP– .........................................................4V
SENSE ......................................................................0.4V
VIN_SENSE .................................................................1mA
DCM .......................................................................±3mA
Operating Temperature Range (Note 2)
LT3798E/LT3798I................................... –40°C to 125°C
Storage Temperature Range .................. –65°C to 150°C
(Note 1)
1
2
3
4
5
6
7
8
CTRL1
CTRL2
CTRL3
VREF
OVP
VC
COMP+
COMP–
16
15
14
13
12
11
10
9
VIN_SENSE
SENSE
GATE
INTVCC
EN/UVLO
VIN
DCM
FB
TOP VIEW
17
GND
MSE PACKAGE
16-LEAD PLASTIC MSOP
θJA = 50°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT3798EMSE#PBF LT3798EMSE#TRPBF 3798 16-Lead Plastic MSOP –40°C to 125°C
LT3798IMSE#PBF LT3798IMSE#TRPBF 3798 16-Lead Plastic MSOP –40°C to 125°C
LT3798HMSE#PBF LT3798HMSE#TRPBF 3798 16-Lead Plastic MSOP –40°C to 150°C
LT3798MPMSE#PBF LT3798MPMSE#TRPBF 3798 16-Lead Plastic MSOP –55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
PARAMETER CONDITIONS MIN TYP MAX UNITS
Input Voltage Range 10 38 V
VIN Quiescent Current VEN/UVLO = 0.2V
VEN/UVLO = 1.5V, Not Switching 45 60
70 70 µA
µA
VIN Quiescent Current, INTVCC Overdriven VINTVCC = 11V 60 µA
VIN Shunt Regulator Voltage I = 1mA 40 V
VIN Shunt Regulator Current Limit 8 mA
INTVCC Quiescent Current VEN/UVLO = 0.2V
VEN/UVLO = 1.5V, Not Switching 12.5
1.8 15.5
2.2 17.5
2.7 µA
mA
EN/UVLO Pin Threshold EN/UVLO Pin Voltage Rising l1.21 1.25 1.29 V
EN/UVLO Pin Hysteresis Current EN/UVLO=1V 8 10 12 µA
VIN_SENSE Threshold Turn Off 27 µA
VREF Voltage 0µA Load
200µA Load
l
l
1.97
1.95 2.0
1.98 2.03
2.03 V
V
CTRL1/CTRL2/CTRL3 Pin Bias Current CTLR1/CTRL2/CTRL3 = 1V ±30 nA
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 18V, INTVCC = 11V, unless otherwise noted.
LT3798
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ELECTRICAL CHARACTERISTICS
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3798E is guaranteed to meet specified performance from
0°C to 125°C junction temperature. Specification over the –40°C and
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 18V, INTVCC = 11V, unless otherwise noted.
LT3798I is guaranteed to meet specified performance from –40°C to 125°C
operating junction temperature range. The LT3798H is guaranteed to meet
performance specifications over the –40°C to 150°C operating junction
temperature range. The LT3798MP is guaranteed to meet performance
specifications over the –55°C to 150°C operating junction temperature
range. High junction temperatures degrade operating lifetimes. Operating
lifetime is derated for junction temperatures greater than 125°C.
Note 3: Current flows out of the FB pin.
PARAMETER CONDITIONS MIN TYP MAX UNITS
SENSE Current Limit Threshold VIN_SENSE = 150µA 96 102 107 mV
Minimum SENSE Current Limit Threshold VIN_SENSE = 34µA 14 mV
Minimum SENSE Current Limit Threshold VIN_SENSE = 21µA 4 mV
SENSE Input Bias Current Current Out of Pin, SENSE = 0V 15 µA
FB Voltage l1.22 1.25 1.28 V
FB voltage Line Regulation 10V < VIN < 35V 0.01 0.03 %/V
FB Pin Bias Current (Note 3), FB = 1V 4.05 4.25 4.4 µA
FB Error Amplifier Voltage Gain ΔVVC/ΔVFB, CTRL1=1V, CTRL2=2V, CTRL3=2V 180 V/V
FB Error Amplifier Transconductance ΔI = 5µA 170 UMHOS
Current Error Amplifier Voltage Gain ΔVCOMP+/ΔVCOMP–, CTRL1 = 1V, CTRL2 = 2V, CTRL3 = 2V 100 V/V
Current Error Amplifier Transconductance ΔI = 5µA 50 UMHOS
Current Loop Voltage Gain ΔVCTRL/ΔVSENSE,1000pF Cap from COMP+ to COMP–21 V/V
DCM Current Turn-On Threshold Current Out of Pin 80 µA
Maximum Oscillator Frequency COMP+ = 0.95V, VIN_SENSE = 150µA 150 kHz
Minimum Oscillator Frequency COMP+ = 0V, VFB <VOVP 4 kHz
Minimum Oscillator Frequency COMP+ = 0V, VFB >VOVP 0.5 kHz
Backup Oscillator Frequency 20 kHz
Linear Regulator
INTVCC Regulation Voltage No Load 9.8 10 10.4 V
Dropout (VIN-INTVCC) IINTVCC = –10mA, VIN = 10V 500 900 mV
Current Limit Below Undervoltage Threshold 12 25 mA
Current Limit Above Undervoltage Threshold 80 120 mA
Gate Driver
tr GATE Driver Output Rise Time CL = 3300pF, 10% to 90% 18 ns
tf GATE Driver Output Fall Time CL = 3300pF, 90% to 10% 18 ns
GATE Output Low (VOL) 0.01 V
GATE Output High (VOH) INTVCC-
50mV V
LT3798
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TYPICAL PERFORMANCE CHARACTERISTICS
VREF vs Temperature VREF vs VIN
SENSE Current Limit Threshold
vs Temperature
Minimum Oscillator Frequency
vs Temperature
Backup Oscillator Frequency
vs Temperature
Maximum Oscillator Frequency
vs Temperature
EN/UVLO Threshold
vs Temperature
EN/UVLO Hysteresis Current
vs Temperature VIN IQ vs Temperature
TA = 25°C, unless otherwise noted.
TEMPERATURE (°C)
–50 0 50 75
–25 25 100 125 150
3798 G01
1.2
EN/UVLO (V)
1.22
1.26
1.24
1.28
1.3
RISING
FALLING
TEMPERATURE (°C)
–50 0 50 75
–25 25 100 125 150
3798 G02
10
10.5
EN/UVLO HYSTERESIS CURRENT (µA)
11
11.5
12
TEMPERATURE (°C)
–50 0 50 75
–25 25 100 125 150
3798 G03
0
IQ (µA)
20
10
30
40
60
50
100
90
80
70 VIN = 12V
VIN = 24V
TEMPERATURE (°C)
–50
1.900
VREF (V)
1.925
1.975
2.000
2.025
2.100
2.075
050 75
1.950
2.050
–25 25 100 125 150
3798 G05
VIN = 24V WITH NO LOAD
VIN = 24V WITH 200µA LOAD
VIN (V)
10
1.95
VREF (V)
1.96
1.98
1.99
2
2.01
2.05
2.03
2.04
1.97
2.02
15 25
20 30 35 40
3798 G05
NO LOAD
200µA LOAD
TEMPERATURE (°C)
–50
0
CURRENT LIMIT (mA)
20
60
80
100
120
050 75
40
–25 25 100 125 150
3798 G06
MAX ILIM
MIN ILIM VIN_SENSE = 34µA
MIN ILIM VIN_SENSE = 21µA
TEMPERATURE (°C)
–50
120
FREQUENCY (kHz)
145
195
220
050 75
170
–25 25 100 125 150
3798 G06a TEMPERATURE (°C)
–50
0
FREQUENCY (kHz)
2
3
4
5
050 75
1
–25 25 125100 150
3799 G07
VFB < VOVP
VOVP > VFB
TEMPERATURE (°C)
–50
0
FREQUENCY (kHz)
10
15
20
25
050 75
5
–25 25 125100 150
3798 G07a
LT3798
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TYPICAL PERFORMANCE CHARACTERISTICS
INTVCC vs Temperature INTVCC vs VIN VIN Shunt Voltage vs Temperature
Maximum Shunt Current
vs Temperature
TA = 25°C, unless otherwise noted.
Leakage Inductance Blanking Time
vs SENSE Current Limit Threshold
TEMPERATURE (°C)
–50
9.5
INTVCC (V)
9.75
10.25
10.5
050 75
10
–25 25 100 125 150
3798 G08
25mA LOAD
10mA LOAD
NO LOAD
VIN (V)
5 15 25 30
10 20 35 40
3798 G09
9
INTVCC (V)
9.2
9.4
9.6
9.8
10
10.2
TEMPERATURE (°C)
–50 0 50 75
–25 25 100 125 150
3798 G10
39
VIN SHUNT VOLTAGE (V)
39.5
40
40.5
41
41.5
42 ISHUNT = 1mA
TEMPERATURE (°C)
–50 0 50 75
–25 25 100 125 150
3798 G11
5
SHUNT CURRENT (mA)
6
7
8
9
10
SENSE CURRENT LIMIT THRESHOLD (mV)
060 80
20 40 100 120
3798 G12
0
0.2
LEAKAGE INDUCTANCE BLANKING TIME (µs)
0.4
0.6
0.8
1
1.2
1.4
1.6
1.8
2
Output Voltage
vs Input Voltage
VOUT vs Temperature
Output Current
vs Input Voltage
VIN (VAC)
90
23.6
VOUT (V)
23.8
24.6
24.2
170 210 230
24.4
24
150130110 190 250 270
3798 G13
PAGE 17 SCHEMATIC:
UNIVERSAL
VIN (VAC)
90
0.90
OUTPUT CURRENT (A)
0.95
1.00
1.05
1.10
210 230
130
110 170150 190 250 270
3798 G14
PAGE 17 SCHEMATIC:
UNIVERSAL
VOUT = 22V
TEMPERATURE (°C)
–50
23
VOUT (V)
24
24.5
25
050 75
23.5
–25 25 125100 150
3798 G012a
PAGE 17 SCHEMATIC:
UNIVERSAL
VAC = 120V
VAC = 220V
LT3798
6
3798fa
CTRL1, CTRL2, CTRL3 (Pin 1, Pin 2, Pin 3): Current
Output Adjustment Pins. These pins control the output
current. The lowest value out of the three CTRL inputs is
compared to negative input of the operational amplifier.
VREF (Pin 4): Voltage Reference Output Pin. Typically 2V.
This pin drives a resistor divider for the CTRL pin, either
for analog dimming or for temperature limit/compensation
of output load. Can supply up to 200µA.
OVP (Pin 5): Overvoltage Protection. This pin accepts a
DC voltage to compare to the sample and hold’s voltage
output information. When output voltage information is
above the OVP, the part divides the minimum switching
frequency by 8, around 500Hz. This protects devices con-
nected to the output. This also allows the part to operate
with very little power consumption with no load to meet
energy star requirements.
VC (Pin 6): Compensation Pin for Internal Error Amplifier.
Connect a series RC from this pin to ground to compensate
the switching regulator. A 100pF capacitor in parallel helps
eliminate noise.
COMP+, COMP– (Pin 7, Pin 8): Compensation Pins for
Internal Error Amplifier. Connect a capacitor between these
two pins to compensate the internal feedback loop.
FB (Pin 9): Voltage Loop Feedback Pin. FB is used to
regulate the output voltage by sampling the third wind-
ing. If the converter is used in current mode, the FB pin
will normally be at a voltage level lower than 1.25V, and
will reach the steady state of 1.25V if it detects an open
output condition.
DCM (Pin 10): Discontinuous Conduction Mode Detection
Pin. Connect a capacitor and resistor in series with this
pin to the third winding.
VIN (Pin 11): Input Voltage. This pin supplies current to
the internal start-up circuitry and to the INTVCC LDO. This
pin must be locally bypassed with a capacitor. A 42V shunt
regulator is internally connected to this pin.
EN/UVLO (Pin 12): Enable/Undervoltage Lockout. A resis-
tor divider connected to VIN is tied to this pin to program
the minimum input voltage at which the LT3798 will turn
on. When below 1.25V, the part will draw 60µA with most
of the internal circuitry disabled and a 10µA hysteresis
current will be pulled out of the EN/UVLO pin. When above
1.25V, the part will be enabled and begin to switch and the
10µA hysteresis current is turned off.
INTVCC (Pin 13): Regulated Supply for Internal Loads
and GATE Driver. Supplied from VIN and regulates to 10V
(typical). INTVCC must be bypassed with a 4.7µF capacitor
placed close to the pin.
PIN FUNCTIONS
Power Factor vs Input Voltage Efficiency vs Input Voltage
VIN (VAC)
90
0.90
0.92
0.91
0.95
0.96
0.97
0.98
0.99
1.00
150 210190 230
0.93
0.94
130110 170 250 270
3798 G15
POWER FACTOR
PAGE 17 SCHEMATIC:
UNIVERSAL
VIN (VAC)
90
60
EFFICIENCY (%)
70
80
90
100
150 210190 230130110 170 250 270
3798 G16
PAGE 17 SCHEMATIC:
UNIVERSAL
TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25°C, unless otherwise noted.
LT3798
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PIN FUNCTIONS
BLOCK DIAGRAM
VOUT+
VOUT–
VRECTIFIED
D1
C7
L1A
T1
N:1
L1B
C2
SW1
R1
R3
C3
1.22V
VIN
R14
R13
D2
L1C
•
C1
R15
R4
R5
VIN_SENSE
EN/UVLODCM
STARTUP
INTERNAL REG
VREF
OVP
FB
S&H
COMP+
COMP–
CTRL1
CTRL3
CTRL2
FB
S&H
VC1.22V
C6
C4
R8
R9
R10
+
–A8
–
+A5 A6
MINIMUM
MULTIPLIER
A3
R12
1M
+
–600mV
ONE
SHOT
+
–A2
+
–A1
CURRENT
COMPARATOR
RQ
S
S
MASTER
LATCH
LOW OUTPUT
CURRENT
OSCILLATOR
A4
R11
R7
DRIVER
GND
GATE
INTVCC
SENSE
R6
C5
3798 BD
–
+A7
A9
GATE (Pin 14): N-Channel FET Gate Driver Output. Switches
between INTVCC and GND. Driven to GND during shutdown
state and stays high during low voltage states.
SENSE (Pin 15): The Current Sense Input for the Control
Loop. Kelvin connect this pin to the positive terminal of the
switch current sense resistor, RSENSE, in the source of the
NFET. The negative terminal of the current sense resistor
should be connected to the GND plane close to the IC.
VIN_SENSE (Pin 16): Line Voltage Sense Pin. The pin is
used for sensing the AC line voltage to perform power
factor correction. Connect a resistor in series with the
line voltage to this pin. If no PFC is needed, connect this
pin to INTVCC with a 25k resistor.
GND (Exposed Pad Pin 17): Ground. The exposed pad
of the package provides both electrical contact to ground
and good thermal contact to the printed circuit board.
The exposed pad must be soldered to the circuit board
for proper operation.
LT3798
8
3798fa
The LT3798 is a current mode switching controller IC
designed specifically for generating a constant current/
constant voltage supply in an isolated flyback topology.
The special problem normally encountered in such circuits
is that information relating to the output voltage and cur-
rent on the isolated secondary side of the transformer
must be communicated to the primary side in order to
maintain regulation. Historically, this has been done with
an opto-isolator. The LT3798 uses a novel method of using
the external MOSFETs peak current information from the
sense resistor to calculate the output current of a flyback
converter without the need of an opto-coupler.
Active power factor correction is becoming a requirement
for offline power supplies and the power levels are de-
creasing. A power factor of one is achieved if the current
drawn is proportional to the input voltage. The LT3798
modulates the peak current limit with a scaled version
of the input voltage. This technique can provide power
factors of 0.97 or greater.
The Block Diagram shows an overall view of the system. The
external components are in a flyback topology configura-
tion. The third winding senses the output voltage and also
supplies power to the part in steady-state operation. The
VIN pin supplies power to an internal LDO that generates
10V at the INTVCC pin. The novel control circuitry consists
of two error amplifiers, a minimum circuit, a multiplier,
a transmission gate, a current comparator, a low output
current oscillator and a master latch, which will be ex-
plained in the following sections. The part also features a
sample-and-hold to sample the output voltage from the
third winding. A comparator is used to detect discontinu-
ous conduction mode (DCM) with a cap connected to the
third winding. The part features a 1.9A gate driver.
The LT3798 is designed for both off-line and DC applica-
tions. The EN/UVLO and a resistor divider can be configured
for a micropower hysteretic start-up. In the Block Diagram,
R3 is used to stand off the high voltage supply voltage.
The internal LDO starts to supply current to the INTVCC
when VIN is above 2.5V. The VIN and INTVCC capacitors are
charged by the current from R3. When VIN exceeds the
turn-on threshold and INTVCC is in regulation at 10V, the
OPERATION
part begins to switch. The VIN hysteresis is set by the EN/
UVLO resistor divider. The third winding provides power
to VIN when its voltage is higher than the VIN voltage. A
voltage shunt is provided for fault protection and can sink
8mA of current when VIN is over 40V.
During a typical cycle, the gate driver turns the external
MOSFET on and a current flows through the primary wind-
ing. This current increases at a rate proportional to the
input voltage and inversely proportional to the magnetizing
inductance of the transformer. The control loop determines
the maximum current and the current comparator turns
the switch off when the current level is reached. When the
switch turns off, the energy in the core of the transformer
flows out the secondary winding through the output diode,
D1. This current decreases at a rate proportional to the
output voltage. When the current decreases to zero, the
output diode turns off and voltage across the secondary
winding starts to oscillate from the parasitic capacitance
and the magnetizing inductance of the transformer. Since
all windings have the same voltage across them, the third
winding rings too. The capacitor connected to the DCM
pin, C1, trips the comparator A2, which serves as a dv/dt
detector, when the ringing occurs. This timing information
is used to calculate the output current and will be described
below. The dv/dt detector waits for the ringing waveform
to reach its minimum value and then the switch turns back
on. This switching behavior is similar to zero volt switching
and minimizes the amount of energy lost when the switch
is turned back on and improves efficiency as much as
5%. Since this part operates on the edge of continuous
conduction mode and discontinuous conduction mode,
the operating mode is called critical conduction mode (or
boundary conduction mode).
Primary Side Control Loops
The LT3798 achieves constant current/constant voltage
operation by using two separate error amplifiers. These
two amplifiers are then fed to a circuit that outputs the
lower voltage of the two, shown as the "minimum" block in
the Block Diagram. This voltage is converted to a current
before being fed into the multiplier.
LT3798
9
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Primary Side Current Control Loop
The CTRL1/CTRL2/CTRL3 pins control the output current
of the flyback controller. To simplify the loop, let’s assume
the VIN_SENSE pin is held at a constant voltage above 1V
eliminating the multiplier from the control loop. The error
amplifier, A5, is configured as integrator with the external
capacitor C6. The COMP+ node voltage is converted to a
current into the multiplier with the V/I converter, A6. Since
A7’s output is constant, the output of the multiplier is
proportional to A6 and can be ignored. The output of the
multiplier controls the peak current with its connection to
the current comparator, A1. The output of the multiplier is
also connected to the transmission gate, SW1, and to a
1M resistor. The transmission gate, SW1, turns on when
the secondary current flows to the output capacitor. This
is called the flyback period when the output diode D1 is
on. The current through the 1M resistor gets integrated by
A5. The lowest CTRL input is equal to the negative input
of A5 in steady state.
A current output regulator normally uses a sense resistor
in series with the output current and uses a feedback loop
to control the peak current of the switching converter. In
this isolated case, the output current information is not
available so instead the LT3798 calculates it using the in-
formation available on the primary side of the transformer.
The output current may be calculated by taking the average
of the output diode current. As shown in Figure 1, the diode
current is a triangle waveform with a base of the flyback
time and a height of the peak secondary winding current.
In a flyback topology, the secondary winding current is N
times the primary winding current, where NPS is the primary
to secondary winding ratio. Instead of taking the area of
the triangle, let’s think of it as a pulse width modulation
(PWM) waveform. During the flyback time, the average
current is half the peak secondary winding current and
zero during the rest of the cycle. The equation to express
the output current is:
IOUT = 0.5 • IPK • NPS • D
where D is equal to the percentage of the cycle that the
flyback time represents. The LT3798 has access to the
primary winding current, the input to the current com-
parator, and when the flyback time starts and ends. Now
the output current can be calculated by averaging a PWM
waveform with a height of the current limit and a duty cycle
of the flyback time over the entire cycle. In the feedback
loop described above, the input to the integrator is such
a waveform. The integrator adjusts the peak current until
calculated output current equals the control voltage. If the
calculated output current is low compared to the control
pin, the error amplifier increases the voltage on the COMP+
node thus increasing the current comparator input.
Primary Side Voltage Control
The output voltage is available through the third winding on
the primary side. A resistor divider attenuates the output
voltage for the voltage error amplifier. A sample-and-hold
circuit samples the attenuated output voltage and feeds
it to the error amplifier. The output of the error amplifier
is the V pin. This node needs a capacitor to compensate
the output voltage control loop.
Power Factor Correction
When the VIN_SENSE voltage is connected to a resistor divider
of the supply voltage, the current limit is proportional to
the supply voltage. The minimum of the two error ampli-
fier outputs is multiplied with the VIN_SENSE pin voltage. If
the LT3798 is configured with a fast control loop, slower
changes from the VIN_SENSE pin would not interfere with
the current limit or the output current. The COMP+ pin
would adjust to the changes of the VIN_SENSE. The only
way for the multiplier to function is to set the control loop
to be an order of magnitude slower than the fundamental
frequency of the VIN_SENSE signal. In an offline case, the
OPERATION
Figure 1. Secondary Diode Current and Switch Waveforms
3798 F01
TFLYBACK
TPERIOD
SECONDARY
DIODE CURRENT
SWITCH
WAVEFORM
IPK(sec)
LT3798
10
3798fa
fundamental frequency of the supply voltage is 120Hz so
the control loop unity gain frequency needs to be set less
than approximately 12Hz. Without a large amount of energy
storage on the secondary side, the output current will be
affected by the supply voltage changes, but the DC com-
ponent of the output current will be accurate. For DC input
or non-PFC AC input applications, connect a 25k resistor
from VIN_SENSE to INTVCC instead of the AC line voltage.
Startup
The LT3798 uses a hysteretic start-up to operate from
high offline voltages. A resistor connected to the supply
voltage protects the part from high voltages. This resistor
is connected to the VIN pin on the part and bypassed with
a capacitor. When the resistor charges the VIN pin to a
turn-on voltage set with the EN/UVLO resistor divider and
the INTVCC pin is at its regulation point, the part begins
to switch. The resistor cannot provide power for the part
in steady state, but relies on the capacitor to start-up the
part, then the third winding begins to provide power to the
VIN pin along with the resistor. An internal voltage clamp
is attached to the VIN pin to prevent the resistor current
from allowing VIN to go above the absolute maximum
voltage of the pin. The internal clamp is set at 40V and is
capable of 8mA(typical) of current at room temperature.
Setting the VIN Turn-On and Turn-Off Voltages
A large voltage difference between the VIN turn-on voltage
and the VIN turn-off voltage is preferred to allow time for the
third winding to power the part. The EN/UVLO sets these
two voltages. The pin has a 10µA current sink when the
pins voltage is below 1.25V and 0µA when above 1.25V.
The VIN pin connects to a resistor divider as shown in
Figure 2. The UVLO threshold for VIN rising is:
VIN(UVLO,RISING) =1.25V•R1+R2
( )
R2 +10µA •R1
The UVLO Threshold for VIN Falling is :
VIN(UVLO,FALLING) =1.25V•R1+R2
( )
R2
Programming Output Voltage
The output voltage is set using a resistor divider from
the third winding to the FB pin. From the Block Diagram,
the resistors R4 and R5 form a resistor divider from the
third winding. The FB also has an internal current source
that compensates for the diode drop. This current source
causes an offset in the output voltage that needs to be ac-
counted for when setting the output voltage. The output
voltage equation is:
VOUT = VBG (R4+R5)/(NST • R5)–(VF + (R4 • ITC)/NST)
where VBG is the internal reference voltage, NST is the
winding ratio between the secondary winding and the third
winding, VF is the forward drop of the output rectifying
diode, and ITC is the internal current source for the FB pin.
The temperature coefficient of the diode's forward drop
needs to be the opposite of the term, (R4 • ITC)/NST. By
taking the partial derivative with respect to temperature,
the value of R4 is found to be the following:
R4 = NST(1/(δITC/δT)(δVF/δT))
δITC/δT = 12.4nA/°C
ITC = 4.25µA
where δITC/δT is the partial derivative of the ITC current
source, and δVF/δT is the partial derivative of the forward
drop of the output rectifying diode.
With R4 set with the above equation, the resistor value
for R5 is found using the following:
R5 = (VBG • R4)/(NST(VOUT+VF)+R4 • ITC-VBG)
OPERATION
LT3798
EN/UVLO
GND
R2
R1
VIN
3798 F02
Figure 2. Undervoltage Lockout (UVLO)
LT3798
11
3798fa
Programming Output Current
The maximum output current depends on the supply volt-
age and the output voltage in a flyback topology. With the
VIN_SENSE pin connected to 100µA current source and a DC
supply voltage, the maximum output current is determined
at the minimum supply voltage, and the maximum output
voltage using the following equation:
IOUT(MAX) =2•(1–D)•
N
PS
42 •RSENSE
where
D=VOUT •NPS
VOUT •NPS +VIN
The maximum control voltage to achieve this maximum
output current is 2V • (1-D).
It is suggested to operate at 95% of these values to give
margin for the part’s tolerances.
When designing for power factor correction, the output
current waveform is going to have a half sine wave squared
shape and will no longer be able to provide the above
currents. By taking the integral of a sine wave squared
over half a cycle, the average output current is found to
be half the value of the peak output current. In this case,
the recommended maximum average output current is
as follows:
IOUT(MAX) =2•(1−D)•NPS
42 •RSENSE
•47.5%
where
D=VOUT •NPS
VOUT •NPS +VIN
The maximum control voltage to achieve this maximum
output current is (1-D) • 47.5%.
For control voltages below the maximum, the output cur-
rent is equal to the following equation:
IOUT =CTRL•NPS
42 •RSENSE
The VREF pin supplies a 2V reference voltage to be used
with the control pins. To set an output current, a resistor
divider is used from the 2V reference to one of the control
pins. The following equation sets the output current with
a resistor divider:
R1=R2 2NPS
42 •IOUT •RSENSE
– 1
where R1 is the resistor connected to the VREF pin and the
CTRL pin and R2 is the resistor connected to the CTRL
pin and ground.
Setting VIN_SENSE Resistor
The VIN_SENSE resistor sets the current feeding the internal
multiplier that modulates the current limit for power factor
correction. At the maximum line voltage, VMAX, the current
is set to 360µA. Under this condition, the resistor value is
equal to (VMAX/360µA).
For DC input or non-PFC AC input applications, connect
a 25k resistor from VIN_SENSE to INTVCC instead of the
AC line voltage.
Critical Conduction Mode Operation
Critical conduction mode is a variable frequency switching
scheme that always returns the secondary current to zero
with every cycle. The LT3798 relies on boundary mode and
discontinuous mode to calculate the critical current because
the sensing scheme assumes the secondary current returns
to zero with every cycle. The DCM pin uses a fast current
input comparator in combination with a small capacitor to
detect dv/dt on the third winding. To eliminate false trip-
ping due to leakage inductance ringing, a blanking time of
between 600ns and 2μs is applied after the switch turns off,
depending on the current limit shown in the Leakage In-
ductance Blanking Time vs SENSE Current Limit Threshold
curve in the Typical Performance Characteristics section.
The detector looks for 80μA of current through the DCM
pin due to falling voltage on the third winding when the
secondary diode turns off. This detection is important since
the output current is calculated using this comparator’s
output. This is not the optimal time to turn the switch on
because the switch voltage is still close to VIN + VOUT • NPS
and would waste all the energy stored in the parasitic ca-
pacitance on the switch node. Discontinuous ringing begins
when the secondary current reaches zero and the energy
in the parasitic capacitance on the switch node transfers
OPERATION
LT3798
12
3798fa
OPERATION
to the input capacitor. This is a second-order network
composed of the parasitic capacitance on the switch node
and the magnetizing inductance of the primary winding
of the transformer. The minimum voltage of the switch
node during this discontinuous ring is VIN – VOUT • NPS.
The LT3798 turns the switch back on at this time, during
the discontinuous switch waveform, by sensing when
the slope of the switch waveform goes from negative to
positive using the dv/dt detector. This switching technique
may increase efficiency by 5%.
Sense Resistor Selection
The resistor, RSENSE, between the source of the external
N-channel MOSFET and GND should be selected to provide
an adequate switch current to drive the application without
exceeding the current limit threshold.
For applications without power factor correction, select a
resistor according to:
RSENSE =2(1–D)NPS
IOUT •42 •95%
where
D=VOUT •NPS
VOUT •NPS +VIN
For applications with power factor correction, select a
resistor according to:
RSENSE =
2(1–D)N
PS
IOUT •42 •47.5%
where
D=VOUT •NPS
VOUT •NPS +VIN
Minimum Current Limit
The LT3798 features a minimum current limit of approxi-
mately 18% of the peak current limit. This is necessary
when operating in critical conduction mode since low
current limits would increase the operating frequency to a
very high frequency. The output voltage sensing circuitry
needs a minimum amount of flyback waveform time to
sense the output voltage on the third winding. The time
needed is 350ns. The minimum current limit allows the
use of smaller transformers since the magnetizing primary
inductance does not need to be as high to allow proper
time to sample the output voltage information.
To help improve crossover distortion of the line input
current, a second minimum current limit of 6% becomes
active when the VIN_SENSE current is lower than 27µA. Since
the off-time becomes very short with this lower minimum
current limit, the sample-and-hold is deactivated.
Universal Input
The LT3798 operates over the universal input voltage
range of 90VAC to 265VAC. In the Typical Performance
Characteristics section, the Output Voltage vs VIN and the
Output Current vs VIN graphs, show the output voltage
and output current line regulation for the first application
picture in the Typical Applications section.
Selecting Winding Turns Ratio
Boundary mode operation gives a lot of freedom in selecting
the turns ratio of the transformer. We suggest to keep the
duty cycle low, lower NPS, at the maximum input voltage
since the duty cycle will increase when the AC waveform
decreases to zero volts. A higher NPS increases the output
current while keeping the primary current limit constant.
Although this seems to be a good idea, it comes at the
expense of a higher RMS current for the secondary-side
diode which might not be desirable because of the primary
side MOSFET’s superior performance as a switch. A higher
NPS does reduce the voltage stress on the secondary-side
diode while increasing the voltage stress on the primary-
side MOSFET. If switching frequency at full output load is
kept constant, the amount of energy delivered per cycle by
the transformer also stays constant regardless of the NPS.
Therefore, the size of the transformer remains the same at
practical NPS’s. Adjusting the turns ratio is a good way to
find an optimal MOSFET and diode for a given application.
LT3798
13
3798fa
Switch Voltage Clamp Requirement
Leakage inductance of an offline transformer is high due
to the extra isolation requirement. The leakage inductance
energy is not coupled to the secondary but goes into
the drain node of the MOSFET. This is problematic since
400V and higher rated MOSFETs cannot always handle
this energy by avalanching. Therefore the MOSFET needs
protection. A transient voltage suppressor (TVS) and diode
are recommended for all offline application and connected,
as shown in Figure 3. The TVS device needs a reverse
breakdown voltage greater than (VOUT + VF) • NPS where
VOUT is the output voltage of the flyback converter, VF is
the secondary diode forward voltage, and NPS is the turns
ratio. An RCD clamp can be used in place of the TVS clamp.
period, as well. Similarly, initial values can be estimated
using stated switch capacitance and transformer leakage
inductance. Once the value of the drain node capacitance
and inductance is known, a series resistor can be added
to the snubber capacitance to dissipate power and criti-
cally dampen the ringing. The equation for deriving the
optimal series resistance using the observed periods
(tPERIOD, and tPERIOD(SNUBBED)) and snubber capacitance
(CSNUBBER) is below, and the resultant waveforms are
shown in Figure 4.
OPERATION
Figure 3. TVS & RCD Switch Voltage Clamps
3798 F03
GATE
VSUPPLY
GATE
VSUPPLY
In addition to clamping the spike, in some designs where
short circuit protection is desired, it will be necessary to
decrease the amount of ringing by using an RC snubber.
Leakage inductance ringing is at its worst during a short
circuit condition, and can keep the converter from cycling
on and off by peak charging the bias capacitor. On/off
cycling is desired to keep power dissipation down in the
output diode. Alternatively, a heat sink can be used to
manage diode temperature.
The recommended approach for designing an RC snubber
is to measure the period of the ringing at the MOSFET
drain when the MOSFET turns off without the snubber
and then add capacitance—starting with something in
the range of 100pF—until the period of the ringing is 1.5
to 2 times longer. The change in period will determine
the value of the parasitic capacitance, from which the
parasitic inductance can be determined from the initial
Figure 4. Observed Waveforms at MOSFET Drain when
Iteratively Implementing an RC Snubber
TIME (µs)
0
0
VDRAIN (V)
10
30
40
50
0.20
90
3798 F04
20
0.10
0.05 0.25
0.15 0.30
60
70
80
NO SNUBBER
WITH SNUBBER
CAPACITOR
WITH RESISTOR
AND CAPACITOR
CPAR =CSNUBBER
tPERIOD(SNUBBED)
tPERIOD
2
– 1
LPAR =tPERIOD2
CPAR •4π2
RSNUBBER =LPAR
CPAR
Note that energy absorbed by a snubber will be converted
to heat and will not be delivered to the load. In high volt-
age or high current applications, the snubber may need to
be sized for thermal dissipation. To determine the power
dissipated in the snubber resistor from capacitive losses,
measure the drain voltage immediately before the MOS-
FET turns on and use the following equation relating that
LT3798
14
3798fa
voltage and the MOSFET switching frequency to determine
the expected power dissipation:
PSNUBBER = fSW • CSNUBBER • VDRAIN2/2
Decreasing the value of the capacitor will reduce the dis-
sipated power in the snubber at the expense of increased
peak voltage on the MOSFET drain, while increasing the
value of the capacitance will decrease the overshoot.
Transformer Design Considerations
Transformer specification and design is a critical part of
successfully applying the LT3798. In addition to the usual
list of caveats dealing with high frequency isolated power
supply transformer design, the following information
should be carefully considered. Since the current on the
secondary side of the transformer is inferred by the current
sampled on the primary, the transformer turns ratio must
be tightly controlled to ensure a consistent output current.
A tolerance of ±5% in turns ratio from transformer to
transformer could result in a variation of more than ±5% in
output regulation. Fortunately, most magnetic component
manufacturers are capable of guaranteeing a turns ratio
tolerance of 1% or better. Linear Technology has worked
OPERATION
Table 1. Predesigned Transformers—Typical Specifications, Unless Otherwise Noted
TRANSFORMER
PART NUMBER
SIZE
(L × W × H)
LPRI
(µH)
NPSA
(NP:NS:NA)
RPRI
(mΩ)
RSEC
(mΩ) MANUFACTURER
TARGET
APPLICATION
(VOUT/IOUT)
JA4429 21.1mm × 21.1mm × 17.3mm 400 1:0.24:0.24 252 126 Coilcraft 22V/1A
7508110210 15.75mm × 15mm × 18.5mm 2000 6.67:1:1.67 5100 165 Würth Elektronik 10V/0.4A
750813002 15.75mm × 15mm × 18.5mm 2000 20:1.0:5.0 6100 25 Würth Elektronik 3.8V/1.1A
750811330 43.2mm × 39.6mm × 30.5mm 300 6:1.0:1.0 150 25 Würth Elektronik 18V/5A
750813144 16.5mm × 18mm × 18mm 600 4:1:0.71 2400 420 Würth Elektronik 28V/0.5A
750813134 16.5mm × 18mm × 18mm 600 8:1:1.28 1850 105 Würth Elektronik 14V/1A
750811291 31mm × 31mm × 25mm 400 1:1:0.24 550 1230 Würth Elektronik 85V/0.4A
750813390 43.18mm × 39.6mm × 30.48mm 100 1:1:0.22 150 688 Würth Elektronik 90V/1A
750811290 31mm × 31mm × 25mm 460 1:1:0.17 600 560 Würth Elektronik 125V/0.32A
X-11181-002 23.5mm × 21.4mm × 9.5mm 500 72:16:10 1000 80 Premo 30V/0.5A
750811248 31mm × 31mm × 25mm 300 4:1.0:1.0 280 25 Würth Elektronik 24V/2A
RLLT-1001 25mm × 22.2mm × 16mm 820 16:1.0:4.0 1150 10 Renco 5V/4A
750312872 43.2mm × 39.6mm × 30.5mm 14 1:1:0.8 11 11 Würth Elektronik 28V/4A
with several leading magnetic component manufacturers
to produce predesigned flyback transformers for use with
the LT3798. Table 1 shows the details of several of these
transformers.
Loop Compensation
The voltage feedback loop is a traditional GM error ampli-
fier. The loop cross-over frequency is set much lower than
twice the line frequency for PFC to work properly.
The current output feedback loop is an integrator con-
figuration with the compensation capacitor between the
negative input and output of the operational amplifier.
This is a one-pole system therefore a zero is not needed
in the compensation. For offline applications with PFC,
the crossover should be set an order of magnitude lower
than the line frequency of 120Hz or 100Hz. In a typical
application, the compensation capacitor is 0.1μF.
In non-PFC applications, the crossover frequency may be
increased to improve transient performance. The desired
crossover frequency needs to be set an order of magnitude
below the switching frequency for optimal performance.
LT3798
15
3798fa
OPERATION
MOSFET and Diode Selection
With a strong 1.9A gate driver, the LT3798 can effectively
drive most high voltage MOSFETs. A low Qg MOSFET is
recommended to maximize efficiency. In most applications,
the RDS(ON) should be chosen to limit the temperature rise
of the MOSFET. The drain of the MOSFET is stressed to
VOUT • NPS + VIN during the time the MOSFET is off and
the secondary diode is conducting current. But in most
applications, the leakage inductance voltage spike exceeds
this voltage. The voltage of this stress is determined by the
switch voltage clamp. Always check the switch waveform
with an oscilloscope to make sure the leakage inductance
voltage spike is below the breakdown voltage of the MOS-
FET. A transient voltage suppressor and diode are slower
than the leakage inductance voltage spike, therefore causing
a higher voltage than calculated.
The secondary diode stress may be as much as VOUT + 2
• VIN/NPS due to the anode of the diode ringing with the
secondary leakage inductance. An RC snubber in parallel
with the diode eliminates this ringing, so that the reverse
voltage stress is limited to VOUT + VIN/NPS. With a high
NPS and output current greater than 3A, the IRMS through
the diode can become very high and a low forward drop
Schottky is recommended.
Discontinuous Mode Detection
The discontinuous mode detector uses AC-coupling to
detect the ringing on the third winding. A 22pF capacitor
with a 30k resistor in series is recommended in most
designs. Depending on the amount of leakage inductance
ringing, an additional current may be needed to prevent
false tripping from the leakage inductance ringing. A resis-
tor from INTVCC to the DCM pin adds this current. Up to
an additional 100μA of current may be needed in some
cases. The DCM pin is roughly 0.7V, therefore the resistor
value is selected using the following equation:
R=
10V – 0.7V
I
where I is equal to the additional current into the DCM pin.
Power Factor Correction/Harmonic Content
The LT3798 attains high power factor and low harmonic
content by making the peak current of the main power
switch proportional to the line voltage by using an internal
multiplier. A power factor of >0.97 is easily attainable for
most applications by following the design equations in
this data sheet. With proper design, LT3798 applications
can easily meet most harmonic standards.
Operation Under Light Output Loads
The LT3798 detects output overvoltage conditions by
looking at the voltage on the third winding. The third
winding voltage is proportional to the output voltage when
the main power switch is off and the secondary diode is
conducting current. Sensing the output voltage requires
delivering power to the output. When the output current is
very low, this periodic delivery of output current can exceed
the load current. The OVP pin sets the output overvolt-
age threshold. When the output of the sample-and-hold
is above this voltage, the minimum switching frequency
is divided by 8 as shown in Figure 5. This OVP threshold
needs to be set above 1.35V and should be set out of the
way of output voltage transients. The output clamp point
is set with the following formula:
VOUT = VOVP(R4 + R5)/(NST • R5)–(VF + (R4•ITC)/NST)
The VOVP pin voltage may be provided by a resistor divider
from the VREF pin. This frequency division greatly reduces
the output current delivered to the output but a Zener or
resistor is required to dissipate the remaining output cur-
rent. The Zener diode’s voltage needs to be 5% higher than
the output voltage set by the resistor divider connected to
the FB pin. Multiple Zener diodes in series may be needed
for higher output power applications to keep the Zener’s
temperature within the specification.
LT3798
16
3798fa
Protection from Shorted Output Conditions
During a shorted output condition as shown in Figure 6, the
LT3798 operates at the minimum operating frequency. In
normal operation, the third winding provides power to the
IC, but the third winding voltage is zero during a shorted
condition. This causes the part’s VIN UVLO to shutdown
switching. The part starts switching again when VIN has
reached its turn-on voltage.
OPERATION
Usage with DC Input Voltage
The LT3798 is flexible enough to operate well from low
voltage to very high voltage DC input voltage applications.
When the supply voltage is less than 40V, the startup re-
sistor is not needed and the part's VIN can be connected
directly to the supply voltage. The startup sequence for
voltages higher than 40V is the same as what is described
for high voltage offline supply voltages.
The loop compensation component values can be chosen
to provide faster loop response since the LT3798 does
not have to provide PFC for the slow 50Hz/60Hz AC input
voltage. For DC input applications, connect a 25k resistor
from VIN_SENSE to INTVCC.
V3RD WINDING
20V/DIV
VOUT
10V/DIV
IOUT
1A/DIV
3798 F05
1ms/DIV
Figure 5. Switching Waveforms When Output
Open-Circuits or at Very Light Load Conditions
Figure 6. Switching Waveforms When Output Short-Circuits
VIN
20V/DIV
V3RD WINDING
50V/DIV
IPRI
1A/DIV
3798 F06
100ms/DIV
LT3798
17
3798fa
TYPICAL APPLICATIONS
3798 TA02
VIN_SENSE
VIN DCM
FB
VREF
CTRL2
CTRL1
GATE
SENSE
INTVCC
GND
LT3798
COMP+
VCCOMP–
C10
560µF
× 2
BR1: DIODES, INC. HD06
C8: VISHAY 440LD22-R
D1: CENTRAL SEMICONDUCTOR CMR1U-06M
D2, D3: DIODES INC. BAV20W
D4: CENTRAL SEMICONDUCTOR CMR1U-02M
M1: FAIRCHILD FDPF15N65
T1: COILCRAFT JA4429-AL
Z1: FAIRCHILD SMBJ170A
Z2: CENTRAL SEMICONDUCTOR CMZ5937B
C4
4.7pF
C5
10µF
C8
2.2nF
4:1:1
C7, 0.1µF
R16
20Ω
R17
20Ω
D2
D4
Z1
D1
RS
0.05Ω
R4
499k
R3
499k R8
100k
R7
100k
R15
4.99k C6
22pF
R14
90.9k
CTRL3
EN/UVLO
R9
40.2k
R11
100k
R10
16.5k C3
2.2µF
OVP
R12
221k
C9
4.7µF
R13
2k
D3
Z2
“Y1 CAP”
M1
24V
1A
R6
95.3k
R5
1M
90V
TO 265V
AC
C2
0.1µF
C1
0.068µF
BR1
L1
33mH
L2
800µH
+
Universal Input 24W PFC Bus Converter
3798 TA03
VIN_SENSE
VIN DCM
FB
VREF
CTRL2
CTRL1
GATE
SENSE
INTVCC
GND
LT3798
COMP+
VCCOMP–
C10
1000µF
×2C11
10µF
×2
BR1: DIODES, INC. HD06
C8: VISHAY 440LD22-R
C11: MURATA GRM32ER7YA106KA12L
D1: CENTRAL SEMICONDUCTOR CMR1U-06M
D2, D3: DIODES INC. BAV20W
D4: DIODES INC. SBR20A200CTB
M1: INFINEON IPB60R165CP
T1: WÜRTH ELEKTRONIK 750811248
Z1: FAIRCHILD SMBJ170A
Z2: CENTRAL SEMICONDUCTOR CMZ5937B
C4
22pF
C5
10µF
C8
2.2nF
4:1:1
C3
1µF
C7, 0.1µF
R16
20Ω
R17
47Ω
D2
D4
M1
Z1
Z2
D1
RS
0.03Ω
R4
499k
R3
499k
R8
100k
R7
100k
R15
5.49k
C6
22pF
R14
100k
CTRL3
EN/UVLO
OVP
R9
40.2k
R10
31.6k
R12
221k
R11
100k
R5
2.4M
R6
301k
C9
4.7µF
R13
33k
D3
24V
2A
90V
TO 265V
AC
C2
0.22µF
C1
0.1µF
BR1
L2
27mH
L1
1mH
+
“Y1 CAP”
Universal Input 48W PFC Application
LT3798
18
3798fa
PACKAGE DESCRIPTION
MSOP (MSE16) 0911 REV E
0.53 ± 0.152
(.021 ± .006)
SEATING
PLANE
0.18
(.007)
1.10
(.043)
MAX
0.17 –0.27
(.007 – .011)
TYP
0.86
(.034)
REF
0.50
(.0197)
BSC
16
16151413121110
12345678
9
9
18
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
0.254
(.010) 0° – 6° TYP
DETAIL “A”
DETAIL “A”
GAUGE PLANE
5.23
(.206)
MIN
3.20 – 3.45
(.126 – .136)
0.889 ± 0.127
(.035 ± .005)
RECOMMENDED SOLDER PAD LAYOUT
0.305 ± 0.038
(.0120 ± .0015)
TYP
0.50
(.0197)
BSC
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ± 0.102
(.112 ± .004)
2.845 ± 0.102
(.112 ± .004)
4.039 ± 0.102
(.159 ± .004)
(NOTE 3)
1.651 ± 0.102
(.065 ± .004)
1.651 ± 0.102
(.065 ± .004)
0.1016 ± 0.0508
(.004 ± .002)
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
0.280 ± 0.076
(.011 ± .003)
REF
4.90 ± 0.152
(.193 ± .006)
DETAIL “B”
DETAIL “B”
CORNER TAIL IS PART OF
THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
0.12 REF
0.35
REF
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev E)
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
LT3798
19
3798fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
REVISION HISTORY
REV DATE DESCRIPTION PAGE NUMBER
A 12/12 Added H- and MP-grade parts 2, 3
LT3798
20
3798fa
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com
LINEAR TECHNOLOGY CORPORATION 2012
LT 1212 REV A • PRINTED IN USA
RELATED PARTS
TYPICAL APPLICATION
PART NUMBER DESCRIPTION COMMENTS
LT3799/LT3799-1 Offline Isolated Flyback LED Controller
with Active PFC No Opto-Coupler Required, TRIAC Dimmable, VIN and VOUT Limited Only by
External Components, MSOP-16E
LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No-Opto Flyback , MSOP-16 with High Voltage Spacing
LT3573/LT3574/LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch
LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch
LT3757/LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive
LT3957/LT3958 40V/100V Flyback/Boost Converters Monolithic with Integrated 5A/3.3A Switch
LTC3803/LTC3803-3/LTC3803-5 200kHz/300kHz Flyback Controllers VIN and VOUT Limited Only by External Components
LTC3805/LTC3805-5 Adjustable Frequency Flyback
Controllers VIN and VOUT Limited Only by External Components
3798 TA04
VIN_SENSE
VIN DCM
FB
VREF
CTRL2
CTRL1
GATE
SENSE
INTVCC
GND
LT3798
COMP+
VCCOMP–
C10
10µF
×4
C2: TDK C5750X7S2A106M
C8: VISHAY 440LD22-R
C10: MURATA GRM32ER7YA106KA12L
D1: DIODES INC. DFLS1150
D2, D3: DIODES INC. BAV20W
D4: ON SEMICONDUCTOR MBR20200CT
D5: DIODES INC. DFLS2100
C4
15pF
C5
10µF
C8
2.2nF
C6
22pF
1:1:0.8
C3
0.1µF
C7, 22nF
R17
20Ω
D2
D4
M1
Z1
Z2
“Y1 CAP”
D1
Z3
D5
RS
0.004Ω
R4
24k
TO
INTVCC R8
6.8k
R7
6.8k
R15
5.9k
R14
100k
CTRL3
EN/UVLO
OVP
R9
40.2k
R10
34.8k
R3
16.2k
R12
221k
R11
100k
R5
402k
R6
51.1k
C9
4.7µF
R13
15k
D3
28V
4A
C2
10µF
VIN
20V TO 60V
M1: FAIRCHILD FDP2532
T1: WÜRTH ELEKTRONIK 750312872
Z1: DIODES INC. SMCJ60A
Z2: CENTRAL SEMICONDUCTOR CMZ59398
Z3: FAIRCHILD SMBJ170A
112W Wide DC Input Industrial Power Supply
