REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
a
AD7878
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700 World Wide Web Site: http://www.analog.com
Fax: 617/326-8703 © Analog Devices, Inc., 1997
LC
2
MOS Complete 12-Bit
100 kHz Sampling ADC with DSP Interface
FUNCTIONAL BLOCK DIAGRAM
FEATURES
Complete ADC with DSP Interface, Comprising:
Track/Hold Amplifier with 2 ms Acquisition Time
7 ms A/D Converter
3 V Zener Reference
8-Word FIFO and Interface Logic
72 dB SNR at 10 kHz Input Frequency
Interfaces to High Speed DSP Processors, e.g.,
ADSP-2100, TMS32010, TMS32020
41 ns max Data Access Time
Low Power, 60 mW typ
APPLICATIONS
Digital Signal Processing
Speech Recognition and Synthesis
Spectrum Analysis
High Speed Modems
DSP Servo Control
GENERAL DESCRIPTION
The AD7878 is a fast, complete, 12-bit A/D converter with a
versatile DSP interface consisting of an 8-word, first-in, first-out
(FIFO) memory and associated control logic.
The FIFO memory allows up to eight samples to be digitized
before the microprocessor is required to service the A/D con-
verter. The eight words can then be read out of the FIFO at
maximum microprocessor speed. A fast data access time of
41 ns allows direct interfacing to DSP processors and high
speed 16-bit microprocessors.
An on-chip status/control register allows the user to program the
effective length of the FIFO and contains the FIFO out of
range, FIFO empty and FIFO word count information.
The analog input of the AD7878 has a bipolar range of ±3 V.
The AD7878 can convert full power signals up to 50 kHz and is
fully specified for dynamic parameters such as signal-to-noise
ratio and harmonic distortion.
The AD7878 is fabricated in Linear Compatible CMOS
(LC
2
MOS), an advanced, mixed technology process that com-
bines precision bipolar circuits with low power CMOS logic.
The part is available in four package styles, 28-pin plastic and
hermetic dual-in-line package (DIP), leadless ceramic chip
carrier (LCCC) or plastic leaded chip carrier (PLCC).
PRODUCT HIGHLIGHTS
1. Complete A/D Function with DSP Interface
The AD7878 provides the complete function for digitizing
ac signals to 12-bit accuracy. The part features an on-chip
track/hold, on-chip reference and 12-bit A/D converter. The
additional feature of an 8-word FIFO reduces the high soft-
ware overheads associated with servicing interrupts in DSP
processors.
2. Dynamic Specifications for DSP Users
The AD7878 is fully specified and tested for ac parameters,
including signal-to-noise ratio, harmonic distortion and
intermodulation distortion. Key digital timing parameters
are also tested and specified over the full operating tempera-
ture range.
3. Fast Microprocessor Interface
Data access time of 41 ns is the fastest ever achieved in a
monolithic A/D converter, and makes the AD7878 compat-
ible with all modern 16-bit microprocessors and digital
signal processors.
AD7878–SPECIFICATIONS
–2– REV. A
(VDD = +5 V 6 5%, VCC = +5 V 6 5%, VSS = –5 V 6 5%, AGND = DGND =
0 V, fCLK = 8 MHz. All Specifications TMIN to TMAX, unless otherwise noted.)
J, A K, L, B S
Parameter Versions
1
Versions Version Units Test Conditions/Comments
DYNAMIC PERFORMANCE
2
Signal-to-Noise Ratio (SNR)
3
@ 25°C 70 72 70 dB min V
IN
= 10 kHz Sine Wave, f
SAMPLE
= 100 kHz
T
MIN
to T
MAX
70 71 70 dB min Typically 71.5 dB for 0 < V
IN
< 50 kHz
Total Harmonic Distortion (THD) –80 –80 –78 dB max V
IN
= 10 kHz Sine Wave, f
SAMPLE
= 100 kHz
Typically –86 dB for 0 < V
IN
< 50 kHz
Peak Harmonic or Spurious Noise –80 –80 –78 dB max V
IN
= 10 kHz, f
SAMPLE
= 100 kHz
Typically –86 dB for 0 < V
IN
< 50 kHz
Intermodulation Distortion (IMD)
Second Order Terms –80 –80 –78 dB max fa = 9 kHz, fb = 9.5 kHz, f
SAMPLE
= 50 kHz
Third Order Terms –80 –80 –78 dB max fa = 9 kHz, fb = 9.5 kHz, f
SAMPLE
= 50 kHz
Track/Hold Acquisition Time 2 2 2 µs max See Throughput Rate Section
DC ACCURACY
Resolution 12 12 12 Bits
Minimum Resolution for Which
No Missing Codes are Guaranteed 12 12 12 Bits
Relative Accuracy ±1/2 ±1/4 ±1/2 LSB typ
Differential Nonlinearity ±1/2 ±1/2 ±1/2 LSB typ
Bipolar Zero Error ±6±6±6 LSB max
Positive Full-Scale Error
4
±6±6±6 LSB max
Negative Full-Scale Error
4
±6±6±6 LSB max
ANALOG INPUT
Input Voltage Range ±3±3±3 Volts
Input Current ±550 ±550 ±550 µA max
REFERENCE OUTPUT
5
REF OUT 3 3 3 V nom
REF OUT Error @ 25°C±10 ±10 ±10 mV max
T
MIN
to T
MAX
±15 ±15 ±15 mV max
Reference Load Sensitivity
(∆REF OUT/∆I) ±1±1±1 mV max Reference Load Current Change (0 µA–500 µA).
Reference Load Should Not Be Changed
During Conversion
LOGIC INPUTS
Input High Voltage, V
INH
+2.4 +2.4 +2.4 V min V
CC
= +5 V ± 5%
Input Low Voltage, V
INL
+0.8 +0.8 +0.8 V max V
CC
= +5 V ± 5%
Input Current, I
IN
±10 ±10 ±10 µA max V
IN
= 0 to V
CC
Input Capacitance, C
IN6
10 10 10 pF max
LOGIC OUTPUTS
Output High Voltage, V
OH
+2.7 +2.7 +2.7 V min I
SOURCE
40 µA
Output Low Voltage, V
OL
+0.4 +0.4 +0.4 V max I
SINK
= 1.6 mA
DB11–DB0
Floating State Leakage Current ±10 ±10 ±10 ±10 µA max
Floating State Output Capacitance
6
15 15 15 15 pF max
CONVERSION TIME
7/7.125 7/7.125 7/7.125 µs min/µs max Assuming No External Read/Write Operations
7/9.250 7/9.250 7/9.250 µs min/µs max Assuming 17 External Read/Write Operations
See Internal Comparator Timing Section
POWER REQUIREMENTS
V
DD
+5 +5 +5 V nom ±5% for Specified Performance
V
CC
+5 +5 +5 V nom ±5% for Specified Performance
V
SS
–5 –5 –5 V nom ±5% for Specified Performance
I
DD
13 13 13 mA max CS = DMWR = DMRD = 5 V
I
CC
100 100 100 µA max CS = DMWR = DMRD = 5 V
I
SS
6 6 6 mA max CS = DMWR = DMRD = 5 V
Power Dissipation 95.5 95.5 95.5 mW max Typically 60 mW
NOTES
1
Temperature range as follows: J, K, L versions: 0°C to +70°C; A, B versions: –25°C to +85°C; S version: –55°C to +125°C.
2
V
IN
= ±3 V. See Dynamic Specifications section.
3
SNR calculation includes distortion and noise components.
4
Measured with respect to the Internal Reference.
5
For capacitive loads greater than 50 pF a series resistor is required (see Internal Reference section).
6
Sample tested @ +25°C to ensure compliance.
Specifications subject to change without notice.
AD7878
–3–
REV. A
Limit at T
MIN
, T
MAX
Limit at T
MIN
, T
MAX
Limit at T
MIN
, T
MAX
Parameter (L Grade) (J, K, A, B Grades) (S Grade) Units Conditions/Comments
t
l
65 65 75 ns max CLK IN to BUSY Low Propagation Delay
t
2
65 65 75 ns max CLK IN to BUSY High Propagation Delay
t
3
2 CLK IN Cycles 2 CLK IN Cycles 2 CLK IN Cycles min CONVST Pulse Width
t
4
0 0 0 ns min CS to DMRD/REGISTER ENABLE Setup Time
t
5
0 0 0 ns min CS to DMRD/ REGISTER ENABLE Hold Time
t
6
45 60 60 ns min DMRD Pulse Width
50 50 50 µs max
t
7
16 16 16 ns min ADD0 to DMRD/REGISTER ENABLE Setup Time
t
8
0 0 0 ns min ADD0 to DMRD/REGISTER ENABLE Hold Time
t
92
41 57 57 ns min Data Access Time after DMRD
t
103
5 5 5 ns min Bus Relinquish Time
45 45 45 ns max
t
11
42 42 55 ns min REGISTER ENABLE Pulse Width
50 50 50 µs max
t
12
20 20 30 ns min Data Valid to REGISTER ENABLE Setup Time
t
13
10 10 10 ns min Data Hold Time after REGISTER ENABLE
t
142
41 57 57 ns min Data Access Time after BUSY
t
RESET
2 CLK IN Cycles 2 CLK IN Cycles 2 CLK IN Cycles min RESET Pulse Width
NOTES
1
Timing Specifications in bold print are 100% production tested. All other times are sample tested at +25°C to ensure compliance. All input signals are specified with
tr = tf = 5 ns (10% to 90% of 5 V) and timed from a voltage level of 1.6 V.
2
t
9
and t
14
are measured with the load circuits of Figure 1 and defined as the time required for an output to cross 0.8 V or 2.4 V.
3
t
10
is defined as the time required for the data lines to change 0.5 V when loaded with the circuits of Figure 2.
Specifications subject to change without notice.
(VDD = 5 V 6 5%, VCC = 5 V 6 5%, VSS = –5 V 6 5%)
TIMING CHARACTERISTICS
1
Figure 1. Load Circuits for Access Time
Figure 2. Load Circuits for Output Float Delay
ABSOLUTE MAXIMUM RATINGS*
(T
A
= +25°C unless otherwise stated)
V
DD
to DGND . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
V
CC
to DGND . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
V
SS
to DGND . . . . . . . . . . . . . . . . . . . . . . . . . +0.3 V to –7 V
V
DD
to V
CC
. . . . . . . . . . . . . . . . . . . . . . . . . .–0.3 V to +0.3 V
AGND to DGND . . . . . . . . . . . . . . . . . –0.3 V to V
DD
+0.3 V
V
IN
to AGND . . . . . . . . . . . . . . . . . . . . . . . . . –15 V to +15 V
REF OUT to AGND . . . . . . . . . . . . . . . . . . . . . . . . . 0 to V
DD
Digital Inputs to DGND
CLK IN, DMWR, DMRD, RESET,
CS, CONVST, ADD0 . . . . . . . . . . . . –0.3 V to V
DD
+0.3 V
Digital Outputs to DGND
ALFL, BUSY . . . . . . . . . . . . . . . . . . –0.3 V to V
DD
+0.3 V
Data Pins
DB11–DB0 . . . . . . . . . . . . . . . . . . . . –0.3 V to V
DD
+0.3 V
Operating Temperature Range
J, K, L Versions . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C
A, B Versions . . . . . . . . . . . . . . . . . . . . . . . –25°C to +85°C
S Version . . . . . . . . . . . . . . . . . . . . . . . . . –55°C to +125°C
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . .+300°C
Power Dissipation (Any Package) to +75°C . . . . . . 1000 mW
Derates above +75°C by . . . . . . . . . . . . . . . . . . 10 mW/°C
*Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. These are stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
sections of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability
WARNING!
ESD SENSITIVE DEVICE
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD7878 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
a. High-Z to V
OH
b. High-Z to V
OL
a. V
OH
to High-Z b. V
OL
to High-Z
AD7878
–4– REV. A
PIN FUNCTION DESCRIPTION
Pin Pin
Number Mnemonic Function
11 ADD0 Address Input. This control input determines whether the word placed on the output data bus during a read operation is a data
word from the FIFO RAM or the contents of the status/control register. A logic low accesses the data word from Location 0 of
the FIFO while a logic high selects the contents of the register (see Status/Control Register section).
12CS Chip Select. Active low logic input. The device is selected when this input is active.
13DMWR Dam Memory Write. Active low logic input. DMWR is used in conjunction with CS low and ADD0 high to write data to the
status/control register. Corresponds to DMWR (ADSP-2100), R/W (MC68000, TMS32020), WE (TMS32010).
14DMRD Data Memory READ. Active low logic input. DMRD is used in conjunction with CS low to enable the three-state output buffers.
Corresponds directly to DMRD (ADSP-2100), DEN (TMS32010).
15BUSY Active Low Logic Output. This output goes low when the ADC receives a CONVST pulse and remains low until the track/hold
has gone into its hold mode. The three-state drivers of the AD7878 can be disabled while the BUSY signal is low (see Extended
READ/WRITE section). This is achieved by writing a logic 0 to DB5 (DISO) of the status/control register. Writing a logic 1 to
DB5 of the status/control register allows data to be accessed from the AD7878 while BUSY is low.
16ALFL FIFO Almost Full. A logic low indicates that the word count (i.e., number of conversion results) in the FIFO memory has
reached the programmed word count in the status/control register. ALFL is updated at the end of each conversion. The ALFL
output is reset to a logic high when a word is read from the FIFO memory and the word count is less than the preprogrammed
word count. It can also be set high by writing a logic 1 to DB7 (ENAF) of the status/control register.
17 DGND Digital Ground. Ground reference for digital circuitry.
18V
CC
Digital supply voltage, +5 V ± 5%. Positive supply voltage for digital circuitry.
19 DB11 Data Bit 11 (MSB). Three-state TTL output. Coding for the data words in FIFO RAM is twos complement.
10–15 DB10–DB5 Data Bit 10 to Data Bit 5. Three-state TTL input/outputs.
16–19 DB4–DB1 Data Bit 4 to Data Bit 1. Three-state TTL outputs.
20 DB0 Data Bit 0 (LSB). Three-state TTL output.
21 V
DD
Analog positive supply voltage, +5 V ± 5%.
22 AGND Analog Ground. Ground reference for track/hold, reference and DAC.
23 REF OUT Voltage Reference Output. The internal 3 V analog reference is provided at this pin. The external load capability of the reference
is 500 µA.
24 V
IN
Analog Input. Analog input range is ±3 V.
25 V
SS
Analog negative supply voltage, –5 V ± 5%.
26 CONVST Convert Start. Logic input. A low to high transition on this input puts the track/hold into its hold mode and starts conversion.
The CONVST input is asynchronous to CLK IN and independent of CS, DMWR and DMRD.
27 RESET Reset. Active low logic input. A logic low sets the words in FIFO memory to 1000 0000 0000 and resets the ALFL output and
status/control register.
28 CLK IN Clock Input. TTL-compatible logic input. Used as the clock source for the A/D converter. The mark-space ratio of this clock can
vary from 35/65 to 65/35.
PIN CONFIGURATIONS
LCCC
PLCC
DIP
AD7878
–5–
REV. A
ORDERING GUIDE
Signal- Data
Temperature to-Noise Access Package
Model
1, 2
Range Ratio Time Options
3
AD7878JN 0°C to +70°C 70 dB 57 ns N-28
AD7878AQ –25°C to +85°C 70 dB 57 ns Q-28
AD7878SQ –55°C to +125°C 70 dB 57 ns Q-28
AD7878KN 0°C to +70°C 72 dB 57 ns N-28
AD7878BQ –25°C to +85°C 72 dB 57 ns Q-28
AD7878LN 0°C to +70°C 72 dB 41 ns N-28
AD7878SE
4
–55°C to +125°C 70 dB 57 ns E-28A
AD7878JP 0°C to +70°C 70 dB 57 ns P-28A
AD7878KP 0°C to +70°C 72 dB 57 ns P-28A
AD7878LP 0°C to +70°C 72 dB 41 ns P-28A
NOTES
1
To order MIL-STD-883, Class B processed parts, add /883B to part number.
Contact our local sales office for military data sheet.
2
Analog Devices reserves the right to ship either ceramic (D-28) packages or
cerdip (Q-28) hermetic packages.
3
E = Leadless Ceramic Chip Carrier; N = Plastic DIP; P = Plastic Leaded Chip
Carrier, Q = Cerdip.
4
Available to /883B processing only.
STATUS/CONTROL REGISTER
The status/control register serves the dual function of providing
control and monitoring the status of the FIFO memory. This
register is directly accessible through the data bus (DB11–DB0)
with a read or write operation while ADD0 is high. A write
operation to the status/control register provides control for the
ALFL output, bus interface and FIFO counter reset. This is
normally done on power-up initialization. The FIFO memory
address pointer is incremented after each conversion and com-
pared with a preprogrammed count in the status/control regis-
ter. When this preprogrammed count is reached, the ALFL
output is asserted if the ENAF control bit is set to zero. This
ALFL can be used to interrupt the microprocessor after any
predetermined number of conversions (between 1 and 8). The
status of the address pointer along with sample overrange and
ALFL status can be accessed at any time by reading the status/
control register. Note: reading the status/control register does
not cause any internal data movement in the FIFO memory.
Status information for a particular word should be read from the
status register before the data word is read from the FIFO
memory.
STATUS/CONTROL REGISTER FUNCTION
DESCRIPTION
DB11 (ALFL)
Almost Full Flag, Read only. This is the same as Pin 6 (ALFL
output) status. A logic low indicates that the word count in
the FIFO memory has reached the preprogrammed count in bit
locations DB10–DB8. ALFL is updated at the end of conversion.
DB10–DB8 (AFC2–AFC0)
Almost Full Word Count, Read/Write. The count value deter-
mines the number of words in the FIFO memory, which will
cause ALFL to be set. When the FIFO word count equals the
programmed count in these three bits, both the ALFL output
and DB11 of the status register are set to a logic low. For ex-
ample, when a code of 011 is written to these bits, ALFL is set
when Location 0 through Location 3 of the FIFO memory
contains valid data. AFC2 is the most significant bit of the word
count. The count value can be read back if required.
DB7 (ENAF)
Enable Almost Full, Read/Write. Writing a 1 to this bit disables
the ALFL output and status register bit DB11.
DB6 (FOVR/RESET)
FIFO Overrun/RESET, Read/Write. Reading a 1 from this bit
indicates that at least one sample has been discarded because
the FIFO memory is full. When the FIFO is full (i.e., contains
eight words) any further conversion results will be lost. Writing
a 1 to this bit causes a system RESET as per the RESET input
(Pin 27).
DB5 (FOOR/DISO)
FIFO Out of RANGE/Disable Outputs, Read/Write. Reading a
1 from this bit indicates that at least one sample in the FIFO
memory is out of range. Writing a 0 to this bit prevents the data
bus from becoming active while BUSY is low, regardless of the
state of CS and DMRD.
DB4 (FEMP)
FIFO Empty, Read Only. Reading a 1 indicates there are no
samples in the FIFO memory. When the FIFO is empty the
internal ripple-down effects of the FIFO are disabled and fur-
ther reads will continue to access the last valid data word in
Location 0.
DB3 (SOOR)
Sample out of Range, Read Only. Reading a 1 indicates the next
sample to be read is out of range, i.e., the sample in Location 0
of the FIFO.
DB–DB0 (FCN2–FCN0)
FIFO Word Count, Read Only. The value read from these bits
indicates the number of samples in the FIFO memory. For
example, reading 011 from these bits indicates that Location 0
through Location 3 contains valid data. Note: reading all 0s
indicates there is either one word or no word in the FIFO
memory; in this case the FIFO Empty determines if there is no
word in memory. FCN2 is the most significant bit.
Table I. Status/Control Bit Function Description
BIT LOCATION DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3 DB2 DB1 DB0
STATUS INFORMATION (READ) ALFL AFC2 AFC1 AFC0 ENAF FOVR FOOR FEMP SOOR FCN2 FCN1 FCN0
CONTROL FUNCTION (WRITE) X AFC2 AFC1 AFC0 ENAF RESET DISO XXXXX
RESET STATUS 1 0 0 0 0 0 0 1 0 0 0 0
X =DON’T CARE
AD7878
–6– REV. A
INTERNAL FIFO MEMORY
The internal FIFO memory of the AD7878 consists of eight
memory locations. Each word in memory contains 13 bits of
information—12 bits of data from the conversion result and one
additional bit which contains information as to whether the 12-
bit result is out of range or not. A block diagram of the AD7878
FIFO architecture is shown in Figure 3.
Figure 3. Internal FIFO Architecture
The conversion result is gathered in the successive approxima-
tion register (SAR) during conversion. At the end of conversion
this result is transferred to the FIFO memory. The FIFO ad-
dress pointer always points to the top of memory, which is the
uppermost location containing valid data. The pointer is incre-
mented after each conversion. A read operation from the FIFO
memory accesses data from the bottom of the FIFO, Location 0.
On completion of the read operation, each data word moves
down one location and the address pointer is decremented by
one. Therefore, each conversion result from the SAR enters at
the top of memory, propagates down with successive reads until
it reaches Location 0 from where it can be accessed by a micro-
processor read operation.
The transfer of information from the SAR to the FIFO occurs in
synchronization with the AD7878 input clock (CLK IN). The
propagation of data words down the FIFO is also synchronous
with this clock. As a result, a read operation to obtain data from
the FIFO must also be synchronous with CLK IN to avoid
Read/Write conflicts in the FIFO (i.e., reading from FIFO Loca-
tion 0 while it is being updated). This requires that the micro-
processor clock and the AD7878 CLK IN are derived from the
same source.
INTERNAL COMPARATOR TIMING
The ADC clock, which is applied to CLK IN, controls the suc-
cessive approximation A/D conversion process. This clock is
internally divided by four to yield a bit trial cycle time of 500 ns
min (CLK IN = 8 MHz clock). Each bit decision occurs 25 ns
after the rising edge of this divided clock. The bit decision is
latched by the rising edge of an internal comparator strobe sig-
nal. There are 12-bit decisions, as in a normal successive ap-
proximation routine, and one extra decision that checks if the
input sample is out of range. In a normal successive approxima-
tion A/D converter, reading data from the device during conver-
sion can upset the conversion in progress. This is due to on-chip
transients, generated by charging or discharging the databus,
concurrent with a bit decision. The scheme outlined below and
shown in Figure 4 describes how the AD7878 overcomes this
problem.
The internal comparator strobe on the AD7878 is gated with
both DMRD and DMWR so that if a read or write operation
occurs when a bit decision is about to be made, the bit decision
point is deferred by one CLK IN cycle. In other words, if
DMRD or DMWR goes low (with CS low) at any time during
the CLK IN low time immediately prior to the comparator
strobing edge (t
LOW
of Figure 4), the bit trial is suspended for a
clock cycle. This makes sure that the bit decision is latched at a
time when the AD7878 is not attempting to charge or discharge
the data bus, thereby ensuring that no spurious transients occur
internally near a bit decision point.
The decision point slippage mechanism is shown in Figure 4 for
the MSB decision. Normally, the MSB decision occurs 25 ns
after the fourth rising CLK IN edge after CONVST goes high.
However, in the timing diagram of Figure 4, CS and DMRD or
DMWR are low in the time period t
LOW
prior to the MSB deci-
sion point on the fourth rising edge. This causes the internal
comparator strobe to be slipped to the fifth rising clock edge.
The AD7878 will again check during a period t
LOW
prior to this
fifth rising clock edge; and if the CS and DMRD or DMWR are
still low, the bit decision point will be slipped a further clock
cycle.
The conversion time for the ADC normally consists of the 13-
bit trials described above and one extra internal clock cycle during
which data is written from the SAR to the FIFO. For an 8 MHz
input clock this results in a conversion time of 7 µs. However,
the software routine servicing the AD7878 has the potential to
read 16 times from the device during conversion—8 reads from
the FIFO and 8 reads from the status/control register. It also has
the potential to write once to the status/control register. If these
Figure 4. Operational Timing Diagram
AD7878
–7–
REV. A
17 (16 read plus 1 write) operations all occur during t
LOW
time
periods, the conversion time will slip by 17 CLK IN cycles.
Therefore, if read or write operations can occur during t
LOW
periods, it means that the conversion time for the ADC can vary
from 7 µs to 9.12 µs (assuming 8 MHz CLK IN). This calcula-
tion assumes there is a slippage of one CLK IN cycle for each
read or write operation.
INITIATING A CONVERSION
Conversion is initiated on the AD7878 by asserting the CONVST
input. This CONVST input is an asynchronous input indepen-
dent of either the ADC or DSP clocks. This is essential for applica-
tions where precise sampling in time is important. In these applica-
tions the signal sampling must occur at exactly equal intervals to
minimize errors due to sampling uncertainty or jitter. In these cases
the CONVST input is driven from a tamer or some precise clock
source. On receipt of a CONVST pulse, the AD7878 acknowl-
edges by taking the BUSY output low. This BUSY output can be
used to ensure no bus activity while the track/hold goes from track
to hold mode (see Extended Read/Write section). The CONVST
input must stay low for at least two CLK IN periods. The track/
hold amplifier switches from the track to hold mode on the rising
edge of CONVST and conversion is also initiated at this point.
The BUSY output returns high after the CONVST input goes high
and the ADC begins its successive approximation routine. Once
conversion has been initiated another conversion start should not
be attempted until the full conversion cycle has been completed.
Figure 5 shows the taming diagram for the conversion start.
In applications where precise sampling is not critical, the
CONVST pulse can be generated from a microprocessor WR
or RD line gated with a decoded address (different from the
AD7878 CS address). Note that the CONVST pulse width
must be a minimum of two AD7878 CLK IN cycles.
Figure 5. Conversion Start Timing Diagram
READ/WRITE OPERATIONS
The AD7878 read/write operations consist of reading from the
FIFO memory and reading and writing from the status/control
register. These operations are controlled by the CS, DMRD,
DMWR and ADD0 logic inputs. A description of these operations
is given in the following sections. In addition to the basic read/write
operations there is an extended read/write operation. This can
occur if a read/write operation occurs during a CONVST pulse.
This extended read/write is intended for use with microproces-
sors that can be driven into a WAIT state, and the scheme is
recommended for applications where an external timer controls
the CONVST input asynchronously to the microprocessor read/
write operations.
Basic Read Operation
Figure 6 shows the timing diagram for a basic read operation on
the AD7878. CS and DMRD going low accesses data from
either the status/control register or the FIFO memory. A read
operation with ADD0 low accesses data from the FIFO while a
read with ADD0 high accesses data from the status/ control
register.
Figure 6. Basic Read Operation
Basic Write Operation
A basic write operation to the AD7878 status/control register
consists of bringing CS and DMWR low with ADD0 high. In-
ternally these signals are gated with CLK IN to provide an
internal REGISTER ENABLE signal (see Figure 7). The pulse
width of this REGISTER ENABLE signal is effectively the
overlap between the CLK IN low time and the DMWR pulse.
This may result in shorter write pulse widths, data setup times
and data hold times than those given by the microprocessor.
The timing on the AD7878 timing diagram of Figure 8 is there-
fore given with respect to the internal REGISTER ENABLE
signal rather than the DMWR signal.
Figure 7.
DMWR
Internal Logic
Figure 8. Basic Write Operation
AD7878
–8– REV. A
Extended Read/Write Operation
As described earlier, a read/write operation to the AD7878 can
cause spurious on-chip transients. Should these transients occur
while the track/hold is going from track to hold mode, it may
result in an incorrect value of V
IN
being held by the track/hold
amplifier. Because the CONVST input has asynchronous capa-
bility, a read/write operation could occur while CONVST is
low. The AD7878 allows the read/write operation to occur but
has the facility to disable its three-state drivers so that there is
no data bus activity and, hence, no transients while the track/
hold goes from track to hold.
Writing a logic 0 to DB5 (DISO) of the status/control register
prevents the output latches from being enabled while the
AD7878 BUSY signal is low. If a microprocessor read/write
operation can occur during the BUSY low time, the BUSY
should be gated with CS of the AD7878 and this gated signal
used to stretch the instruction cycle using DMACK (ADSP-
2100), READY (TMS32020) or DTACK (68000).
When CONVST goes low, the AD7878 acknowledges it by
bringing BUSY low on the next rising edge of CLK IN. With a
logic 0 in DB5, the AD7878 data bus cannot now be enabled. If
a read/write operation now occurs, the BUSY and CS gated
signal drives the microprocessor into a WAIT state, thereby
extending the read/write operation. BUSY goes high on the
second rising edge of CLK IN after CONVST goes high. The
AD7878 data outputs are now enabled and the microprocessor
is released from its WAIT state, allowing it to complete its read/
write operation to the AD7878.
The microprocessor cycle time for the read/write operation is
extended by the CONVST pulse width plus two CLK IN peri-
ods worst case. This is the maximum length of time for which
BUSY can be low. Assuming a CONVST pulse width of two
CLK IN periods and an 8 MHz CLK IN, the instruction cycle
is extended by 500 ns maximum. Figure 9 shows the timing
diagram for an extended read operation. In a similar manner, a
write operation will be extended if it occurs during a CONVST
pulse.
For processors that cannot be forced into a WAIT state, writing
a logic 1 into DB5 of the status/control register allows the out-
put latches to be enabled while BUSY is low. In this case BUSY
still goes low as before, but it would not be used to stretch the
read/write cycle and the instruction cycle continues as normal
(see Figures 6 and 8).
Figure 9. Extended Read Operation
AD7878 DYNAMIC SPECIFICATIONS
The AD7878 is specified and 100% tested for dynamic perfor-
mance specifications rather than for traditional dc specifications
such as Integral and Differential Nonlinearity. These ac specifi-
cations provide information on the AD7878’s effect on the spec-
tral content of the input signal. Hence, the parameters for which
the AD7878 is specified include SNR, Harmonic Distortion, inter-
modulation Distortion and Peak Harmonics. These terms are dis-
cussed in more detail in the following sections.
Signal-to-Noise Ratio (SNR)
SNR is the measured signal-to-noise ratio at the output of the
ADC. The signal is the rms magnitude of the fundamental.
Noise is the rms sum of all the nonfundamental signals (excluding
dc) up to half the sampling frequency (f
S
/2). SNR is dependent
upon the number of quantization levels used in the digitization
process; the more levels, the smaller the quantization noise. The
theoretical signal-to-noise ratio for a sine wave input is given by
SNR = (6.02 N + 1.76) dB (1)
where N is the number of bits. Thus for an ideal 12-bit con-
verter, SNR = 74 dB.
The output spectrum from the ADC is evaluated by applying a
sine-wave signal of very low distortion to the V
IN
input, which is
sampled at a 100 kHz sampling rate. A Fast Fourier Transform
(FFT) plot is generated from which the SNR data can be ob-
tained. Figure 10 shows a typical 2048 point FFT plot of the
AD7878KN with an input signal of 25 kHz and a sampling
frequency of 100 kHz. The SNR obtained from this graph is
72.6 dB. It should be noted that the harmonics are included in
the SNR calculation.
Figure 10. AD7878 FFT Plot
Effective Number of Bits
The formula given in (1) relates the SNR to the number of bits.
Rewriting the formula, as in (2), it is possible to get a measure of
performance expressed in effective number of bits (N). The
effective number of bits for a device can be calculated directly
from its measured SNR.
N=SNR –1.76
6.02
(2)
AD7878
–9–
REV. A
Figure 11 shows a typical plot of effective number of bits versus
frequency for an AD7878KN with a sampling frequency of
100 kHz. The effective number of bits typically falls between
11.7 and 11.85 corresponding to SNR figures of 72.2 and
73.1 dB.
Figure 11. Effective Number of Bits vs. Frequency
Harmonic Distortion
Harmonic Distortion is the ratio of the rms sum of harmonics to
the fundamental. For the AD7878, Total Harmonic Distortion
(THD) is defined as:
THD =20 log
(
V
22
+V
32
+V
42
+V
52
+V
62
)
V
1
where V
1
is the rms amplitude of the fundamental and V
2
, V
3
,
V
4
, V
5
and V
6
are the rms amplitudes of the second to the sixth
harmonic. The THD is also derived from the FFT plot of the
ADC output spectrum.
Intermodulation Distortion
With inputs consisting of sine waves at two frequencies, fa and
fb, any active device with nonlinearities will create distortion
products at sum and difference frequencies of mfa + nfb where
m, n = 0, 1, 2, 3 . . . . , etc. Intermodulation terms are those for
which neither m nor n is equal to zero. For example, the second
order terms include (fa + fb) and (fa – fb) while the third order
terms include (2fa + fb), (2fa – fb), (fa + 2fb) and (fa – 2fb).
Using the CCIF standard, where two input frequencies near the
top end of the input bandwidth are used, the second and third
order terms are of different significance. The second order terms
are usually distanced in frequency from the original sine waves,
while the third order terms are usually at a frequency close to the
input frequencies. As a result, the second and third order terms
are specified separately. The calculation of the intermodulation
distortion is as per the THD specification where it is the ratio of
the rms sum of the individual distortion products to the rms am-
plitude of the fundamental expressed in dBs.
Intermodulation distortion is calculated using an FFT algorithm
but, in this case, the input consists of two equal amplitude, low
distortion sine waves. Figure 12 shows a typical IMD plot for
the AD7878.
Peak Harmonic or Spurious Noise
Peak harmonic or spurious noise is defined as the ratio of the
rms value of the next largest component in the ADC output
spectrum (up to FS/2 and excluding dc) to the rms value of the
fundamental. Normally, the value of this specification will be
determined by the largest harmonic in the spectrum, but for
parts where the harmonics are buried in the noise floor the
largest peak will be a noise peak.
Figure 12. AD7878 IMD Plot
Histogram Plot
When a sine wave of a specified frequency is applied to the V
IN
input of the AD7878 and several million samples are taken, it is
possible to plot a histogram showing the frequency of occur-
rence of each of the 4096 ADC codes. If a particular step is
wider than the ideal 1 LSB width, then the code associated with
that step will accumulate more counts than for the code for an
ideal step. Likewise, a step narrower than ideal will have fewer
counts. Missing codes are easily seen in the histogram plot because
a missing code means zero counts for a particular code. Large
spikes in the plot indicate large differential nonlinearity.
Figure 13 shows a histogram plot for the AD7878KN with a
sampling frequency of 100 kHz and an input frequency of
25 kHz. For a sine-wave input, a perfect ADC would produce a
cusp probability density function described by the equation:
p(V)=1
π(A2–V2)
where A is the peak amplitude of the sine wave and p (V) is the
probability of occurrence at a voltage V. The histogram plot of
Figure 13 corresponds very well with this cusp shape. The ab-
sence of large spikes in this plot indicates small dynamic differ-
ential nonlinearity (the largest spike in the plot represents less
than 1/4 LSB of DNL error). The AD7878 has no missing
codes under these conditions since no code records zero counts.
Figure 13. AD7878 Histogram Plot
AD7878
–10– REV. A
CONVERSION TIMING
The track-and-hold on the AD7878 goes from track-to-hold
mode on the rising edge of CONVST, and the value of V
IN
at
this point is the value which will be converted. However, the
conversion actually sorts on the next rising edge of CLK IN
after CONVST goes high. If CONVST goes high within ap-
proximately 30 ns prior to a rising edge of CLK IN, that CLK
IN edge will not be seen as the first CLK IN edge of the con-
version process, and conversion will not actually start until one
CLK IN cycle later. As a result, the conversion time (from
CONVST to FIFO update) will vary by one clock cycle de-
pending on the relationship between CONVST and CLK IN.
A conversion cycle normally consists of 56 CLK IN cycles
(assuming no read/write operations) which corresponds to a 7
As conversion time. If CONVST goes high within 30 ns prior
to a rising edge of CLK IN, the conversion time will consist of
57 CLK IN cycles, i.e., 7.125 µs. This effect does not cause
track/hold jitter.
INTERNAL REFERENCE
The AD7878 has an on-chip temperature compensated buried
Zener reference (see Figure 14) that is factory trimmed to 3 V
± 1%. Internally, it provides both the DAC reference and the
dc bias required for bipolar operation. The reference output is
available (REF OUT) and is capable of providing up to 500 µA
to an external load.
Figure 14. AD7878 Reference Circuit
The maximum recommended capacitance on REF OUT for
normal operation is 50 pF. If the reference is required for use
external to the AD7878, it should be decoupled with a
200 Ω resistor in series with a parallel combination of a 10 µF
tantalum capacitor and a 0.1 µF ceramic capacitor. These
decoupling components are required to remove voltage spikes
caused by the internal operation of the AD7878.
TRACK-AND-HOLD AMPLIFIER
The track-and-hold amplifier on the analog input of the
AD7878 allows the ADC to accurately convert an input sine
wave of 6 V peak-peak amplitude to 12-bit accuracy. The input
bandwidth of the track/hold amplifier is much greater than the
Nyquist rate of the ADC even when operated at its minimum
conversion time. The 0.1 dB cutoff frequency occurs typically
at 500 kHz. The track/hold amplifier acquires an input signal to
12-bit accuracy in less than 2 µs.
The operation of the track/hold amplifier is transparent to the
user. The track/hold amplifier goes from its tracking mode to
its hold mode at the start of conversion on the rising edge of
CONVST and returns to track mode at the end of conversion.
ANALOG INPUT
Figure 15 shows the AD7878 analog input. The analog input
range is ±3 V into an input resistance of typically 15 kΩ. The
designed code transitions occur midway between successive
integer LSB values (i.e., 1/2 LSB, 3/2 LSBs, 5/2 LSBs . . . .
FS–3/2 LSBs). The output code is 2s complement binary with
1 LSB = FS/4096 = 6 V/4096 = 1.46 mV. The ideal input/
output transfer function is shown in Figure 16.
Figure 15. AD7878 Analog Input
Figure 16. Input/Output Transfer Function
OFFSET AND FULL-SCALE ADJUSTMENT
In most Digital Signal Processing (DSP) applications offset and
full-scale error have little or no effect on system performance.
Offset error can always be eliminated in the analog domain by
ac coupling. Full-scale error effect is linear and does not cause
problems as long as the input signal is within the full dynamic
range of the ADC. Some applications may require that the input
signal span the full analog input dynamic range and, accord-
ingly, offset and full-scale error will have to be adjusted to zero.
Where adjustment is required, offset must be adjusted before
full-scale error. This is achieved by trimming the offset of the
op amp driving the analog input of the AD7878 while the input
voltage is 1/2 LSB below ground. The trim procedure is as
follows: apply a voltage of –0.73 mV (–1/2 LSB) at V
1
and
adjust the op amp offset voltage until the ADC output code
flickers between 1111 1111 1111 and 0000 0000 0000.
Gain error can be adjusted at either the first code transition
(ADC negative full scale) or the last code transition (ADC
positive full scale). The trim procedures for both cases are as
follows:
AD7878
–11–
REV. A
Positive Full-Scale Adjust
Apply a voltage of 2.9978 V (FS/2 – 3/2 LSBs) at V
1
. Adjust R2
until the ADC output code flickers between 0111 1111 1110
and 0111 1111 1111.
Negative Full-Scale Adjust
Apply a voltage of –2.9993 V (–FS/2 + 1/2 LSB) at V
l
and ad-
just R2 until the ADC output code flickers between 1000 0000
0000 and 1000 0000 0001.
Figure 17. AD7878 Full-Scale Adjust Circuit
MICROPROCESSOR INTERFACING
The AD7878 high speed bus timing allows direct interfacing to
DSP processors. Due to the complexity of the AD7878 internal
logic, only synchronous interfacing is allowed. This means that
the ADC clock must be the same as, or a derivative of, the pro-
cessor clock. Suitable processor interfaces are shown in Figures
18 to 21.
AD7878–ADSP-2100/TMS32010/TMS32020
All three interfaces use an external timer for conversion control,
allowing the ADC to sample the analog input asynchronously to
the microprocessor. The AD7878 ALFL output interrupts the
processor when the FIFO preprogrammed word count is
reached. The processor then reads the conversion results from
the AD7878 internal FIFO memory.
Figure 18. AD7878–ADSP-2700 Interface
Figure 19. AD7878–TMS32020 Interface
The interfaces to the ADSP-2100 and the TMS32020 gate the
AD7878 CS and the BUSY to provide a signal which drives the
processor into a wait state if a read/write operation to the ADC
is attempted while the ADC track/hold amplifier is going from
the track to the hold mode. This avoids digital feedthrough to
the analog circuitry. The TMS32020 does not have separate
RD and WR outputs to drive the AD7878 DMWR and
DMRD inputs. These are generated from the processor STRB
and R/W outputs with the addition of some logic gates.
Figure 20. AD7878–TMS32020 Interface
AD7878–M CC8000
This interface also uses an external timer for conversion control
as described for the previous three interfaces. It is discussed
separately because it needs extra logic due to the nature of its
interrupts. The MC68000 has eight levels of external interrupt.
When interrupting this processor one of these levels (0 to 7)
has to be encoded onto the IPL2–IPL0 inputs. This is achieved
with a 74148 encoder in Figure 21, (interrupt Level 1 is taken
for example purposes only). The MC68000 places this inter-
rupt level on address bits A3 to A1 at the start of the interrupt
service routine. Additional logic is used to decode this interrupt
level on the address bus and the FC2–FC0 outputs to generate
a VPA signal for the MC68000. This results in an autovectored
interrupt, the start address for the service routine must be
loaded into the appropriate auto vector location during initial-
ization. For further information on the 68000 interrupts con-
sult the 68000 User’s Manual.
AD7878
–12– REV. A
The MC68000 AS and R/W outputs are used to generate sepa-
rate DMWR and DMRD inputs for the AD7878. As with the
three interfaces previously described, WAIT states are inserted if
a read/write operation is attempted while the track/hold amplifier
is going from the track to the hold mode.
Figure 21. AD7878–MC68000 Interface
Typical AD7878 Microprocessor Operating Sequence
After power-up or reset, the status/control register is initialized
by writing to the AD7878. This enables the ALFL output if
required for a microprocessor interrupt and sets the effective word
length of the FIFO memory. The processor now executes the
main body of the program while waiting for an ADC interrupt.
This interrupt will occur when the preprogrammed number of
samples are collected in the FIFO memory. The interrupt ser-
vice routine first interrogates DB5(FOOR) of the status/control
register to determine if any sample in the FIFO memory is out
of range. If all data samples are valid, then the program pro-
ceeds to read the FIFO memory. If, on the other hand, at least
one sample is out of range, then an overrange routine is called.
There are many actions that can be taken by the out of range
routine, the selection of which is application dependent. One
option is to ignore all the current samples residing in the FIFO
memory, reinitialize the status/control register and return to the
main body of the program. Another option is to check the indi-
vidual out of range status of each word in the FIFO memory
and to discard the invalid ones. The underrange or overrange
status of each word can also be determined and the analog input
adjusted accordingly before returning to the main program.
Note: there is no need to check the out-of-range status if the
analog input is always assured to be within range.
THROUGHPUT RATE
The AD7878 has a maximum specified throughput rate (sample
rate) of 100 kHz. This is a worst-case test condition and specifi-
cations apply for reduced sampling rates, provided that Nyquist
criterion is obeyed. The throughput rate must take into account
ADC CONVST pulse width, ADC conversion time and the
track/hold amplifier acquisition time. The time required for each
of these tasks is shown in Table II for a selection of DSP proces-
sors. Since the ADC clock has to be synchronized to the micro-
processor dock, the conversion time depends on the micro-
processor used. In addition, time must be allowed for reading data
from the AD7878. If this task is performed during the track/
hold amplifier acquisition period, then it does not impact the
overall throughput rate. However, if the read operations occur
during a conversion, they may stretch the conversion time and
reduce the track/hold amplifier acquisition time. The track/hold
amplifier requires a minimum of 2 µs to operate to specification.
The time required to read from the AD7878 depends on the
number of FIFO memory locations to be read and the software
organization.
As an example, consider an application using the ADSP-2100
and the AD7878 with a throughput rate of 100 kHz. The time
required for the CONVST pulse and the ADC conversion is
7.375 µs. This leaves 2.625 µs for the track/hold acquisition
time and for reading the ADC (both operations occurring in
parallel). The ADSP-2100, when operating from a 32 MHz
clock, has an instruction cycle of 125 ns and an interrupt re-
sponse time of 500 ns. This allows adequate time to perform
16 read operations within the time budget allowed.
Table II. AD7878 Throughput Rate
CONVST Conversion T/H Acquisition
Pulse Width Time Time
Number of Non-
Clock Cycles 2 min 57 max Applicable
ADSP-2100
1
250 ns min 7.125 µs max 2 µs min
TMS32010
2
400 ns min 11.14 µs max 2 µs min
TMS32020
2
400 ns min 11.14 µs max 2 µs min
NOTES
1
ADSP-2100 Clock Frequency = 32 MHz.
2
TMS320XX Clock Frequency = 20 MHz.
APPLICATION HINTS
Good printed circuit board (PCB) layout is as important as the
overall circuit design itself in achieving high speed A/D perfor-
mance. The AD7878 is required to make bit decisions on an
LSB size of 1.465 mV. To achieve this, the designer has to be
conscious of noise both in the ADC itself and in the preceding
analog circuitry. Switching mode power supplies are not recom-
mended as the switching spikes will feed through to the com-
parator, causing noisy code transitions. Other concerns are
ground loops and digital feedthrough from microprocessors.
These factors influence any ADC, and a proper PCB layout that
minimizes these effects is essential for best performance.
LAYOUT HINTS
Ensure that the layout for the printed circuit board has the
digital and analog signal lines separated as much as possible.
Take care not to run any digital track alongside an analog signal
track. Guard (screen) the analog input with AGND.
AD7878
–13–
REV. A
Establish a single point analog ground (star ground) separate
from the logic system ground at Pin 22 (AGND) or as dose as
possible to the AD7878, as shown in Figure 22. Connect all
other grounds and Pin 7 (AD7878 DGND) to this single analog
ground point. Do not connect any other digital grounds to this
analog ground point. Low impedance analog and digital power
supply common returns are essential to low noise operation of
the ADC, so make the foil width for these tracks as wide as
possible. The use of ground planes minimizes impedance paths
and also guards the analog circuitry from digital noise. The
circuit layouts of Figures 25 and 26 have both analog and digital
ground planes, which are kept separated and only joined to-
gether at the AD7878 AGND pin.
NOISE
Keep the input signal leads to V
IN
and signal return leads from
AGND (Pin 22) as short as possible to minimize input noise
coupling. In applications where this is not possible, use a
shielded cable between the source and the ADC. Reduce the
ground circuit impedance as much as possible since any poten-
tial difference in grounds between the signal source and the
ADC appears as an error voltage in series with the input signal.
Figure 22. Power Supply Grounding Practice
DATA ACQUISITION BOARD
Figure 23 shows the AD7878 in a data acquisition circuit that
will interface directly to either the ADSP-2100, TMS32010 or
the TMS32020. The corresponding printed circuit board (PCB)
layout and silkscreen are shown in Figures 24 to 26.
The only additional component required for a full data acquisi-
tion system is an antialiasing filter. There is a component grid
provided near the analog input on the PCB which may be used
for such a filter or any other conditioning circuitry. To facilitate
this option, a wire link (labelled LK1 on the PCB) is required
on the analog input track. This link connects the input signal to
either the component grid or directly to the buffer amplifier
driving the AD7878 analog input.
Microprocessor connections to the PCB can be made by either
of two ways:
1. 96-contact (3 ROW) Eurocard connector.
2. 26-contact (2 ROW) IDC connector.
The 96-contact Eurocard connector is directly compatible with
the ADSP-2100 Evaluation Board Prototype Expansion Con-
nector. The expansion connector on the ADSP-2100 has eight
decoded drip enable outputs labelled ECE8 to ECE1. ECE6 is
used to drive the AD7878 CS input on the data acquisition
board. To avoid selecting onboard RAM sockets at the same
time, LK6 on the ADSP-2100 board must be removed. In addi-
tion, the expansion connector on the ADSP-2100 has four inter-
rupts labelled EIRQ3 to EIRQ0. The AD7878 ALFL output
connects to EIRQ0. The AD7878 and ADSP-2100 data lines
are aligned for left justified data transfer.
The 26-way IDC connector contains all the necessary contacts
for both the TMS32010 and TMS32020. There are two
switches on the data acquisition board that must be set to en-
able the appropriate interface configuration (see Table III). The
interface connections for the TMS32010/32020 and IDC signal
contact numbers are shown in Table IV and Figure 23. Note the
AD7878 CS input must be decoded from the address bus prior
to the AD7878 evaluation board for the TMS320XX interfaces.
Connections to the analog input (V
IN
) and the CONVST input
are made via two BNC sockets labelled SKT1 and SKT2 on the
silkscreen. If the CONVST input is derived from either the
microprocessor or ADC clock, the effects of clock noise cou-
pling will be reduced.
Table III. AD7878 PCB Switch Settings
SWITCH SETTING
Microprocessor SW1 SW2
ADSP-2100 A A
TMS32010 B A
TMS32020 B B
POWER SUPPLY CONNECTIONS
The PCB requires two analog supplies and one 5 V digital sup-
ply. Connections to the analog supplies are made directly to the
PCB as shown on the silk screen in Figure 24. The connections
are labelled V+ and V– and the range for both of these supplies
is 12 V to 15 V. Connection to the 5 V digital supply is made
through either of the two microprocessor connectors. The +5 V
and –5 V analog power supplies required by the AD7878 are
generated from two voltage regulators on the V+ and V– power
supply inputs (IC3 and IC4 in Figure 23).
COMPONENT LIST
IC1 AD711 Op Amp
IC2 AD7878 Analog-to-Digital
Converter
IC3 MC78L05 5 V Regulator
IC4 MC79L05 –5 V Regulator
IC5* 74HC00 Quad NAND Gate
IC6* 74HC04 Hex Inverter
IC7 74HC02 Quad NOR Gate
SW1 Single Pole Double Throw
SW2 Double Pole Double Throw
LK1 Wire Link for Analog Input
C1, C3, C5, C7, C9 10 µF Capacitors
C11, C13, C15
C2, C4, C6, C8, C10 0.1 µF Capacitors
C12, C14, C16
R1*, R2* 10 kΩ Resistors
SKT1, SKT2 BNC Sockets
SKT3 26-Contact (2 Row) IDC
Connector
SKT4 96-Contact (3 Row) Eurocard
Connector
*Not required for ADSP-2100 Interface.
AD7878
–14– REV. A
Figure 23. Data Acquisition Circuit Using the AD7878
Figure 24. PCB Silkscreen for Figure 23
AD7878
–15–
REV. A
Figure 25. PCB Component Side Layout for Figure 23
Figure 26. PCB Solder Side Layout for Figure 23
AD7878
–16– REV. A
C1204a–1–5/97
PRINTED IN U.S.A.
Table IV. TMS32010/TMS32020 Interface Connections
IDC Signal Connect TMS32010 TMS32020
Contact No. Mnemonic Signal Signal
1R/W—R/W
2STRB —STRB
3DMRD DEN —
4DMWR WE —
5CS CS CS
6 READY — READY
7RESET RESET RESET
8ALFL INT INT
9 ADD0 PA0 A0
10 CLK CLKOUT CLKOUT2
11 DB10 D10 D10
12 DB11 D11 D11
13 DB8 D8 D8
14 DB9 D9 D9
15 DB6 D6 D6
16 DB7 D7 D7
17 DB4 D4 D4
18 DB5 D5 D5
19 DB2 D2 D2
20 DB3 D3 D3
21 DB0 D0 D0
22 DB1 D1 D1
23 5 V 5 V 5 V
24 5 V 5 V 5 V
25 GND GND GND
26 GND GND GND
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
28-Pin Plastic DIP (N-28)
28-Pin Cerdip (Q-28)
28-Pin Ceramic DIP (D-28)
28-Terminal PLCC (P-28A)
28-Terminal LCCC (E-28A)
NOTE
1
Analog Devices reserves the right to ship either cerdip or ceramic hermetic
packages.
