LTC3864
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For more information www.linear.com/LTC3864
Typical applicaTion
FeaTures DescripTion
60V Low IQ Step-Down
DC/DC Controller with
100% Duty Cycle Capability
The LTC
®
3864 is a robust, high voltage step-down DC/DC
controller optimized for automotive and industrial applica-
tions. It drives a P-channel power MOSFET switch allowing
100% duty cycle operation. The wide input and output volt-
age ranges cover a multitude of applications. This device
has been verified with the failure mode and effects analysis
(FMEA) procedure for operation during failure conditions.
The LTC3864 offers excellent light load efficiency, draw-
ing only 40µA quiescent current in a user programmable
Burst Mode operation. Its peak current mode, constant
frequency PWM architecture provides for good control of
switching frequency and output current limit. The switch-
ing frequency can be programmed from 50kHz to 850kHz
with an external resistor and can be synchronized to an
external clock from 75kHz to 750kHz.
The LTC3864 offers programmable soft-start or output
tracking. Safety features include overvoltage protection,
overcurrent and short-circuit protection including fre-
quency foldback and a power good output signal.
The LTC3864 is available in thermally enhanced 12-Pin
MSOP and 3mm × 4mm DFN packages.
5.2V to 60V Input, 5V/2A Output, 350kHz Step-Down Converter
applicaTions
n Wide Operating VIN Range: 3.5V to 60V
n Wide VOUT Range: 0.8V to VIN
n Low Operating IQ = 40µA
n Very Low Dropout Operation: 100% Duty Cycle
n Strong High Voltage MOSFET Gate Driver
n Constant Frequency Current Mode Architecture
n Verified FMEA for Adjacent Pin Open/Short
n Selectable High Efficiency Burst Mode
®
Operation or
Pulse-Skipping Mode at Light Loads
n Programmable Fixed Frequency: 50kHz to 850kHz
n Phase-Lockable Frequency: 75kHz to 750kHz
n Accurate Current Limit
n Programmable Soft-Start or Voltage Tracking
n Internal Soft-Start Guarantees Smooth Start-Up
n Power Good Output Voltage Monitor
n Low Shutdown IQ = 7µA
n Available in Small 12-Pin Thermally Enhanced MSOP
and DFN Packages
n Industrial and Automotive Power Supplies
n Telecom Power Supplies
n Distributed Power Systems L, LT, LTC, LTM, OPTI-LOOP, Linear Technology, Burst Mode and the Linear logo are registered
trademarks and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks
are the property of their respective owners. Protected by U.S. Patents including 5731694.
Efficiency
350kHz
100k
25mΩ
10µH
CAP
0.47µF 10µF
PGND
LTC3864
3864 TA01a
SS
ITH
FREQ
SGND
RUN VIN
PLLIN/MODE
SENSE
GATE
PGOOD
VFB
9.09k
422k 47µF
×2
VOUT*
5V
2A
80.6k
VIN*
5.2V TO 60V
3.3nF
*VOUT FOLLOWS VIN
WHEN 3.5V ≤ VIN ≤ 5.2V
LOAD CURRENT (A)
0.01
50
EFFICIENCY (%)
80
70
60
90
100
0.1 1
3864 TA01b
PULSE-SKIPPING
Burst Mode
OPERATION
VIN = 12V
VOUT = 5V
LTC3864
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For more information www.linear.com/LTC3864
pin conFiguraTion
absoluTe MaxiMuM raTings
Input Supply Voltage (VIN) ......................... –0.3V to 65V
VIN-VSENSE Voltage ...................................... –0.3V to 6V
VIN-VCAP Voltage ........................................ –0.3V to 10V
RUN Voltage............................................... –0.3V to 65V
PGOOD, PLLIN/MODE Voltages ................... –0.3V to 6V
SS, ITH, FREQ, VFB Voltages ........................ –0.3V to 5V
(Note 1)
12
11
10
9
8
7
13
PGND
1
2
3
4
5
6
GATE
VIN
SENSE
CAP
RUN
PGOOD
PLLIN/MODE
FREQ
SGND
SS
VFB
ITH
TOP VIEW
DE PACKAGE
12-LEAD (4mm × 3mm) PLASTIC DFN
TJMAX = 150°C, θJA = 43°C/W, θJC = 5.5°C/W
EXPOSED PAD (PIN 13) IS PGND, MUST BE SOLDERED TO PCB FOR OPTIMAL
THERMAL PERFORMANCE
1
2
3
4
5
6
PLLIN/MODE
FREQ
SGND
SS
VFB
ITH
12
11
10
9
8
7
GATE
VIN
SENSE
CAP
RUN
PGOOD
TOP VIEW
13
PGND
MSE PACKAGE
12-LEAD PLASTIC MSOP
TJMAX = 150°C, θJA = 40°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 13) IS PGND, MUST BE SOLDERED TO PCB FOR OPTIMAL
THERMAL PERFORMANCE
orDer inForMaTion
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LTC3864EMSE#PBF LTC3864EMSE#TRPBF 3864 12-Lead Plastic MSOP –40°C to 125°C
LTC3864IMSE#PBF LTC3864IMSE#TRPBF 3864 12-Lead Plastic MSOP –40°C to 125°C
LTC3864HMSE#PBF LTC3864HMSE#TRPBF 3864 12-Lead Plastic MSOP –40°C to 150°C
LTC3864MPMSE#PBF LTC3864MPMSE#TRPBF 3864 12-Lead Plastic MSOP –55°C to 150°C
LTC3864EDE#PBF LTC3864EDE#TRPBF 3864 12-Lead (4mm × 3mm) Plastic DFN –40°C to 125°C
LTC3864IDE#PBF LTC3864IDE#TRPBF 3864 12-Lead (4mm × 3mm) Plastic DFN –40°C to 125°C
LTC3864HDE#PBF LTC3864HDE#TRPBF 3864 12-Lead (4mm × 3mm) Plastic DFN –40°C to 150°C
LTC3864MPDE#PBF LTC3864MPDE#TRPBF 3864 12-Lead (4mm × 3mm) Plastic DFN –55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
Operating Junction Temperature Range (Notes 2, 3, 4)
LTC3864E,I ....................................... –40°C to 125°C
LTC3864H ..........................................–40°C to 150°C
LTC3864MP ....................................... –55°C to 150°C
Storage Temperature Range .................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
MSOP Package .................................................300°C
LTC3864
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For more information www.linear.com/LTC3864
elecTrical characTerisTics
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Input Supply
VIN Input Voltage Operating Range 3.5 60 V
VUVLO Undervoltage Lockout (VIN-VCAP) Ramping Up Threshold
(VIN-VCAP) Ramping Down Threshold
Hysteresis
l
l
3.25
3.00 3.50
3.25
0.25
3.8
3.50 V
V
V
IQInput DC Supply Current
Pulse-Skipping Mode PLLIN/MODE = 0V, FREQ = 0V,
VFB = 0.83V (No Load) 0.77 1.2 mA
Burst Mode Operation PLLIN/MODE = Open, FREQ = 0V,
VFB = 0.83V (No Load) 40 60 µA
Shutdown Supply Current RUN = 0V 7 12 µA
Output Sensing
VREG Regulated Feedback Voltage VITH = 1.2V (Note 5) l0.792 0.800 0.809 V
∆VREG
∆VIN
Feedback Voltage Line Regulation VIN = 3.8V to 60V (Note 5) –0.005 0.005 %/V
∆VREG
∆VITH
Feedback Voltage Load Regulation VITH = 0.6V to 1.8V (Note 5) –0.1 –0.015 0.1 %
gm(EA) Error Amplifier Transconductance VITH = 1.2V, ∆IITH = ±5µA (Note 5) 1.8 mS
IFB Feedback Input Bias Current –50 –10 50 nA
Current Sensing
VILIM Current Limit Threshold (VIN-VSENSE) VFB = 0.77V l85 95 103 mV
ISENSE SENSE Pin Input Current VSENSE = VIN 0.1 2 µA
Start-Up and Shutdown
VRUN RUN Pin Enable Threshold VRUN Rising l1.22 1.26 1.32 V
VRUNHYS RUN Pin Hysteresis 150 mV
ISS Soft-Start Pin Charging Current VSS = 0V 10 µA
Switching Frequency and Clock Synchronization
f Programmable Switching Frequency RFREQ = 24.9kΩ
RFREQ = 64.9kΩ
RFREQ = 105kΩ
375 105
440
810
505 kHz
kHz
kHz
fLO Low Switching Frequency FREQ = 0V 320 350 380 kHz
fHI High Switching Frequency FREQ = Open 485 535 585 kHz
fSYNC Synchronization Frequency l75 750 kHz
VCLK(IH) Clock Input High Level into PLLIN/MODE l2 V
VCLK(LO) Clock Input Low Level into PLLIN/MODE l0.5 V
fFOLD Foldback Frequency as Percentage of
Programmable Frequency VFB = 0V, FREQ = 0V 18 %
tON(MIN) Minimum On-Time 220 ns
Gate Driver
VCAP Gate Bias LDO Output Voltage (VIN-VCAP) IGATE = 0mA l7.6 8.0 8.5 V
VCAPDROP Gate Bias LDO Dropout Voltage VIN = 5V, IGATE = 15mA 0.2 0.5 V
∆VCAP(LINE) Gate Bias LDO Line Regulation 9V ≤ VIN ≤ 60V, IGATE = 0mA 0.002 0.03 %/V
∆VCAP(LOAD) Gate Bias LDO Load Regulation Load = 0mA to 20mA –3.5 %
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C. (Notes 3, 4) VIN = 12V, unless otherwise noted.
LTC3864
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For more information www.linear.com/LTC3864
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
RUP Gate Pull-Up Resistance Gate High 2 Ω
RDN Gate Pull-Down Resistance Gate Low 0.9 Ω
PGOOD and Overvoltage
VPGL PGOOD Voltage Low IPGOOD = 2mA 0.2 0.4 V
IPG PGOOD Leakage Current VPGOOD = 5V 1 µA
%PGD PGOOD Trip Level VFB Ramping Negative with Respect to VREG
Hysteresis –13 –10
2.5 –7 %
%
VFB Ramping Positive with Respect to VREG
Hysteresis 7 10
2.5 13 %
%
tPGDLY PGOOD Delay PGOOD Going High to Low
PGOOD Going Low to High 100
100 µs
µs
VFBOV VFB Overvoltage Lockout Threshold GATE Going High without Delay,
VFB(OV)-VFB(NOM) in Percent 10 %
elecTrical characTerisTics
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C. (Notes 3, 4) VIN = 12V, unless otherwise noted.
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: Continuous operation above the specified maximum operating
junction temperature may impair device reliability or permanently damage
the device.
Note 3: The junction temperature (TJ in °C) is calculated from the ambient
temperature (TA in °C) and power dissipation (PD in Watts) as follows:
TJ = TA + (PD • θJA)
where θJA (in °C/W) is the package thermal impedance provided in the Pin
Configuration section for the corresponding package.
Note 4: The LTC3864 is tested under pulsed loading conditions such that
TJ ≈ TA. The LTC3864E is guaranteed to meet performance specifications
from 0°C to 85°C operating junction temperature range. The LTC3864E
specifications over the –40°C to 125°C operating junction temperature
range are assured by design, characterization and correlation with statistical
process controls. The LTC3864I is guaranteed to meet performance
specifications over the –40°C to 125°C operating junction temperature
range, the LTC3864H is guaranteed over the –40°C to 150°C operating
junction temperature range, and the LTC3864MP is guaranteed and tested
over the full –55°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes; operating lifetime is
derated for junction temperatures greater than 125°C. The maximum
ambient temperature consistent with these specifications is determined by
specific operating conditions in conjunction with board layout, the rated
package thermal impedance and other environmental factors.
Note 5: The LTC3864 is tested in a feedback loop that adjust VFB to achieve
a specified error amplifier output voltage (on ITH pin).
LTC3864
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For more information www.linear.com/LTC3864
Typical perForMance characTerisTics
Transient Response:
Burst Mode Operation
Dropout Behavior (100% Duty
Cycle) Low VIN Operation
Normal Soft Start-Up
Soft Start-Up into a Prebiased
Output Output Tracking
Pulse-Skipping Mode Operation
Waveforms
Burst Mode Operation
Waveforms
Transient Response:
Pulse-Skipping Mode Operation
TA = 25°C, unless otherwise noted.
VIN = 12V, VOUT = 5V
FIGURE 8 CIRCUIT
VOUT
1V/DIV
SS
200mV/DIV
VIN
5V/DIV
1ms/DIV 3864 G07
VIN = 12V, VOUT = 5V
ILOAD = 0.5mA
FIGURE 8 CIRCUIT
SS
200mV/DIV
VOUT
1V/DIV
RUN
5V/DIV
1ms/DIV 3864 G08
VOUT PREBIASED
TO 2.9V
VIN = 12V, VOUT = 5V
ILOAD = 100mA
FIGURE 8 CIRCUIT
VOUT
2V/DIV
SS
200mV/DIV
20ms/DIV 3864 G09
VIN TRANSIENT: 12V TO 4V
AND BACK TO 12V
VOUT = 5V, ILOAD = 100mA, FIGURE 8 CIRCUIT
GATE
10V/DIV
VOUT
2V/DIV
VIN
2V/DIV
50ms/DIV 3864 G05
VOUT = VIN IN DROPOUT
VIN = 0V TO 3.8V
THEN BACK TO 0V
ILOAD = 100mA
FIGURE 8 CIRCUIT
SW
5V/DIV
VOUT
1V/DIV
VIN
1V/DIV
20ms/DIV 3864 G06
VOUT PROGRAMMED TO 5V,
BUT STARTS UP IN DROPOUT
SINCE VIN < 5V
VIN = 12V
VOUT = 5V
ILOAD = 100mA
FIGURE 8 CIRCUIT
IL
500mA/DIV
VSW
10V/DIV
VOUT
50mV/DIV
10µs/DIV 3864 G02
VIN = 12V
VOUT = 5V
ILOAD = 100mA
FIGURE 8 CIRCUIT
IL
500mA/DIV
VSW
10V/DIV
VOUT
50mV/DIV
2µs/DIV
3864 G01
VIN = 12V
VOUT = 5V
TRANSIENT = 100mA TO 2A
FIGURE 8 CIRCUIT
IL
2A/DIV
VOUT
500mV/DIV
ILOAD
2A/DIV
100µs/DIV 3864 G03
VIN = 12V
VOUT = 5V
TRANSIENT = 100mA TO 2A
FIGURE 8 CIRCUIT
IL
2A/DIV
VOUT
500mV/DIV
ILOAD
2A/DIV
100µs/DIV 3864 G04
LTC3864
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For more information www.linear.com/LTC3864
Typical perForMance characTerisTics
Burst Mode Input Current Over
Input Voltage (No Load)
Pulse-Skipping Mode Input
Current Over Input Voltage
(No Load)
Shutdown Current Over Input
Voltage
Output Regulation Over Input
Voltage
Output Regulation Over Load
Current
Output Regulation Over
Temperature
Overcurrent Protection Short-Ciruit Protection VIN Line Transient Behavior
TA = 25°C, unless otherwise noted.
VIN = 12V, VOUT = 5V
FIGURE 8 CIRCUIT
VOUT
500mV/DIV
IL
1A/DIV
ILOAD
1A/DIV
1A 1A
20ms/DIV 3864 G10
3.2A
VOUT DROOPS DUE TO
REACHING CURRENT LIMIT
VIN = 12V, VOUT = 5V
FIGURE 8 CIRCUIT
IL
2A/DIV
VOUT
5V/DIV
SHORT-
CIRCUIT
TRIGGER
500µs/DIV 3864 G11
SHORT-CIRCUIT REGION
SOFT RECOVERY
FROM SHORT
VIN(V)
0
30
IVIN (µA)
35
55
50
45
40
60
65
70
10 20 30 40
3864 G13
50 60
VIN = 12V, VOUT = 5V
ILOAD = 0A
FIGURE 8 CIRCUIT
VIN (V)
0
700
IVIN (µA)
850
800
750
900
950
10 20 30 40
3864 G14
50 60
VIN = 12V, VOUT = 5V
ILOAD = 0A
FIGURE 8 CIRCUIT
VIN (V)
0
0
IVIN (µA)
15
10
5
20
25
10 20 30 40
3864 G15
50 60
FIGURE 8 CIRCUIT
VIN = 12V, SURGE TO 48V
VOUT = 5V
ILOAD = 200mA, FIGURE 8 CIRCUIT
VOUT
50mV/DIV
GATE
20V/DIV
VIN
20V/DIV
2ms/DIV 3864 G12
VIN (V)
0
–0.010
NORMALIZED ∆VOUT (%)
0
–0.005
0.005
0.010
10 20 30 40
3864 G16
50 60
VOUT = 5V
ILOAD = 200mA
VOUT NORMALIZED AT VIN = 12V
FIGURE 8 CIRCUIT
Burst Mode OPERATION
PULSE-SKIPPING
ILOAD (A)
–0.5
–0.010
NORMALIZED ∆VOUT (%)
0
–0.005
0.005
0.010
00.5 1 1.5
3864 G17
2 2.5
VIN = 12V, VOUT = 5V
ILOAD NORMALIZED AT ILOAD = 1A
FIGURE 8 CIRCUIT
Burst Mode OPERATION
PULSE-SKIPPING
TEMPERATURE (°C)
–75
–1.0
NORMALIZED ∆VOUT (%)
0
1.0
0.8
0.6
0.4
0.2
–0.2
–0.4
–0.6
–0.8
–25 25 75 125
3864 G18
175
VIN = 12V, VOUT = 5V
ILOAD = 200mA
VOUT NORMALIZED TO TA = 25°C
FIGURE 8 CIRCUIT
Burst Mode OPERATION
PULSE-SKIPPING
LTC3864
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For more information www.linear.com/LTC3864
Typical perForMance characTerisTics
GATE Bias LDO (VIN - VCAP) Load
Regulation
GATE Bias LDO (VIN - VCAP)
Dropout Behavior
Current Sense Voltage Over ITH
Voltage
Current Sense Voltage Over
Temperature
SS Pin Pull-Up Current Over
Temperature
RUN Pin Pull-Up Current Over
Temperature
Free Running Frequency Over
Input Voltage
Free Running Frequency Over
Temperature
Frequency Foldback % Over
Feedback Voltage
TA = 25°C, unless otherwise noted.
VIN (V)
0
300
f (kHz)
450
600
550
500
400
350
10 20 30 40 50
3864 G19
60
FREQ = 0V
FREQ = OPEN
VFB (mV)
0
0
FREQUENCY FOLDBACK %
60
120
100
80
40
20
200 400 600
3864 G21
800
IGATE (mA)
0
–3.5
(VIN - VCAP) REGULATION (%)
–2.0
0.5
–1.0
–0.5
0.0
–1.5
–2.5
–3.0
510 15
3864 G22
20
IGATE (mA)
0
–0.5
(VIN - VCAP) DROPOUT (V)
0.1 VIN = 5V
–0.1
0.0
–0.2
–0.3
–0.4
510 15
3864 G23
20
TEMPERATURE (°C)
–75
90
CURRENT LIMIT SENSE VOLTAGE (mV)
100
98
94
92
96
–25 25 75 125
3864 G25
175
TEMPERATURE (°C)
–75
0.25
RUN PULL-UP CURRENT (µA)
0.65
0.55
0.35
0.45
–25 25 75 125
3864 G27
175
TEMPERATURE (°C)
–75
300
f (kHz)
450
600
550
500
400
350
–25 25 75 125
3864 G20
175
FREQ = 0V
FREQ = OPEN
TEMPERATURE (°C)
–75
6
SS PULL-UP CURRENT (µV)
14
12
8
10
–25 25 75 125
3864 G26
175
ITH VOLTAGE (V)
0
–10
CURRENT SENSE VOLTAGE (mV)
100
80
90
40
30
20
10
0
70
60
50
0.4 0.8 1.2 1.6
3864 G24
2
Burst Mode OPERATION
PULSE-SKIPPING
LTC3864
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For more information www.linear.com/LTC3864
pin FuncTions
PLLIN/MODE (Pin 1): External Reference Clock Input
and Burst Mode Enable/Disable. When an external clock
is applied to this pin, the internal phase-locked loop will
synchronize the turn-on edge of the gate drive signal with
the rising edge of the external clock. When no external
clock is applied, this input determines the operation during
light loading. Floating this pin selects low IQ (40μA) Burst
Mode operation. Pulling to ground selects pulse-skipping
mode operation.
FREQ (Pin 2): Switching Frequency Set Point Input. The
switching frequency is programmed by an external set-
point resistor RFREQ connected between the FREQ pin and
signal ground. An internal 20µA current source creates
a voltage across the external setpoint resistor to set the
internal oscillator frequency. Alternatively, this pin can
be driven directly by a DC voltage to set the oscillator
frequency. Grounding selects a fixed operating frequency
of 350kHz. Floating selects a fixed operating frequency
of 535kHz.
SGND (Pin 3): Ground Reference for Small Signal Analog
Component (Signal Ground). Signal ground should be used
as the common ground for all small signal analog inputs
and compensation components. Connect signal ground to
power ground (ground reference for power components)
only at one point using a single PCB trace.
SS (Pin 4): Soft-Start and External Tracking Input. The
LTC3864 regulates the feedback voltage to the smaller of
0.8V or the voltage on the SS pin. An internal 10μA pull-up
current source is connected to this pin. A capacitor to
ground at this pin sets the ramp time to the final regulated
output voltage. Alternatively, another voltage supply con-
nected through a resistor divider to this pin allows the
output to track the other supply during start-up.
VFB (Pin 5): Output Feedback Sense. A resistor divider
from the regulated output point to this pin sets the output
voltage. The LTC3864 will nominally regulate VFB to the
internal reference value of 0.8V. If VFB is less than 0.4V, the
switching frequency will linearly decrease and fold back
to about one-fifth of the internal oscillator frequency to
reduce the minimum duty cycle.
ITH (Pin 6): Current Control Threshold and Controller
Compensation Point. This pin is the output of the error
amplifier and the switching regulator’s compensation
point. The voltage ranges from 0V to 2.9V, with 0.8V cor-
responding to zero sense voltage (zero current).
PGOOD (Pin 7): Power Good Indicator Output. This open
drain logic output is pulled to ground when the output
voltage is outside of a ±10% window around the regulation
point. The PGOOD switches states only after a 100µs delay.
RUN (Pin 8): Digital Run Control Input. A RUN voltage
above the 1.26V threshold enables normal operation, while
a voltage below the threshold shuts down the controller.
An internal 0.4µA current source pulls the RUN pin up to
about 3.3V. The RUN pin can be connected to an external
power supply up to 60V.
CAP (Pin 9): Gate Driver (–) Supply. A low ESR ceramic
bypass capacitor of at least 0.47µF or 10X the effective
CMILLER of the P-channel power MOSFET, is required from
VIN to this pin to serve as a bypass capacitor for the in-
ternal regulator. To insure stable low noise operation, the
bypass capacitor should be placed adjacent to the VIN and
CAP pins and connected using the same PCB metal layer.
SENSE (Pin 10): Current Sense Input. A sense resistor
RSENSE from VIN pin to the SENSE pin sets the maximum
current limit. The peak inductor current limit is equal to
95mV/RSENSE. For accuracy, it is important that the VIN
pin and the SENSE pin route directly to the current sense
resistor and make a Kelvin (4-wire) connection.
VIN (Pin 11): Chip Power Supply. A minimum bypass
capacitor of 0.1µF is required from the VIN pin to power
ground. For best performance use a low ESR ceramic
capacitor placed near the VIN pin.
GATE (Pin 12): Gate Drive Output for External P-Channel
MOSFET. The gate driver bias supply voltage (VIN-VCAP)
is regulated to 8V when VIN is greater than 8V. The gate
driver is disabled when (VIN-VCAP) is less than 3.5V (typi-
cal), 3.8V maximum in startup and 3.25V (typical) 3.5V
maximum in normal operation.
PGND (Exposed Pad Pin 13): Ground Reference for Power
Components (Power Ground). The PGND exposed pad must
be soldered to the circuit board for electrical contact and
for rated thermal performance of the package. Connect
signal ground to power ground only at one point using a
single PCB trace.
LTC3864
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For more information www.linear.com/LTC3864
–+
EA
(Gm = 1.8mS)
0.8V
10µA
LOGIC
CONTROL
LDO
IN
OUT
PLL
SYSTEM
Q
S R
MODE/CLOCK
DETECT
DELAY
100µs
VCO
OV O.88V
SLOPE
COMPENSATION
O.72V
UV
3.25V
GATE
CAP
SS
VFB
VIN – 8V
SENSE
VIN
1.26V
–
+
+
RPGD
PLLIN/MODE
PGND
CCAP
MP
D1
VOUT
UVLO
RFREQ
SGND
FREQ
RUN RUN
0.4µA
20µA
3864 FD
+
–
–
+
–
DRV
CLOCK
PGOOD
+
–
+
–
O.425V
Burst Mode
OPERATION
+
–
ITH
RITH
CITH1
L
CSS
CIN
VIN
RSENSE
COUT
VOUT
RFB2
RFB1
ICMP
+
FuncTional DiagraM
LTC3864
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For more information www.linear.com/LTC3864
operaTion
Main Control Loop (Refer to Functional Diagram)
The LTC3864 uses a peak current-mode control architec-
ture to regulate the output in an asynchronous step-down
DC/DC switching regulator. The VFB input is compared to
an internal reference by a transconductance error ampli-
fier (EA). The internal reference can be either a fixed 0.8V
reference VREF or the voltage input on the SS pin. In normal
operation VFB regulates to the internal 0.8V reference
voltage. In soft-start or tracking mode, when the SS pin
voltage is less than the internal 0.8V reference voltage,
VFB will regulate to the SS pin voltage. The error amplifier
output connects to the ITH (current [I] threshold [TH])
pin. The voltage level on the ITH pin is then summed with
a slope compensation ramp to create the peak inductor
current set point.
The peak inductor current is measured through a sense
resistor RSENSE placed across the VIN and SENSE pins.
The resultant differential voltage from VIN to SENSE is
proportional to the inductor current and is compared to the
peak inductor current set point. During normal operation
the P-channel power MOSFET is turned on when the clock
leading edge sets the SR latch through the S input. The
P-channel MOSFET is turned off through the SR latch R
input when the differential voltage from VIN to SENSE is
greater than the peak inductor current set point and the
current comparator, ICMP, trips high.
Power CAP and VIN Undervoltage Lockout (UVLO)
Power for the P-channel MOSFET gate driver is derived
from the CAP pin. The CAP pin is regulated to 8V below
VIN in order to provide efficient P-channel operation. The
power for the VCAP supply comes from an internal LDO,
which regulates the VIN-CAP differential voltage. A mini-
mum capacitance of 0.47µF (low ESR ceramic) is required
between VIN and CAP to assure stability.
For VIN ≤ 8V, the LDO will be in dropout and the CAP volt-
age will be at ground, i.e. the VIN-CAP differential voltage
will equal VIN. If VIN-CAP is less than 3.25V (typical), the
LTC3864 enters a UVLO state where the GATE is prevented
from switching and most internal circuitry is shut down.
In order to exit UVLO, the VIN-CAP voltage would have to
exceed 3.5V (typical).
Shutdown and Soft-Start
When the RUN pin is below 0.7V, the controller and most
internal circuits are disabled. In this micropower shutdown
state, the LTC3864 draws only 7µA. Releasing the RUN
pin allows a small internal pull up current to pull the RUN
pin above 1.26V and enable the controller. The RUN pin
can be pulled up to an external supply of up to 60V or it
can be driven directly by logic levels.
The start-up of the output voltage VOUT is controlled by
the voltage on the SS pin. When the voltage on the SS
pin is less than the 0.8V internal reference, the VFB pin is
regulated to the voltage on the SS pin. This allows the SS
pin to be used to program a soft-start by connecting an
external capacitor from the SS pin to signal ground. An
internal 10µA pull-up current charges this capacitor, creat-
ing a voltage ramp on the SS pin. As the SS voltage rises
from 0V to 0.8V
, the output voltage VOUT rises smoothly
from zero to its final value.
Alternatively, the SS pin can be used to cause the start-
up of VOUT to track that of another supply. Typically, this
requires connecting the SS pin to an external resistor
divider from the other supply to ground. (See Applications
Information section.) Under shutdown or UVLO, the SS
pin is pulled to ground and prevented from ramping up.
If the slew rate of the SS pin is greater than 1.2V/ms, the
output will track an internal soft-start ramp instead of the
SS pin. The internal soft-start will guarantee a smooth
start-up of the output under all conditions, including in the
case of a short-circuit recovery where the output voltage
will recover from near ground.
Light Load Current Operation (Burst Mode Operation
or Pulse-Skipping Mode)
The LTC3864 can be enabled to enter high efficiency Burst
Mode operation or pulse-skipping mode at light loads. To
select pulse-skipping operation, tie the PLLIN/MODE pin
to signal ground. To select Burst Mode operation, float
the PLLIN/MODE pin.
In Burst Mode operation, if the VFB is higher than the refer-
ence voltage, the error amplifier will decrease the voltage
on the ITH pin. When the ITH voltage drops below 0.425V,
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the internal sleep signal goes high, enabling sleep mode.
The ITH pin is then disconnected from the output of the
error amplifier and held at 0.55V.
In sleep mode, much of the internal circuitry is turned
off, reducing the quiescent current to 40µA while the load
current is supplied by the output capacitor. As the output
voltage and hence the feedback voltage decreases, the
error amplifier’s output will rise. When the output voltage
drops enough, the ITH pin is reconnected to the output
of the error amplifier, the sleep signal goes low, and the
controller resumes normal operation by turning on the
external P-MOSFET on the next cycle of the internal oscil-
lator. In Burst Mode operation, the peak inductor current
has to reach at least 25% of current limit for the current
comparator, ICMP, to trip and turn the P-MOSFET back off,
even though the ITH voltage may indicate a lower current
setpoint value.
When the PLLIN/MODE pin is connected for pulse-skipping
mode, the LTC3864 will skip pulses during light loads. In
this mode, ICMP may remain tripped for several cycles and
force the external MOSFET to stay off, thereby skipping
pulses. This mode offers the benefits of smaller output
ripple, lower audible noise, and reduced RF interference,
at the expense of lower efficiency when compared to Burst
Mode operation.
Frequency Selection and Clock Synchronization
The switching frequency of the LTC3864 can be selected
using the FREQ pin. If the PLLIN/MODE pin is not being
driven by an external clock source, the FREQ pin can be
tied to signal ground, floated, or programmed through an
external resistor. Tying FREQ pin to signal ground selects
350kHz, while floating selects 535kHz. Placing a resistor
between FREQ pin and signal ground allows the frequency
to be programmed between 50kHz and 850kHz.
The phase-locked loop (PLL) on the LTC3864 will syn-
chronize the internal oscillator to an external clock source
when connected to the PLLIN/MODE pin. The PLL forces
the turn-on edge of the external P-channel MOSFET to be
aligned with the rising edge of the synchronizing signal.
The oscillator’s default frequency is based on the operating
frequency set by the FREQ pin. If the oscillator’s default
frequency is near the external clock frequency, only slight
adjustments are needed for the PLL to synchronize the
external P-channel MOSFET’s turn-on edge to the rising
edge of the external clock. This allows the PLL to lock
rapidly without deviating far from the desired frequency.
The PLL is guaranteed from 75kHz to 750kHz. The clock
input levels should be greater than 2V for HI and less
than 0.5V for LO.
Power Good and Fault Protection
The PGOOD pin is an open-drain output. An internal
N-channel MOSFET pulls the PGOOD pin low when the VFB
pin voltage is outside a ±10% window from the 0.8V internal
voltage reference. The PGOOD pin is also pulled low when
the RUN pin is low (shut down). When the VFB pin voltage
is within the ±10% window, the MOSFET is turned off and
the pin is allowed to be pulled up by an external resistor
to a source no greater than 6V. The PGOOD open-drain
output has a 100µs delay before it can transition states.
When the VFB voltage is above +10% of the regulated
voltage of 0.8V, this is considered as an overvoltage con-
dition and the external P-MOSFET is immediately turned
off and prevented from ever turning on until VFB returns
below +7.5%.
In the event of an output short circuit or overcurrent con-
dition that causes the output voltage to drop significantly
while in current limit, the LTC3864 operating frequency
will fold back. Anytime the output feedback VFB voltage is
less than 50% of the 0.8V internal reference (i.e., 0.4V),
frequency foldback is active. The frequency will continue
to drop as VFB drops until reaching a minimum foldback
frequency of about 18% of the setpoint frequency. Fre-
quency foldback is designed, in combination with peak
current limit, to limit current in start-up and short-circuit
conditions. Setting the foldback frequency as a percentage
of operating frequency assures that start-up characteristics
scale appropriately with operating frequency.
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The LTC3864 is a current mode, constant frequency PWM
controller for an asynchronous step-down DC/DC regulator
with a P-channel power MOSFET acting as the main switch
and a Schottky power diode acting as the commutating
(catch) diode. The input range extends from 3.5V to 60V.
The output range can be programmed from 0.8V to all the
way up to VIN. The LTC3864 can transition from regulation
to 100% duty cycle when the input voltage drops below
the programmed output voltage. Additionally, the LTC3864
offers Burst Mode operation with 40µA quiescent current,
which delivers outstanding efficiency in light load opera-
tion. The LTC3864 is a low pin count, robust and easy to
use solution in applications which require high efficiency
and operate with widely varying high voltage inputs.
The typical application on the front page is a basic LTC3864
application circuit. The LTC3864 can sense the inductor
current through a high side series sense resistor, RSENSE,
placed between VIN and the source of the external P-
MOSFET. Once the required output voltage and operating
frequency have been determined, external component
selection is driven by load requirements, and begins with
the selection of inductor and RSENSE. Next, the power
MOSFET and catch diode are selected. Finally, input and
output capacitors are selected.
Output Voltage Programming
The output voltage is programmed by connecting a
feedback resistor divider from the output to the VFB pin
as shown in Figure 1. The output voltage in steady state
operation is set by the feedback resistors according to
the equation:
VOUT =0.8V •1+RFB2
R
FB1
To improve the transient response, a feedforward capacitor
CFF may be used. Great care should be taken to route the
VFB line away from noise sources, such as the inductor
or the GATE signal that drives the external P-MOSFET.
applicaTions inForMaTion
Switching Frequency and Clock Synchronization
The choice of operating frequency is a trade-off between
efficiency and component size. Lowering the operating fre-
quency improves efficiency by reducing MOSFET switching
losses but requires larger inductance and/or capacitance
to maintain low output ripple voltage. Conversely, raising
the operating frequency degrades efficiency but reduces
component size.
The LTC3864 can free run at a user programmed switch-
ing frequency, or it can synchronize with an external
clock to run at the clock frequency. When the LTC3864
is synchronized, the GATE pin will phase synchronize
with the rising edge of the applied clock in order to turn
the external P-MOSFET on. The switching frequency of
the LTC3864 is programmed with the FREQ pin, and the
external clock is applied at the PLLIN/MODE pin. Table 1
highlights the different states in which the FREQ pin can
be used in conjunction with the PLLIN/MODE pin.
Table 1
FREQ PIN PLLIN/MODE PIN FREQUENCY
0V DC Voltage 350kHz
Floating DC Voltage 535kHz
Resistor to GND DC Voltage 50kHz to 850kHz
Any of the Above External Clock Phase Locked to External
Clock
LTC3864
VFB
VOUT
RFB2 CFF
RFB1
3864 F01
Figure 1. Setting the Output Voltage
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The free-running switching frequency can be programmed
from 50kHz to 850kHz by connecting a resistor from FREQ
pin to signal ground. The resulting switching frequency as
a function of resistance on FREQ pin is shown in Figure 2.
Set the free-running frequency to the desired synchroni-
zation frequency using the FREQ pin so that the internal
oscillator is prebiased to approximately the synchronization
frequency. While it is not required that the free-running
frequency be near the external clock frequency, doing so
will minimize synchronization time.
Inductor Selection
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use of
smaller inductor and capacitor values. A higher frequency
generally results in lower efficiency because of MOSFET
gate charge and transition losses. In addition to this basic
trade-off, the effect of inductor value on ripple current and
low current operation must also be considered.
Given the desired input and output voltages, the inductor
value and operation frequency determine the ripple current:
∆IL=VOUT
f•L
1– VOUT
V
IN
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors and results in lower
output ripple. Highest efficiency operation is obtained at
low frequency with small ripple current. However, achieving
FREQ PIN RESISTOR (kΩ)
15
FREQUENCY (kHz)
600
800
1000
35 45 5525
3864 F02
400
200
500
700
900
300
100
065 75 85 95 105 115 125
Figure 2. Switching Frequency vs Resistor on FREQ Pin
this requires a large inductor. There is a trade-off between
component size, efficiency, and operating frequency.
A reasonable starting point for ripple current is 40% of
IOUT(MAX) at nominal VIN. The largest ripple current occurs
at the highest VIN. To guarantee that the ripple current does
not exceed a specified maximum, the inductance should
be chosen according to:
L=VOUT
f•∆IL(MAX)
1– VOUT
VIN(MAX)
Once the inductance value has been determined, the type
of inductor must be selected. Core loss is independent of
core size for a given inductor value, but it is very depen-
dent on the inductance selected. As inductance increases,
core losses decrease. Unfortunately, increased inductance
requires more turns of wire and therefore copper losses
will increase.
High efficiency converters generally cannot tolerate the
core loss of low cost powdered iron cores, forcing the use
of more expensive ferrite materials. Ferrite designs have
very low core loss and are preferred at high switching
frequencies, so design goals can concentrate on cop-
per loss and preventing saturation. Ferrite core material
saturates hard, which means that inductance collapses
abruptly when the peak design current is exceeded. This
will result in an abrupt increase in inductor ripple current
and output voltage ripple. Do not allow the core to saturate!
A variety of inductors are available from manufacturers
such as Sumida, Panasonic, Coiltronics, Coilcraft, Toko,
Vishay, Pulse, and Würth.
Current Sensing and Current Limit Programming
The LTC3864 senses the inductor current through a cur-
rent sense resistor, RSENSE, placed across the VIN and
SENSE pins. The voltage across the resistor, VSENSE, is
proportional to inductor current and in normal operation
is compared to the peak inductor current setpoint. A
current limit condition is detected when VSENSE exceeds
95mV. When the current limit threshold is exceeded, the
P-channel MOSFET is immediately turned off by pulling
the GATE voltage to VIN regardless of the controller input.
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The peak inductor current limit is equal to:
IL(PEAK) ≅95mV
R
SENSE
This inductor current limit would translate to an output
current limit based on the inductor ripple:
ILIMIT ≅
95mV
R
SENSE
–
∆I
L
2
The SENSE pin is a high impedance input with a maximum
leakage of ±2µA. Since the LTC3864 is a peak current
mode controller, noise on the SENSE pin can create pulse
width jitter. Careful attention must be paid to the layout of
RSENSE. To ensure the integrity of the current sense signal,
VSENSE, the traces from VIN and SENSE pins should be
short and run together as a differential pair and Kelvin
(4-wire) connected across RSENSE (Figure 3).
drain current ID(MAX), and the MOSFET’s thermal resistance
θJC(MOSFET) and θJA(MOSFET).
The gate driver bias voltage VIN-VCAP is set by an internal
LDO regulator. In normal operation, the CAP pin will be
regulated to 8V below VIN. A minimum 0.1µF capacitor
is required across the VIN and CAP pins to ensure LDO
stability. If required, additional capacitance can be added
to accommodate higher gate currents without voltage
droop. In shutdown and Burst Mode operation, the CAP
LDO is turned off. In the event of CAP leakage to ground,
the CAP voltage is limited to 9V by a weak internal clamp
from VIN to CAP. As a result, a minimum 10V VGS rated
MOSFET is required.
The power dissipated by the P-channel MOSFET when the
LTC3864 is in continuous conduction mode is given by:
PMOSFET ≅D•IOUT
2
•ρt•RDS(ON) +
VIN2•IOUT
2
•CMILLER
( )
•
RDN
V
IN
– V
CAP
( )
– V
MILLER
+RUP
V
MILLER
•f
where D is duty factor, RDS(ON) is on-resistance of
P-MOSFET, ρt is temperature coefficient of on-resistance,
RDN is the pull-down driver resistance specified at 0.9Ω
typical and RUP is the pull-up driver resistance specified at
2Ω typical. VMILLER is the Miller effective VGS voltage and
is taken graphically from the power MOSFET data sheet.
The power MOSFET input capacitance CMILLER is
the most important selection criteria for determin-
ing the transition loss term in the P-channel MOSFET
but is not directly specified on MOSFET data sheets.
CMILLER is a combination of several components, but
it can be derived from the typical gate charge curve
included on most data sheets (Figure 4). The curve is
The LTC3864 has internal filtering of the current sense
voltage which should be adequate in most applications.
However
, adding a provision for an external filter offers
added flexibility and noise immunity, should it be neces-
sary. The filter can be created by placing a resistor from the
RSENSE resistor to the SENSE pin and a capacitor across
the VIN and SENSE pins.
Power MOSFET Selection
The LTC3864 drives a P-channel power MOSFET that
serves as the main switch for the asynchronous step-
down converter. Important P-channel power MOSFET
parameters include drain-to-source breakdown voltage
VBR(DSS), threshold voltage VGS(TH), on-resistance RDS(ON),
gate-to-drain reverse transfer capacitance CRSS, maximum
Figure 3. Inductor Current Sensing
Figure 4. (a) Typical P-MOSFET Gate Charge Characteristics
and (b) Test Set-Up to Generate Gate Charge Curve
S
D
G
VSD(TEST)
RLOAD
IGATE
3864 F04
MILLER EFFECT
QIN
a b
CMILLER = (QB – QA)/VSD(TEST)
VSG
–
+
(a) (b)
VIN
RSENSE
LTC3864
VIN
SENSE RF
MP
OPTIONAL
FILTERING
3864 F03
CF
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generated by forcing a constant current out of the gate of a
common-source connected P-MOSFET that is loaded with
a resistor, and then plotting the gate voltage versus time.
The initial slope is the effect of the gate-to-source and
gate-to-drain capacitances. The flat portion of the curve
is the result of the Miller multiplication effect of the drain-
to-gate capacitance as the drain voltage rises across the
resistor load. The Miller charge (the increase in coulombs
on the horizontal axis from a to b while the curve is flat) is
specified for a given VSD test voltage, but can be adjusted
for different VSD voltages by multiplying by the ratio of
the adjusted VSD to the curve specified VSD value. A way
to estimate the CMILLER term is to take the change in gate
charge from points a and b (or the parameter QGD on a
manufacturer’s data sheet) and dividing it by the specified
VSD test voltage, VSD(TEST).
CMILLER ≅
Q
GD
VSD(TEST)
The term with CMILLER accounts for transition loss, which
is highest at high input voltages. For VIN < 20V, the high-
current efficiency generally improves with larger MOSFETs,
while for VIN > 20V, the transition losses rapidly increase
to the point that the use of a higher RDS(ON) device with
lower CMILLER actually provides higher efficiency.
Schottky Diode Selection
When the P-MOSFET is turned off, a power Schottky diode
is required to function as a commutating diode to carry the
inductor current. The average diode current is therefore
dependent on the P-MOSFET’s duty factor. The worst case
condition for diode conduction is a short-circuit condition
where the Schottky must handle the maximum current
as its duty factor approaches 100% (and the P-channel
MOSFET’s duty factor approaches 0%). The diode there-
fore must be chosen carefully to meet worst case voltage
and current requirements. The equation below describes
the continuous or average forward diode current rating
required, where D is the regulator duty factor.
I
F(AVG)
≅I
OUT(MAX)
•1–D
( )
Once the average forward diode current is calculated,
the power dissipation can be determined. Refer to the
Schottky diode data sheet for the power dissipation
PDIODE as a function of average forward current IF(AVG).
PDIODE can also be iteratively determined by the two
equations below, where VF(IOUT, TJ) is a function of both
IF(AVG) and junction temperature TJ. Note that the thermal
resistance θJA(DIODE) given in the data sheet is typical and
can be highly layout dependent. It is therefore important
to make sure that the Schottky diode has adequate heat
sinking.
T
J
≅P
DIODE
•θ
JA(DIODE)
PDIODE ≅IF(AVG)•VF(IOUT,TJ)
The Schottky diode forward voltage is a function of both
IF(AVG) and TJ, so several iterations may be required to
satisfy both equations. The Schottky forward voltage VF
should be taken from the Schottky diode data sheet curve
showing Instantaneous Forward Voltage. The forward
voltage will increase as a function of both TJ and IF(AVG).
The nominal forward voltage will also tend to increase as
the reverse breakdown voltage increases. It is therefore
advantageous to select a Schottky diode appropriate to
the input voltage requirements.
CIN and COUT Selection
The input capacitance CIN is required to filter the square
wave current through the P-channel MOSFET. Use a low
ESR capacitor sized to handle the maximum RMS current.
ICIN(RMS) ≅IOUT(MAX)•VOUT
VIN
•VIN
VOUT
–1
The formula has a maximum at VIN = 2VOUT, where
ICIN(RMS) = IOUT(MAX)/2. This simple worst-case condition
is commonly used for design because even significant
deviations do not offer much relief. Note that ripple cur-
rent ratings from capacitor manufacturers are often based
on only 2000 hours of life, which makes it advisable to
derate the capacitor.
The selection of COUT is primarily determined by the ESR
required to minimize voltage ripple and load step transients.
The ∆VOUT is approximately bounded by:
∆VOUT ≤∆ILESR+1
8•f•COUT
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Since ∆IL increases with input voltage, the output ripple
is highest at maximum input voltage. Typically, once the
ESR requirement is satisfied, the capacitance is adequate
for filtering and has the necessary RMS current rating.
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Dry tantalum, specialty polymer, aluminum electrolytic
and ceramic capacitors are all available in surface mount
packages. Specialty polymer capacitors offer very low
ESR but have lower specific capacitance than other types.
Tantalum capacitors have the highest specific capacitance,
but it is important to only use types that have been surge
tested for use in switching power supplies. Aluminum
electrolytic capacitors have significantly higher ESR, but
can be used in cost-sensitive applications provided that
consideration is given to ripple current ratings and long-
term reliability. Ceramic capacitors have excellent low ESR
characteristics but can have a high voltage coefficient and
audible piezoelectric effects.
The high Q of ceramic capacitors with trace inductance
can also lead to significant ringing. When used as input
capacitors, care must be taken to ensure that ringing from
inrush currents and switching does not pose an overvolt-
age hazard to the power switch and controller. To dampen
input voltage transients, add a small 5μF to 40μF aluminum
electrolytic capacitor with an ESR in the range of 0.5Ω to
2Ω. High performance through-hole capacitors may also
be used, but an additional ceramic capacitor in parallel
is recommended to reduce the effect of lead inductance.
Discontinuous and Continuous Operation
The LTC3864 operates in discontinuous conduction (DCM)
until the load current is high enough for the inductor
current to be positive at the end of the switching cycle.
The output load current at the continuous/discontinuous
boundary IOUT(CDB) is given by the following equation:
IOUT(CDB) ≅(VIN – VOUT)( VOUT
+
VF)
2•L•f•(V
IN
+V
F
)
The continuous/discontinuous boundary is inversely
proportional to the inductor value. Therefore, if required,
IOUT(CDB) can be reduced by increasing the inductor value.
External Soft-Start and Output Tracking
Start-up characteristics are controlled by the voltage on
the SS pin. When the voltage on the SS pin is less than
the internal 0.8V reference, the LTC3864 regulates the VFB
pin voltage to the voltage on the SS pin. When the SS pin
is greater than the internal 0.8V reference, the VFB pin
voltage regulates to the 0.8V internal reference. The SS
pin can be used to program an external soft-start function
or to allow VOUT to track another supply during start-up.
Soft-start is enabled by connecting a capacitor from
the SS pin to ground. An internal 10µA current source
charges the capacitor, providing a linear ramping voltage
at the SS pin that causes VOUT to rise smoothly from 0V
to its final regulated value. The total soft-start time will
be approximately:
tSS =CSS •0.8V
10µA
When the LTC3864 is configured to track another supply,
a voltage divider can be used from the tracking supply to
the SS pin to scale the ramp rate appropriately. Two com-
mon implementations of tracking as shown in Figure 5a
are coincident and ratiometric. For coincident tracking,
make the divider ratio from the external supply the same
as the divider ratio for the feedback voltage. Ratiometric
tracking could be achieved by using a different ratio than
the feedback (Figure 5b).
Note that the soft-start capacitor charging current is always
flowing, producing a small offset error
. To minimize this
error, select the tracking resistive divider values to be small
enough to make this offset error negligible.
Short-Circuit Faults: Current Limit and Foldback
The maximum inductor current is inherently limited in a
current mode controller by the maximum sense voltage.
In the LTC3864, the maximum sense voltage is 95mV,
measured across the inductor sense resistor RSENSE,
placed across the VIN and SENSE pins. The output current
limit is approximately:
ILIMIT ≅
95mV
R
SENSE
–
∆I
L
2
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The current limit must be chosen to ensure that ILIMIT(MIN)
> IOUT(MAX) under all operating conditions. The minimum
current limit value should be greater than the inductor
current required to produce maximum output power at
worst case efficiency. Worst-case efficiency typically oc-
curs at the highest VIN.
Short-circuit fault protection is assured by the combination
of current limit and frequency foldback. When the output
feedback voltage VFB drops below 0.4V, the operating
frequency f will fold back to a minimum value of 0.18 • f
when VFB reaches 0V. Both current limit and frequency
foldback are active in all modes of operation. In a short-
circuit fault condition, the output current is first limited by
current limit and then further reduced by folding back the
operating frequency as the short becomes more severe.
Short-Circuit Recovery and Internal Soft-Start
An internal soft-start feature guarantees a maximum posi-
tive output voltage slew rate in all operational cases. In a
short-circuit recovery condition for example, the output
recovery rate is limited by the internal soft-start so that
output voltage overshoot and excessive inductor current
buildup is prevented.
The internal soft-start voltage and the external SS pin
operate independently. The output will track the lower of
the two voltages. The slew rate of the internal soft-start
voltage is roughly 1.2V/ms, which translates to a total
soft-start time of 650µs. If the slew rate of the SS pin
is greater than 1.2V/ms the output will track the internal
soft-start ramp. To assure robust fault recovery, the
Figure 5(a). Two Different Modes of Output Tracking
TIME
Coincident Tracking
EXTERNAL
SUPPLY EXTERNAL
SUPPLY
VOUT
VOLTAGE
VOUT
TIME 3864 F05a
Ratiometric Tracking
VOLTAGE
Figure 5(b): Setup for Ratiometric and Coincident Tracking
RFB2
EXT. V
RFB1
Coincident Tracking Setup
TO SS
RFB2
VOUT
TO VFB
RFB1
R1
EXT. V
R2
R1+ R2 ≥
R2
TO SS
RFB2
VOUT
TO VFB
RFB1
3864 F05b
Ratiometric Tracking Setup
0.8V
EXT. V
LTC3864
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applicaTions inForMaTion
internal soft-start feature is active in all operational cases.
If a short-circuit condition occurs which causes the output
to drop significantly, the internal soft-start will assure a
soft recovery when the fault condition is removed.
The internal soft-start assures a clean soft ramp-up from
any fault condition that causes the output to droop, guar-
anteeing a maximum ramp rate in soft-start, short-circuit
fault release, or output recovery from drop out. Figure 6
illustrates how internal soft-start controls the output
ramp-up rate under varying scenarios.
guaranteed to operate down to a VIN of 3.5V over the full
temperature range.
The implications of both the UVLO rising and UVLO falling
specifications must be carefully considered for low VIN
operation. The UVLO threshold with VIN rising is typi-
cally 3.5V (with a maximum of 3.8V) and UVLO falling is
typically 3.25V (with a maximum of 3.5V). The operating
input voltage range of the LTC3864 is guaranteed to be
3.5V to 60V over temperature, but the initial VIN ramp
must exceed 3.8V to guarantee start-up.
For example, Figure 7 illustrates LTC3864 operation when
an automotive battery droops during a cold crank condi-
tion. The typical automotive battery is 12V to 14V, which is
more than enough headroom above 3.8V for the LTC3864
to start up. Onboard electronics which are powered by a
DC/DC regulator require a minimum supply voltage for
seamless operation during the cold crank condition, and
the battery may droop close to these minimum supply
requirements during a cold crank. The DC/DC regulator
should not exacerbate the situation by having excessive
dropout between the already suppressed battery voltage
input and the output of the regulator which power these
electronics. As seen in Figure 7, the LTC3864’s 100%
duty cycle capability allows virtually no dropout (only the
IOUT • (RSENSE + RDS(ON)) drop across the sense resistor
and P-MOSFET if there is a significant IOUT) from the battery
to the output. The 3.5V guaranteed UVLO point assures
sufficient margin for continuous, uninterrupted operation in
extreme cold crank battery drooping conditions. However,
additional input capacitance or slower soft start-up time
may be required at low VIN (e.g. 3.5V to 4.5V) in order to
limit VIN droop caused by inrush currents, especially if
the battery or input source has a sufficiently large input
impedance.
Figure 6. Internal Soft-Start (a) Allows Soft Start-Up without
an External Soft-Start Capacitor and Allows Soft Recovery from
(b) a Short-Circuit or (c) a VIN Dropout
Figure 7. Typical Automotive Cold Crank
VIN Undervoltage Lockout (UVLO)
The LTC3864 is designed to accommodate applications
requiring widely varying power input voltages from 3.5V
to 60V. To accommodate the cases where VIN drops
significantly once in regulation, the LTC3864 is
3864 F07
TIME
VOUT
VBATTERY
12V
LTC3864’s 100% DUTY CYCLE CAPABILITY ALLOWS
VOUT TO RIDE VIN WITHOUT SIGNIFICANT DROP-OUT
5V
VOLTAGE
TIME~ 650µs
(a)
VOUT
VIN
VOLTAGE
3864 F06
INTERNAL SOFT-START INDUCED START-UP
(NO EXTERNAL SOFT-START CAPACITOR)
TIME
SHORT-CIRCUIT
(b)
VOUT
VOLTAGE
INTERNAL SOFT-START
INDUCED RECOVERY
INTERNAL SOFT-START
INDUCED RECOVERY
TIME
(c)
VOUT
VIN
VIN
DROPOUT
VOLTAGE
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applicaTions inForMaTion
Minimum On-Time Considerations
The minimum on-time, tON(MIN), is the smallest time
duration that the LTC3864 is capable of turning on the
power MOSFET, and is typically 220ns. It is determined
by internal timing delays and the gate charge required to
turn on the MOSFET. Low-duty-cycle applications may
approach this minimum on-time limit, so care should be
taken to ensure that:
tON(MIN)<VOUT
VIN(MAX) •f
If the duty cycle falls below what can be accommodated
by the minimum on-time, the controller will skip cycles.
However, the output voltage will continue to regulate.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
the dominant contributors and therefore where efficiency
improvements can be made. Percent efficiency can be
expressed as:
% Efficiency = 100% - (L1+L2+L3+…)
where L1, L2, L3, etc., are the individual losses as a per-
centage of input power.
Although all dissipative elements in the circuit produce
losses, four main sources account for most of the losses
in LTC3864 application circuits.
1. I2R Loss: I2R losses result from the P-channel MOSFET
resistance, inductor resistance, the current sense resis-
tor, and input and output capacitor ESR. In continuous
mode operation the average output current flows through
L but is chopped between the P-channel MOSFET and
the bottom side Schottky diode. The following equation
may be used to determine the total I2R loss:
PI2R≈(I2OUT+∆I2L/12)•[RDCR+D•(RDS(ON)+RSENSE
+RESR(CIN))]+∆I2L/12•RESR(COUT)
2. Transition Loss: Transition loss of the P-channel MOS-
FET becomes significant only when operating at high
input voltages (typically 20V or greater.) The P-channel
transition losses (PPMOSTRL) can be determined from
the following equation:
PPMOSTRL =VIN2•IOUT
2
•(CMILLER)•
RDN
(V
IN
– V
CAP
) – V
MILLER
+RUP
V
MILLER
•f
3. Gate Charging Loss: Charging and discharging the gate
of the MOSFET will result in an effective gate charg-
ing current. Each time the P-channel MOSFET gate is
switched from low to high and low again, a packet of
charge dQ moves from the capacitor across VIN – VCAP
and is then replenished from ground by the internal VCAP
regulator. The resulting dQ/dt current is a current out
of VIN flowing to ground. The total power loss in the
controller including gate charging loss is determined
by the following equation:
P
CNTRL
=V
IN
•(I
Q
+f•Q
G(PMOSFET)
)
4. Schottky Loss: The Schottky diode loss is most signifi-
cant at low duty factors (high step down ratios). The
critical component is the Schottky forward voltage as
a function of junction temperature and current. The
Schottky power loss is given by the equation below.
P
DIODE
≅(1–D)•I
OUT
•V
F(IOUT,TJ)
When making adjustments to improve efficiency, the in-
put current is the best indicator of changes in efficiency.
If changes cause the input current to decrease, then the
efficiency has increased. If there is no change in input
current, there is no change in efficiency.
OPTI-LOOP
®
Compensation
OPTI-LOOP compensation, through the availability of the
ITH pin, allows the transient response to be optimized for
a wide range of loads and output capacitors. The ITH pin
not only allows optimization of the control loop behavior
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but also provides a test point for the step-down regulator ’s
DC-coupled and AC-filtered closed-loop response. The DC
step, rise time and settling at this test point truly reflects the
closed-loop response. Assuming a predominantly second
order system, phase margin and/or damping factor can be
estimated using the percentage of overshoot seen at this
pin. The bandwidth can also be estimated by examining
the rise time at this pin.
The ITH series RITH-CITH1 filter sets the dominant pole-zero
loop compensation. Additionally, a small capacitor placed
from the ITH pin to signal ground, CITH2, may be required to
attenuate high frequency noise. The values can be modified
to optimize transient response once the final PCB layout
is done and the particular output capacitor type and value
have been determined. The output capacitors need to be
selected because their various types and values determine
the loop feedback factor gain and phase. An output current
pulse of 20% to 100% of full load current having a rise
time of 1μs to 10μs will produce output voltage and ITH
pin waveforms that will give a sense of the overall loop
stability without breaking the feedback loop. The general
goal of OPTI-LOOP compensation is to realize a fast but
stable ITH response with minimal output droop due to
the load step. For a detailed explanation of OPTI-LOOP
compensation, refer to Application Note 76.
Switching regulators take several cycles to respond to a
step in load current. When a load step occurs, VOUT im-
mediately shifts by an amount equal to ∆ILOAD • ESR, where
ESR is the effective series resistance of COUT
. ∆ILOAD also
begins to charge or discharge COUT
, generating a feedback
error signal used by the regulator to return VOUT to its
steady-state value. During this recovery time, VOUT can
be monitored for overshoot or ringing that would indicate
a stability problem.
Connecting a resistive load in series with a power MOSFET,
then placing the two directly across the output capacitor
and driving the gate with an appropriate signal generator
is a practical way to produce a realistic load-step condi-
tion. The initial output voltage step resulting from the step
change in output current may not be within the bandwidth
of the feedback loop, so this signal cannot be used to
determine phase margin. This is why it is better to look
at the ITH pin signal which is in the feedback loop and
is the filtered and compensated feedback loop response.
The gain of the loop increases with RITH and the bandwidth
of the loop increases with decreasing CITH1. If RITH is
increased by the same factor that CITH1 is decreased, the
zero frequency will be kept the same, thereby keeping the
phase the same in the most critical frequency range of the
feedback loop. In addition, a feedforward capacitor, CFF
, can
be added to improve the high frequency response, as shown
in Figure 1. Capacitor CFF provides phase lead by creating
a high frequency zero with RFB2 which improves the phase
margin. The output voltage settling behavior is related to
the stability of the closed-loop system and will demonstrate
overall performance of the step-down regulator.
In some applications, a more severe transient can be caused
by switching in loads with large (>10μF) input capacitors.
If the switch connecting the load has low resistance and
is driven quickly, then the discharged input capacitors are
effectively put in parallel with COUT
, causing a rapid drop in
VOUT
. No regulator can deliver enough current to prevent
this problem. The solution is to limit the turn-on speed of
the load switch driver. A Hot Swap™ controller is designed
specifically for this purpose and usually incorporates cur-
rent limiting, short-circuit protection and soft starting.
Design Example
Consider a step-down converter with the following
specifications: VIN = 5V to 55V, VOUT = 5V, IOUT(MAX) = 2A,
and f = 350kHz (Figure 8).
The output voltage is programmed according to:
VOUT =0.8V •1+RFB2
R
FB1
If RFB1 is chosen to be 80.6k, then RFB2 would have to
be 422k.
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The FREQ pin is tied to signal ground in order to program
the switching frequency to 350kHz. The on-time required
at 55V to generate a 5V output can be calculated as:
tON =
V
OUT
V
IN
•f=
5V
55V •350kHz ≈260ns
This on-time is larger than LTC3864’s minimum on-time
with sufficient margin to prevent cycle skipping.
Next, set the inductor value to give 60% worst-case ripple
at maximum VIN = 55V.
L=5V
350kHz •(0.6 •2A)
1– 5V
55V
≈10.8µH
Select 10µH, which is a standard value.
The resulting maximum ripple current is:
∆IL=5V
350kHz •10µH
1– 5V
55V
≈1.3A
Figure 8. Design Example (5V, 2A 350kHz Step-Down Converter)
Next, set the RSENSE resistor value to ensure that the
converter can deliver a maximum output current of 2.0A
with sufficient margin to account for component varia-
tions and worst-case operating conditions. Using a 30%
margin factor:
RSENSE ≅
95mV
1.3•2A +1.3A
2
≈27.5mΩ
Use a more readily available 25mΩ sense resistor.
The current limit is:
ILIMIT ≅
95mV
25mΩ
–
1.3A
2
≈3.15A
Next choose a P-channel MOSFET with the appropri-
ate BVDSS and ID rating. In this example, a good choice
is the Fairchild FDMC5614P (BVDSS = 60V, ID = 5.7A,
RDS(ON) = 105mΩ, ρ100°C = 1.5, CMILLER = 100pF,
θJA = 60°C/W). The expected power dissipation and the
Efficiency
CSS
0.1µF
RRUN
100k
RPGD
100k
MP
D1
SW
L1
10µH
CAP
CCAP
0.47µF CIN2
4.7µF
CIN1
12µF
63V
PGND
LTC3864
3864 F08a
SS
ITH
FREQ
SGND
RUN VIN
MODE/PLLN
SENSE
GATE
PGOOD
VFB
RSENSE
25mΩ
RITH 9.53k
RFB2
422k
47µF
×2
VOUT*
5V
2A
RFB1
80.6k
VIN*
5.2V TO 55V
CVIN
0.1µF
CITH1
3.3nF
CITH2 100pF
+
CIN1: NICHICON UPJ1J120MDD
D1: DIODES INC SBR3U100LP
L1: TOKO 1217AS-H-100M
MP: FAIRCHILD FDMC5614P *VOUT FOLLOWS VIN WHEN 3.5V ≤ VIN ≤ 5.2V
SEE DROPOUT BEHAVIOR IN TYPICAL PERFORMANCE CHARACTERISTICS
CFF
47pF
LOAD CURRENT (A)
0.01
50
EFFICIENCY (%)
80
70
60
90
100
0.1 1
3864 F08b
PULSE-SKIPPING
Burst Mode
OPERATION
VIN = 12V
VOUT = 5V
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resulting junction temperature for the MOSFET can be
calculated at TA = 70°C, VIN(MAX) = 55V and IOUT(MAX) = 2A:
PPMOS =
5V
55V (2A)2•1.5•105mΩ+
(55V)2•(2A / 2)•100pF •0.9Ω
8V – 3V
+2Ω
3V •350kHz
≈57mW +90mW =147mW
T
J
=70°C+147mW •60°C/W ≈80°C
The calculations can be repeated for VIN(MIN) = 5V:
PPMOS =
5V
5V (2A)2•1.5•105mΩ+
(5.2V)2•100pF •0.9Ω
5.2V – 3V +2Ω
3V
•350kHz
≈630mW +1mW ≈631mW
T
J
=70°C+631mW •60°C / W ≈108°C
Next choose an appropriate Schottky diode that will handle
the power requirements. The Diodes Inc. SBR3U100LP
Schottky diode is selected (VF(2A,125°C) = 0.5V, θJA = 61°C/W)
for this application. The power dissipation and junction
temperature at TA = 70°C can be calculated as:
PDIODE =2A •1– 5V
55V
•0.5V ≈909mW
TJ=70°C+909mW •61°C/W =125°C
These power dissipation calculations show that careful
attention to heat sinking will be necessary.
For the input capacitance, a combination of ceramic and
electrolytic capacitors are chosen to handle the maximum
RMS current of 1A. COUT will be selected based on the
ESR that is required to satisfy the output voltage ripple
requirement. For this design, two 47µF ceramic capacitors
are chosen to offer low ripple in both normal operation
and in Burst Mode operation.
A soft-start time of 8ms can be programmed through a
0.1µF capacitor on the SS pin:
CSS =
8ms•10µA
0.8V
=0.1µF
Loop compensation components on the ITH pin are chosen
based on load step transient behavior (as described under
OPTI-LOOP Compensation) and is optimized for stability. A
pull-up resistor is used on the RUN pin for FMEA compli-
ance (see Failure Modes and Effects Analysis).
Gate Driver Component Placement,
Layout and Routing
It is important to follow recommended power supply PC
board layout practices such as placing external power ele-
ments to minimize loop area and inductance in switching
paths. Be careful to pay particular attention to gate driver
component placement, layout and routing.
The effective CCAP capacitance should be greater than 0.1µF
minimum in all operating conditions. Operating voltage
and temperature both decrease the rated capacitance to
varying degrees depending on dielectric type. The LTC3864
is a PMOS controller with an internal gate driver and boot-
strapped LDO that regulates the differential CAP voltage
(VIN – VCAP) to 8V nominal. The CCAP capacitance needs
to be large enough to assure stability and provide cycle-
to-cycle current to the PMOS switch with minimum series
inductance. We recommend a ceramic 0.47µF 16V capacitor
with a high quality dielectric such as X5R or X7R. Some
high current applications with large Qg PMOS switches
may benefit from an even larger CCAP capacitance.
Figure 9 shows the LTC3864 Generic Application Sche-
matic which includes an optional current sense filter and
series gate resistor. Figure 10 illustrates the recommended
gate driver component placement, layout and routing of
the GATE, VIN, SENSE and CAP pins and key gate driver
components. It is recommended that the gate driver layout
follow the example shown in Figure 10 to assure proper
operation and long term reliability.
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The LTC3864 gate driver should connect to the external
power elements in the following manner. First route the
VIN pin using a single low impedance isolated trace to
the positive RSENSE resistor PAD without connection to
the VIN plane. The reason for this precaution is that the
VIN pin is internally Kelvin connected to the current sense
comparator, internal VIN power and the PMOS gate driver.
Connecting the VIN pin to the VIN power plane adds noise
and can result in jitter or instability. Figure 10 shows a
single VIN trace from the positive RSENSE pad connected
to CSF, CCAP, VIN pad and CINB. The total trace length to
RSENSE should be minimized and the capacitors CSF, CCAP
and CINB should be placed near the VIN pin of the LTC3864.
CCAP should be placed near the VIN and CAP pins. Figure 10
shows CCAP placed adjacent to the VIN and CAP pins with
SENSE routed between the pads. This is the recommended
layout and results in the minimum parasitic inductance.
The gate driver is capable of providing high peak current.
Parasitic inductance in the gate drive and the series in-
ductance between VIN to CAP can cause a voltage spike
between VIN and CAP on each switching cycle. The voltage
spike can result in electrical over-stress to the gate driver
and can result in gate driver failures in extreme cases. It
is recommended to follow the example shown in Figure 10
for the placement of CCAP as close as is practical.
RGATE resistor pads can be added with a 0Ω resistor to
allow the damping resistor to be added later. The total
length of the gate drive trace to the PMOS gate should
be minimized and ideally be less than 1cm. In most cases
with a good layout the RGATE resistor is not needed. The
RGATE resistor should be located near the gate pin to re-
duce peak current through GATE and minimize reflected
noise on the gate pin.
The RSF and CSF pads can be added with a zero ohm resis-
tor for RSF and CSF not populated. In most applications,
external filtering is not needed. The current sense filter
RSF and CSF can be added later if noise if demonstrated
to be a problem.
The bypass capacitor CINB is used to locally filter the
VIN supply. CINB should be tied to the VIN pin trace and
to the PGND exposed pad. The CINB positive pad should
connect to RSENSE positive though the VIN pin trace. The
CINB ground trace should connect to the PGND exposed
pad connection.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3864.
1. Multilayer boards with dedicated ground layers are
preferable for reduced noise and for heat sinking pur-
applicaTions inForMaTion
CSF
L1
Q1 D1
CAP
CCAP
PGND
LTC3864
3864 F09
SS
ITH
FREQ
SGND
GROUND
PLANE
TO PGND
RUN VIN
PLLIN/MODE
SENSE
GATE
PGOOD
VFB
RITH
RSF RSENSE
RGATE
RFREQ RPGD
RFB2
VOUT
VIN
CIN
+
–
CITH
CPITH
CINB
CSS
COUT
RFB1
RGATE
TO Q1 GATE
TO RSENSE+
3864 F10
CINB
CCAP
GATE
SENSE
CAP
VIN
CSF
RSF
TO RSENSE–
Figure 9. LTC3864 Generic Application Schematic with Optional
Current Sense Filter and Series Gate Resistor
Figure 10. LTC3864 Recommended Gate Driver PC
Board Placement, Layout and Routing
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poses. Use wide rails and/or entire planes for VIN, VOUT
and GND for good filtering and minimal copper loss. If
a ground layer is used, then it should be immediately
below (and/or above) the routing layer for the power
train components which consist of CIN, sense resistor,
P-MOSFET, Schottky diode, inductor, and COUT. Flood
unused areas of all layers with copper for better heat
sinking.
2. Keep signal and power grounds separate except at the
point where they are shorted together. Short signal and
power ground together only at a single point with a
narrow PCB trace (or single via in a multilayer board).
All power train components should be referenced to
power ground and all small signal components (e.g.,
CITH1, RFREQ, CSS etc.) should be referenced to signal
ground.
3. Place CIN, sense resistor, P-MOSFET, inductor, and
primary COUT capacitors close together in one compact
area. The junction connecting the drain of P-MOSFET,
cathode of Schottky, and (+) terminal of inductor (this
junction is commonly referred to as switch or phase
node) should be compact but be large enough to handle
the inductor currents without large copper losses. Place
the sense resistor and source of P-channel MOSFET
as close as possible to the (+) plate of CIN capacitor(s)
that provides the bulk of the AC current (these are
normally the ceramic capacitors), and connect the
anode of the Schottky diode as close as possible to
the (–) terminal of the same CIN capacitor(s). The high
dI/dt loop formed by CIN, the MOSFET, and the Schottky
diode should have short leads and PCB trace lengths to
minimize high frequency EMI and voltage stress from
inductive ringing. The (–) terminal of the primary COUT
capacitor(s) which filter the bulk of the inductor ripple
current (these are normally the ceramic capacitors)
should also be connected close to the (–) terminal of CIN.
4. Place pins 7 to 12 facing the power train components.
Keep high dV/dt signals on GATE and switch away from
sensitive small signal traces and components.
5. Place the sense resistor close to the (+) terminal of CIN
and source of P-MOSFET. Use a Kelvin (4-wire) con-
nection across the sense resistor and route the traces
together as a differential pair into the VIN and SENSE
pins. An optional RC filter could be placed near the VIN
and SENSE pins to filter the current sense signal.
6. Place the resistive feedback divider RFB1/2 as close as
possible to the VFB pin. The (+) terminal of the feedback
divider should connect to the output regulation point
and the (–) terminal of feedback divider should connect
to signal ground.
7. Place the ceramic CCAP capacitor as close as possible
to VIN and CAP pins. This capacitor provides the gate
discharging current for the power P-MOSFET.
8. Place small signal components as close to their respec-
tive pins as possible. This minimizes the possibility of
PCB noise coupling into these pins. Give priority to
VFB, ITH, and FREQ pins. Use sufficient isolation when
routing a clock signal into PLLIN /MODE pin so that the
clock does not couple into sensitive small signal pins.
Failure Mode and Effects Analysis (FMEA)
A FMEA study on the LTC3864 has been conducted through
adjacent pin opens and shorts. The device was tested
in a step-down application (Figure 8) from VIN = 12V to
VOUT = 5V with a current load of 1A on the output. One
group of tests involved the application being monitored
while each pin was disconnected from the PC board
and left open while all other pins remained intact. The
other group of tests involved each pin being shorted to
its adjacent pins while all other pins were connected as
it would be normally in the application. The results are
shown in Table 2.
For FMEA compliance, the following design implementa-
tions are recommended:
• If the RUN pin is being pull-up to a voltage greater than
6V, then it is done so through a pull-up resistor (100k
to 1M) so that the PGOOD pin is not damaged in case
of a RUN to PGOOD short.
• The gate of the external P-MOSFET be pulled through
a resistor (20k to 100k) to the input supply, VIN so that
the P-MOSFET is guaranteed to turn off in case of a
GATE open.
applicaTions inForMaTion
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applicaTions inForMaTion
Table 2
FAILURE MODE VOUT IOUT IVIN f
RECOVERY
WHEN
FAULT IS
REMOVED? BEHAVIOR
None 5V 1A 453mA 350kHz N/A Normal Operation.
Pin Open
Open Pin 1 (PLLIN/MODE) 5V 1A 453mA 350kHz OK Pin already left open in normal application, so no difference.
Open Pin 2 (FREQ) 5V 1A 453mA 535kHz OK Frequency jumps to default open value.
Open Pin 3 (GND) 5V 1A 453mA 350kHz OK Exposed pad still provides GND connection to device.
Open Pin 4 (SS) 5V 1A 453mA 350kHz OK External soft-start removed, but internal soft-start still available.
Open Pin 5 (VFB) 0V 0A 0.7mA 0kHz OK Controller stops switching. VFB internally self biases HI to prevent
switching.
Open Pin 6 (ITH) 5V 1A 507mA 40kHz OK Output still regulating, but the switching is erratic. Loop not stable.
Open Pin 7 (PGOOD) 5V 1A 453mA 350kHz OK No PGOOD output, but controller regulates normally.
Open Pin 8 (RUN) 5V 1A 453mA 350kHz OK Controller does not start-up.
Open Pin 9 (CAP) 5V 1A 453mA 350kHz OK More jitter during switching, but regulates normally.
Open Pin 10 (SENSE) 0V 0A 0.7mA 0kHz OK SENSE internally prebiases to 0.6V below VIN. This prevents
controller from switching.
Open Pin 11 (VIN) 5.4V 1A 597mA 20kHz OK VIN able to bias internally through SENSE. Regulates with high VOUT
ripple.
Open Pin 12 (GATE) 0V 0A 0.7mA 0kHz OK Gate does not drive external power FET, preventing output regulation.
Open Pin 13 (PGND) 5V 453mA 350kHz OK Pin 3 (GND) still provides GND connection to device.
Pins Shorted
Short Pins 1, 2
(PLLIN/MODE and FREQ) 5V 1A 453mA 350kHz OK Burst Mode operation disabled, but runs normally as in pulse-skipping
mode.
Short Pins 2, 3
(FREQ and GND) 5V 1A 453mA 0kHz OK FREQ already shorted to GND, so regulates normally.
Short Pins 3, 4
(GND and SS) 0V 0A 0.7mA 0kHz OK SS short to GND prevents device from starting up.
Short Pins 4, 5
(SS and VFB) 1V(DC)
3VP-P
50mA 9mA Erratic OK VOUT oscillates from 0V to 3V.
Short Pins 5, 6
(VFB and ITH) 3.15V 625mA 181mA 350kHz OK Controller loop does not regulate to proper output voltage.
Short Pins 7, 8
(PGOOD and RUN) 5V 1A 453mA 350kHz OK Controller does not start-up.
Short Pins 8, 9
(RUN and CAP) 5V 1A 453mA 350kHz OK Able to start-up and regulate normally.
Short Pins 9, 10
(CAP and SENSE) 0V 0A 181mA 0kHz OK CAP ~ VIN, which prevents turning on external P-MOSFET.
Short Pins 10, 11
(SENSE and VIN) 5V 1A 453mA 50kHz OK Regulates with high VOUT ripple.
Short Pins 11, 12
(VIN and GATE) 0V 0A 29mA 0kHz OK Power MOSFET is always kept OFF, preventing regulation.
LTC3864
26
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For more information www.linear.com/LTC3864
Typical applicaTions
24V to 60V Input, 24V/1A Output at 750kHz
Efficiency
3.5V to 48V Input, 1.8V/4A Output at 100kHz
Efficiency
CSS
0.1µF
RPGD
100k
MP
D1
L1
10µH
CAP
CCAP
0.47µF CIN2
10µF
×2
CIN1
33µF
63V
PGND
LTC3864
3864 TA03a
SS
ITH
FREQ
SGND
RUN VIN
MODE/PLLN
SENSE
GATE
PGOOD
VFB
RSENSE
15mΩ
RITH 14k
RFREQ 24.3k RFB2
102k
100µF
×2
VOUT
1.8V
4A
330µF
6.3V
RFB1
80.6k
VIN
3.5V TO 48V
CITH1
10nF
+
CIN1: SANYO 63ME33AX
D1: VISHAY V10P10
L1: WÜRTH 7447709100
MP: VISHAY/SILICONIX SI7461DP
CVIN
0.1µF
+
CITH2 100pF
RPGD2
768k
MP
D1
L1
47µH
CAP
CCAP
0.47µF CIN2
2.2µF
CIN1
33µF
63V
PGND
LTC3864
3864 TA02a
SS
ITH
CIN1: NICHICON UPJ1J100MPD
D1: DIODES INC SBR3U100LP
L1: TOKO 1217AS-H-470M
MP: VISHAY/SILICONIX SI7113DN *VOUT FOLLOWS VIN WHEN 3.5V ≤ VIN ≤ 24V
RUN VIN
MODE/PLLN
SENSE
GATE
PGOOD
VFB
RSENSE
50mΩ
RITH 30.1k
FREQ
SGND
RFREQ 97.6k RFB2
887k
10µF
VOUT*
24V
1A
RFB1
30.1k
VIN
24V TO 60V
CVIN
0.1µF
CITH1
6.8nF
CITH2 100pF
RPGD1
200k
+
LOAD CURRENT (A)
0.01
30
EFFICIENCY (%)
80
70
60
50
40
90
100
0.1 1
3864 TA02b
PULSE-SKIPPING
Burst Mode
OPERATION
VIN = 48V
VOUT = 24V
LOAD CURRENT (A)
0.01
30
EFFICIENCY (%)
70
60
50
40
80
0.1 1
3864 TA03b
PULSE-SKIPPING
Burst Mode
OPERATION
VIN = 12V
VOUT = 1.8V
LTC3864
27
3864fa
For more information www.linear.com/LTC3864
RPGD2
549k
MP
D1
L1
22µH
CAP
CCAP
0.47µF CIN2
4.7µF
CIN1
33µF
63V
PGND
*VOUT FOLLOWS VIN WHEN 3.5V ≤ VIN ≤ 12V
LTC3864
3864 TA04a
SS
ITH
FREQ
SGND
RUN VIN
MODE/PLLN
SENSE
GATE
PGOOD
VFB
RSENSE
30mΩ
RITH 11.3k
RFB2
845k
10µF
×2
VOUT*
12V
2A
RFB1
60.4k
VIN
12V TO 58V
CVIN
0.1µF
CITH1
3300pF
CITH2 100pF
+
CIN1: SANYO 63ME33AX
D1: DIODES INC SBR3U100LP
L1: TOKO 1217AS-H-220M
MP: VISHAY/SILICONIX SI7465DP
RPGD1
402k
Typical applicaTions
12V to 58V Input, 12V/2A Output at 535kHz
Efficiency
LOAD CURRENT (A)
0.01
50
EFFICIENCY (%)
80
70
60
90
0.1 1
3864 TA04b
PULSE-SKIPPING
Burst Mode
OPERATION
VIN = 48V
VOUT = 12V
DE/UE Package
12-Lead Plastic DFN (4mm × 3mm)
(Reference LTC DWG # 05-08-1695 Rev D)
4.00 ±0.10
(2 SIDES)
3.00 ±0.10
(2 SIDES)
NOTE:
1. DRAWING PROPOSED TO BE A VARIATION OF VERSION
(WGED) IN JEDEC PACKAGE OUTLINE M0-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
0.40 ± 0.10
BOTTOM VIEW—EXPOSED PAD
1.70 ± 0.10
0.75 ±0.05
R = 0.115
TYP
R = 0.05
TYP
2.50 REF
16
127
PIN 1 NOTCH
R = 0.20 OR
0.35 × 45°
CHAMFER
PIN 1
TOP MARK
(NOTE 6)
0.200 REF
0.00 – 0.05
(UE12/DE12) DFN 0806 REV D
3.30 ±0.10
0.25 ± 0.05 0.50 BSC
2.50 REF
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
2.20 ±0.05
0.70 ±0.05
3.60 ±0.05
PACKAGE
OUTLINE
1.70 ± 0.05
3.30 ±0.05
0.50 BSC
0.25 ± 0.05
package DescripTion
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
LTC3864
28
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For more information www.linear.com/LTC3864
package DescripTion
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
MSOP (MSE12) 0911 REV F
0.53 ±0.152
(.021 ±.006)
SEATING
PLANE
0.18
(.007)
1.10
(.043)
MAX
0.22 –0.38
(.009 – .015)
TYP
0.86
(.034)
REF
0.650
(.0256)
BSC
12
12 11 10 9 8 7
7
DETAIL “B”
16
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
0.254
(.010) 0° – 6° TYP
DETAIL “A”
DETAIL “A”
GAUGE PLANE
RECOMMENDED SOLDER PAD LAYOUT
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ±0.102
(.112 ±.004)
2.845 ±0.102
(.112 ±.004)
4.039 ±0.102
(.159 ±.004)
(NOTE 3)
1.651 ±0.102
(.065 ±.004)
1.651 ±0.102
(.065 ±.004)
0.1016 ±0.0508
(.004 ±.002)
1 2 3 4 5 6
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
0.406 ±0.076
(.016 ±.003)
REF
4.90 ±0.152
(.193 ±.006)
DETAIL “B”
CORNER TAIL IS PART OF
THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
0.12 REF
0.35
REF
5.23
(.206)
MIN
3.20 – 3.45
(.126 – .136)
0.889 ±0.127
(.035 ±.005)
0.42 ±0.038
(.0165 ±.0015)
TYP
0.65
(.0256)
BSC
MSE Package
12-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1666 Rev F)
LTC3864
29
3864fa
For more information www.linear.com/LTC3864
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
revision hisTory
REV DATE DESCRIPTION PAGE NUMBER
A 6/14 Modified VIN to CAP capacitance
Updated Notes 2 and 3
1, 8, 10, 21,
25, 26, 28
2
LTC3864
30
3864fa
For more information www.linear.com/LTC3864
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
LINEAR TECHNOLOGY CORPORATION 2012
LT 0614 REV A • PRINTED IN USA
LOAD CURRENT (A)
0.01
40
EFFICIENCY (%)
80
70
60
50
90
0.1 1
3864 TA05b
PULSE-SKIPPING
Burst Mode
OPERATION
VIN = 12V
VOUT = 3.3V
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LTC3864
relaTeD parTs
Typical applicaTion
PART NUMBER DESCRIPTION COMMENTS
LTC3891 60V, Low IQ, Synchronous Step-Down DC/DC
Controller Phase-Lockable Fixed Frequency 50kHz to 900kHz 4V ≤ VIN ≤ 60V,
0.8V ≤ VOUT ≤ 24V, IQ = 50µA
LTC3890 60V, Low IQ, Dual 2-Phase Synchronous
Step-Down DC/DC Controller Phase-Lockable Fixed Frequency 50kHz to 900kHz 4V ≤ VIN ≤ 60V,
0.8V ≤ VOUT ≤ 24V, IQ = 50µA
LTC3824 60V, Low IQ, Step-Down DC/DC Controller, 100%
Duty Cycle Selectable Fixed Frequency 200kHz to 600kHz 4V≤ VIN ≤ 60V,
0.8V ≤ VOUT ≤ VIN, IQ = 40µA, MSOP-10E
LT3845A 60V, Low IQ, Single Output Synchronous
Step-Down DC/DC Controller Synchronizable Fixed Frequency 100kHz to 600kHz 4V ≤ VIN ≤ 60V,
1.23V ≤ VOUT ≤ 36V, IQ = 120µA, TSSOP-16
LTC3863 60V Low IQ Inverting DC/DC Controller PLL Fixed Frequency 75kHz to 750kHz, 3.5V ≤ VIN ≤ 60V –150V ≤ VOUT ≤
–0.4V, IQ = 70µA, 3mm × 4mm DFN-12, MSOP-12
LTC3834/LTC3834-1
LTC3835/LTC3835-1 Low IQ, Single Output Synchronous Step-Down
DC/DC Controller with 99% Duty Cycle Phase-Lockable Fixed Frequency 140kHz to 650kHz, 4V ≤ VIN ≤ 36V,
0.8V ≤ VOUT ≤ 10V, IQ = 30µA/80µA
LTC3857/LTC3857-1
LTC3858/LTC3858-1 Low IQ, Dual Output 2-Phase Synchronous
Step-Down DC/DC Controllers with 99% Duty
Cycle
Phase-Lockable Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 38V,
0.8V ≤ VOUT ≤ 24V, IQ = 50µA/170µA
LTC3859AL Low IQ, Triple Output Buck/Buck/Boost
Synchronous DC/DC Controller All Outputs Remain in Regulation Through Cold Crank 2.5V ≤ VIN ≤ 38V,
VOUT(BUCKS) Up to 24V, VOUT(BOOST) Up to 60V, IQ = 28µA
3.5V to 38V Input, 3.3V/3A Output at 300kHz
Efficiency
RPGD
100k
MP
D1
L1
6.8µH
CAP
CCAP
0.47µF
CVIN
0.1µF
CIN1
33µF
63V
PGND
LTC3864
3864 TA05a
SS
ITH
FREQ
SGND
RUN VIN
MODE/PLLN
SENSE
GATE
PGOOD
VFB
RSENSE
20mΩ
RITH 20k
RFREQ 42.2k RFB2
634k
47µF
×2
VOUT
3.3V
3A
RFB1
200k
VIN
3.5V TO 60V
CITH1
10nF
CITH2 100pF
CSS
0.1µF
+
CIN2
10µF
×2
CIN1: SANYO 63ME33AX
D1: VISHAY V15P45S
L1: WÜRTH 7447709100
MP: VISHAY/SILICONIX Si7611DN
