MAX12553ETL+T - Maxim Integrated Products, Inc.

General Description

The MAX12553 is a 3.3V, 14-bit, 65Msps analog-to-digital

converter (ADC) featuring a fully differential wideband

track-and-hold (T/H) input amplifier, driving a low-noise

internal quantizer. The analog input stage accepts single-

ended or differential signals. The MAX12553 is optimized

for low-power, small size, and high dynamic perfor-

mance. Excellent dynamic performance is maintained

from baseband to input frequencies of 175MHz and

beyond, making the MAX12553 ideal for intermediate-

frequency (IF) sampling applications.

Powered from a single 3.15V to 3.60V supply, the

MAX12553 consumes only 363mW while delivering a

typical signal-to-noise (SNR) performance of 71dB at

an input frequency of 175MHz. In addition to low oper-

ating power, the MAX12553 features a 150µW power-

down mode to conserve power during idle periods.

A flexible reference structure allows the MAX12553 to use

the internal 2.048V bandgap reference or accept an

externally applied reference. The reference structure

allows the full-scale analog input range to be adjusted

from ±0.35V to ±1.10V. The MAX12553 provides a com-

mon-mode reference to simplify design and reduce exter-

nal component count in differential analog input circuits.

The MAX12553 supports both a single-ended and dif-

ferential input clock drive. Wide variations in the clock

duty cycle are compensated with the ADC’s internal

duty-cycle equalizer (DCE).

ADC conversion results are available through a 14-bit,

parallel, CMOS-compatible output bus. The digital out-

put format is pin selectable to be either two’s comple-

ment or Gray code. A data-valid indicator eliminates

external components that are normally required for reli-

able digital interfacing. A separate digital power input

accepts a wide 1.7V to 3.6V supply, allowing the

MAX12553 to interface with various logic levels.

The MAX12553 is available in a 6mm x 6mm x 0.8mm,

40-pin thin QFN package with exposed paddle (EP),

and is specified for the extended industrial (-40°C to

+85°C) temperature range.

See the Pin-Compatible Versions table for a complete

family of 14-bit and 12-bit high-speed ADCs.

Applications

IF and Baseband Communication Receivers

Cellular, Point-to-Point Microwave, HFC, WLAN

Ultrasound and Medical Imaging

Portable Instrumentation

Low-Power Data Acquisition

Features

♦Direct IF Sampling Up to 400MHz

♦Excellent Dynamic Performance

74.0dB/71dB SNR at fIN = 3MHz/175MHz

90.6dBc/80.7dBc SFDR at fIN = 3MHz/175MHz

♦Low Noise Floor: -76dBFS

♦3.3V Low-Power Operation

337mW (Single-Ended Clock Mode)

363mW (Differential Clock Mode)

150µW (Power-Down Mode)

♦Fully Differential or Single-Ended Analog Input

♦Adjustable Full-Scale Analog Input Range: ±0.35V

to ±1.10V

♦Common-Mode Reference

♦CMOS-Compatible Outputs in Two’s Complement

or Gray Code

♦Data-Valid Indicator Simplifies Digital Interface

♦Data Out-of-Range Indicator

♦Miniature, 40-Pin Thin QFN Package with Exposed

Paddle

♦Evaluation Kit Available (Order MAX12555EVKIT)

MAX12553

14-Bit, 65Msps, 3.3V ADC

________________________________________________________________ Maxim Integrated Products 1

Ordering Information

19-3343; Rev 0; 8/04

For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at

1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.

EVALUATION KIT

AVAILABLE

PART TEMP

RANGE PIN-PACKAGE PKG

CODE

MAX12553ETL -40°C to

+85°C

40 Thin QFN

(6mm x 6mm x 0.8mm) T4066-3

Pin-Compatible Versions

PART SAMPLING

RATE (Msps)

RESOLUTION

(BITS)

TARGET

APPLICATION

MAX12553 65 14 IF/Baseband

MAX1209 80 12 IF

MAX1211 65 12 IF

MAX1208 80 12 Baseband

MAX1207 65 12 Baseband

MAX1206 40 12 Baseband

Pin Configuration appears at end of data sheet.

MAX12553

14-Bit, 65Msps, 3.3V ADC

2_______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS

(VDD = 3.3V, OVDD = 2.0V, GND = 0, REFIN = REFOUT (internal reference), VIN = -0.5dBFS, CLKTYP = high, DCE = high, PD = low,

G/T= low, fCLK = 65MHz (50% duty cycle), TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional

operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to

absolute maximum rating conditions for extended periods may affect device reliability.

VDD to GND...........................................................-0.3V to +3.6V

OVDD to GND........-0.3V to the lower of (VDD + 0.3V) and +3.6V

INP, INN to GND ...-0.3V to the lower of (VDD + 0.3V) and +3.6V

REFIN, REFOUT, REFP, REFN, COM

to GND................-0.3V to the lower of (VDD + 0.3V) and +3.6V

CLKP, CLKN, CLKTYP, G/T, DCE,

PD to GND ........-0.3V to the lower of (VDD + 0.3V) and +3.6V

D13–D0, DAV, DOR to GND....................-0.3V to (OVDD + 0.3V)

Continuous Power Dissipation (TA= +70°C)

40-Pin Thin QFN 6mm x 6mm x 0.8mm

(derated 26.3mW/°C above +70°C)........................2105.3mW

Operating Temperature Range ...........................-40°C to +85°C

Junction Temperature......................................................+150°C

Storage Temperature Range .............................-65°C to +150°C

Lead Temperature (soldering 10s) ..................................+300°C

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

DC ACCURACY (Note 2)

Resolution 14 Bits

Integral Nonlinearity INL fIN = 3MHz (Note 5) ±1.4 ±4.2 LSB

Differential Nonlinearity DNL fIN = 3MHz, no missing codes over

temperature (Note 3) ±0.5 ±1.0 LSB

Offset Error VREFIN = 2.048V ±0.1 ±0.55 %FS

Gain Error VREFIN = 2.048V ±0.5 ±4.9 %FS

ANALOG INPUT (INP, INN)

Differential Input Voltage Range VDIFF Differential or single-ended inputs ±1.024 V

Common-Mode Input Voltage VDD/2 V

CPAR Fixed capacitance to ground 2

Input Capacitance

(Figure 3) CSAMPLE Switched capacitance 4.5 pF

CONVERSION RATE

Maximum Clock Frequency fCLK 65 MHz

Minimum Clock Frequency 5MHz

Data Latency Figure 6 8.5 Clock

cycles

DYNAMIC CHARACTERISTICS (differential inputs, Note 2)

Small-Signal Noise Floor SSNF Input at less than -35dBFS -76.0 dBFS

fIN = 3MHz at -0.5dBFS (Note 8) 69.3 74.0

fIN = 32.5MHz at -0.5dBFS 73.9

fIN = 70MHz at -0.5dBFS 73.4

Signal-to-Noise Ratio SNR

fIN = 175MHz at -0.5dBFS (Notes 7, 8) 68.0 71.0

fIN = 3MHz at -0.5dBFS (Note 8) 69.2 73.9

fIN = 32.5MHz at -0.5dBFS 73.1

fIN = 70MHz at -0.5dBFS 73.1

Signal-to-Noise and Distortion SINAD

fIN = 175MHz at -0.5dBFS (Notes 7, 8) 67.6 70.0

MAX12553

14-Bit, 65Msps, 3.3V ADC

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS (continued)

(VDD = 3.3V, OVDD = 2.0V, GND = 0, REFIN = REFOUT (internal reference), VIN = -0.5dBFS, CLKTYP = high, DCE = high, PD = low,

G/T= low, fCLK = 65MHz (50% duty cycle), TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

fIN = 3MHz at -0.5dBFS 79.8 90.6

fIN = 32.5MHz at -0.5dBFS 84.0

fIN = 70MHz at -0.5dBFS 87.8

Spurious-Free Dynamic Range SFDR

fIN = 175MHz at -0.5dBFS (Note 7) 75.9 80.7

dBc

fIN = 3MHz at -0.5dBFS -90.6 -80.2

fIN = 32.5MHz at -0.5dBFS -81.0

fIN = 70MHz at -0.5dBFS -85.4

Total Harmonic Distortion THD

fIN = 175MHz at -0.5dBFS -78.9 -71.3

dBc

fIN = 3MHz at -0.5dBFS -99

fIN = 32.5MHz at -0.5dBFS -91

fIN = 70MHz at -0.5dBFS -92

Second Harmonic HD2

fIN = 175MHz at -0.5dBFS -81

dBc

fIN = 3MHz at -0.5dBFS -94

fIN = 32.5MHz at -0.5dBFS -84

fIN = 70MHz at -0.5dBFS -88

Third Harmonic HD3

fIN = 175MHz at -0.5dBFS -86

dBc

fIN1 = 68.5MHz at -7dBFS

fIN2 = 71.5MHz at -7dBFS -87

Intermodulation Distortion IMD fIN1 = 172.5MHz at -7dBFS

fIN2 = 177.5MHz at -7dBFS -80

dBc

fIN1 = 68.5MHz at -7dBFS

fIN2 = 71.5MHz at -7dBFS -91

Third-Order Intermodulation IM3 fIN1 = 172.5MHz at -7dBFS

fIN2 = 177.5MHz at -7dBFS -83

dBc

fIN1 = 68.5MHz at -7dBFS

fIN2 = 71.5MHz at -7dBFS 90

Two-Tone Spurious-Free

Dynamic Range SFDRTT fIN1 = 172.5MHz at -7dBFS

fIN2 = 177.5MHz at -7dBFS 81

dBc

Aperture Delay tAD Figure 4 1.2 ns

Aperture Jitter tAJ Figure 4 <0.2 psRMS

Output Noise nOUT INP = INN = COM 0.95 LSBRMS

Overdrive Recovery Time ±10% beyond full scale 1 Clock

cycles

MAX12553

14-Bit, 65Msps, 3.3V ADC

4_______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(VDD = 3.3V, OVDD = 2.0V, GND = 0, REFIN = REFOUT (internal reference), VIN = -0.5dBFS, CLKTYP = high, DCE = high, PD = low,

G/T= low, fCLK = 65MHz (50% duty cycle), TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

INTERNAL REFERENCE (REFIN = REFOUT; VREFP, VREFN, and VCOM are generated internally)

REFOUT Output Voltage VREFOUT 2.002 2.048 2.066 V

COM Output Voltage VCOM VDD/2 1.65 V

Differential Reference Output

Voltage VREF VREF = VREFP - VREFN = VREFIN x 3/4 1.536 V

REFOUT Load Regulation 35 mV/mA

REFOUT Temperature Coefficient TCREF +50 ppm/°C

Short to VDD—sinking 0.24

REFOUT Short-Circuit Current Short to GND—sourcing 2.1 mA

B U F F ER ED EXT ER N A L R EF ER EN C E ( R EF IN d r iv e n e x t e r n a lly ; VR EF IN

= 2.0 4 8 V, VR EF P, VR EF N

, a n d VC OM

a r e g e n e r a t e d in t e r n a lly )

REFIN Input Voltage VREFIN 2.048 V

REFP Output Voltage VREFP (VDD/2) + (VREFIN x 3/8) 2.418 V

REFN Output Voltage VREFN (VDD/2) - (VREFIN x 3/8) 0.882 V

COM Output Voltage VCOM VDD/2 1.60 1.65 1.70 V

Differential Reference Output

Voltage VREF VREF = VREFP - VREFN = VREFIN x 3/4 1.463 1.536 1.601 V

Differential Reference

Temperature Coefficient ±25 ppm/°C

REFIN Input Resistance >50 MΩ

UNBUFFERED EXTERNAL REFERENCE (REFIN = GND; VREFP, VREFN, and VCOM are applied externally)

COM Input Voltage VCOM VDD/2 1.65 V

REFP Input Voltage VREFP - VCOM 0.768 V

REFN Input Voltage VREFN - VCOM -0.768 V

Differential Reference Input

Voltage VREF VREF = VREFP - VREFN = VREFIN x 3/4 1.536 V

REFP Sink Current IREFP VREFP = 2.418V 1 mA

REFN Source Current IREFN VREFN = 0.882V 0.7 mA

COM Sink Current ICOM 0.7 mA

REFP, REFN Capacitance 13 pF

COM Capacitance 6pF

CLOCK INPUTS (CLKP, CLKN)

Single-Ended Input High

Threshold VIH CLKTYP = GND, CLKN = GND 0.8 x

VDD V

Single-Ended Input Low

Threshold VIL CLKTYP = GND, CLKN = GND 0.2 x

VDD V

Differential Input Voltage Swing CLKTYP = high 1.4 VP-P

Differential Input Common-Mode

Voltage CLKTYP = high VDD / 2 V

MAX12553

14-Bit, 65Msps, 3.3V ADC

_______________________________________________________________________________________ 5

ELECTRICAL CHARACTERISTICS (continued)

(VDD = 3.3V, OVDD = 2.0V, GND = 0, REFIN = REFOUT (internal reference), VIN = -0.5dBFS, CLKTYP = high, DCE = high, PD = low,

G/T= low, fCLK = 65MHz (50% duty cycle), TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

Input Resistance RCLK Figure 5 5 kΩ

Input Capacitance CCLK 2pF

DIGITAL INPUTS (CLKTYP, G/T, PD)

Input High Threshold VIH 0.8 x

OVDD V

Input Low Threshold VIL 0.2 x

OVDD V

VIH = OVDD ±5

Input Leakage Current VIL = 0 ±5µA

Input Capacitance CDIN 5pF

DIGITAL OUTPUTS (D13–D0, DAV, DOR)

D13–D0, DOR, ISINK = 200µA 0.2

Output Voltage Low VOL DAV, ISINK = 600µA 0.2 V

D13–D0, DOR, ISOURCE = 200µA OVDD -

0.2

Output Voltage High VOH

DAV, ISOURCE = 600µA OVDD -

0.2

Tri-State Leakage Current ILEAK (Note 4) ±5 µA

D13–D0, DOR Tri-State Output

Capacitance COUT (Note 4) 3 pF

DAV Tri-State Output

Capacitance CDAV (Note 4) 6 pF

POWER REQUIREMENTS

Analog Supply Voltage VDD 3.15 3.3 3.60 V

Digital Output Supply Voltage OVDD 1.7 2.0 VDD +

0.3V V

Normal operating mode,

fIN = 175MHz at -0.5dBFS, CLKTYP = GND,

single-ended clock

102

Normal operating mode,

fIN = 175MHz at -0.5dBFS,

CLKTYP = OVDD, differential clock

110 123

Analog Supply Current IVDD

Power-down mode clock idle, PD = OVDD 0.045

MAX12553

14-Bit, 65Msps, 3.3V ADC

6_______________________________________________________________________________________

Note 1: Specifications ≥+25°C guaranteed by production test; <+25°C guaranteed by design and characterization.

Note 2: See definitions in the Parameter Definitions section at the end of this data sheet.

Note 3: Specifications guaranteed by design and characterization. Devices tested for performance during production test.

Note 4: During power-down, D13–D0, DOR, and DAV are high impedance.

Note 5: Guaranteed by design and characterization.

Note 6: Digital outputs settle to VIH or VIL.

Note 7: Due to test-equipment-jitter limitations at 175MHz, 0.15% of the spectrum on each side of the fundamental is excluded from

the spectral analysis.

Note 8: Limit specifications include performance degradations due to a production test socket. Performance is improved when the

MAX12553 is soldered directly to the PC board.

ELECTRICAL CHARACTERISTICS (continued)

(VDD = 3.3V, OVDD = 2.0V, GND = 0, REFIN = REFOUT (internal reference), VIN = -0.5dBFS, CLKTYP = high, DCE = high, PD = low,

G/T= low, fCLK = 65MHz (50% duty cycle), TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

Normal operating mode,

fIN = 175MHz at -0.5dBFS, CLKTYP = GND,

single-ended clock

337

Normal operating mode,

fIN = 175MHz at -0.5dBFS,

CLKTYP = OVDD, differential clock

363 406

Analog Power Dissipation PDISS

Power-down mode clock idle, PD = OVDD 0.15

Normal operating mode,

fIN = 175MHz at -0.5dBFS, OVDD = 2.0V,

CL ≈ 5pF

8.2 mA

Digital Output Supply Current IOVDD

Power-down mode clock idle, PD = OVDD 20 µA

TIMING CHARACTERISTICS (Figure 6)

Clock Pulse-Width High tCH 7.7 ns

Clock Pulse-Width Low tCL 7.7 ns

Data-Valid Delay tDAV CL = 5pF (Note 6) 6.9 ns

Data Setup Time Before Rising

Edge of DAV tSETUP CL = 5pF (Notes 5, 6) 8.5 ns

Data Hold Time After Rising Edge

of DAV tHOLD CL = 5pF (Notes 5, 6) 6.3 ns

Wake-Up Time from Power-Down tWAKE VREFIN = 2.048V 10 ms

MAX12553

14-Bit, 65Msps, 3.3V ADC

_______________________________________________________________________________________ 7

SINGLE-TONE FFT PLOT

(8192-POINT DATA RECORD)

MAX12553 toc01

FREQUENCY (MHz)

AMPLITUDE (dBFS)

24 2881216204

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

-120

-110

032

fCLK = 65MHz

fIN = 3.00720215MHz

AIN = -0.542dBFS

SNR = 74.223dB

SINAD = 74.147dB

THD = -91.794dBc

SFDR = 91.499dBc

HD3

HD2

SINGLE-TONE FFT PLOT

(8192-POINT DATA RECORD)

MAX12553 toc02

FREQUENCY (MHz)

AMPLITUDE (dBFS)

24 2881216204

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

-120

-110

032

fCLK = 65MHz

fIN = 32.39685059MHz

AIN = -0.469dBFS

SNR = 74.206dB

SINAD = 73.534dB

THD = -81.965dBc

SFDR = 86.015dBc

HD3

HD2

SINGLE-TONE FFT PLOT

(8192-POINT DATA RECORD)

MAX12553 toc03

FREQUENCY (MHz)

AMPLITUDE (dBFS)

24 2881216204

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

-120

-110

032

fCLK = 65MHz

fIN = 69.89562988MHz

AIN = -0.460dBFS

SNR = 73.772dB

SINAD = 73.615dB

THD = -88.110dBc

SFDR = 88.325dBc

HD3

HD2

SINGLE-TONE FFT PLOT

(8192-POINT DATA RECORD)

MAX12553 toc04

FREQUENCY (MHz)

AMPLITUDE (dBFS)

24 2881216204

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

-120

-110

032

fCLK = 65MHz

fIN = 174.9017334MHz

AIN = -0.499dBFS

SNR = 70.971dB

SINAD = 70.260dB

THD = -78.475dBc

SFDR = 80.267dBc

HD5

HD2

SINGLE-TONE FFT PLOT

(8192-POINT DATA RECORD)

MAX12553 toc05

FREQUENCY (MHz)

AMPLITUDE (dBFS)

24 2881216204

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

-120

-110

032

fCLK = 64.96256MHz

fIN = 250.00911MHz

AIN = -0.494dBFS

SNR = 69.39dB

SINAD = 68.67dB

THD = -76.8dBc

SFDR = 78.6dBc

HD2 HD3

TWO-TONE FFT PLOT

(16,384-POINT DATA RECORD)

MAX12553 toc06

FREQUENCY (MHz)

AMPLITUDE (dBFS)

24 2881216204

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

-120

-110

032

fCLK = 65MHz

fIN1 = 68.50311279MHz

AIN1 = -7.018dBFS

fIN2 = 71.50238037MHz

AIN2 = -7.087dBFS

SFDRTT = 90.085dBc

IMD = -87.812dBc

IM3 = -91.844dBc

fIN1

fIN2

2 x fIN2 - fIN1

TWO-TONE FFT PLOT

(16,384-POINT DATA RECORD)

MAX12553 toc07

FREQUENCY (MHz)

AMPLITUDE (dBFS)

24 2881216204

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

-120

-110

032

fCLK = 65.00352MHz

fIN1 = 172.4870625MHz

AIN1 = -7.047dBFS

fIN2 = 177.4861125MHz

AIN2 = -6.984dBFS

SFDRTT = 81.484dBc

IMD = -80.035dBc

IM3 = -83.511dBc

fIN1

fIN2

2 x fIN2 - fIN1fIN2 - fIN1 fIN1 + fIN2

INTEGRAL NONLINEARITY

MAX12553 toc08

DIGITAL OUTPUT CODE

INL (LSB)

1228881924096

-1.5

-1.0

-0.5

0.5

1.0

1.5

2.0

-2.0

0 16384

DIFFERENTIAL NONLINEARITY

MAX12553 toc09

DIGITAL OUTPUT CODE

DNL (LSB)

1228881924096

-0.4

-0.6

-0.8

-0.2

0.2

0.4

0.6

0.8

1.0

-1.0

0 16384

Typical Operating Characteristics

(VDD = 3.3V, OVDD = 2.0V, GND = 0, REFIN = REFOUT (internal reference), VIN = -0.5dBFS, CLKTYP = high, DCE = high, PD = low,

G/T= low, fCLK ≈65MHz (50% duty cycle), TA = +25°C, unless otherwise noted.)

MAX12553

14-Bit, 65Msps, 3.3V ADC

8_______________________________________________________________________________________

SNR, SINAD

vs. SAMPLING RATE

MAX12553 toc10

fCLK (MHz)

SNR, SINAD (dB)

604020

080

fIN ≈ 70MHz

SNR

SINAD

SFDR, -THD

vs. SAMPLING RATE

MAX12553 toc11

fCLK (MHz)

SFDR, -THD (dB)

604020

100

080

fIN ≈ 70MHz

SFDR

-THD

POWER DISSIPATION

vs. SAMPLING RATE

MAX12553 toc12

fCLK (MHz)

POWER DISSIPATION (mW)

604020

250

300

350

400

450

500

200

080

DIFFERENTIAL CLOCK

fIN ≈ 70MHz

CL ≈ 5pF

ANALOG + DIGITAL POWER

ANALOG POWER

SNR, SINAD

vs. SAMPLING RATE

MAX12553 toc13

fCLK (MHz)

SNR, SINAD (dB)

604020

080

fIN ≈ 175MHz

SNR

SINAD

SFDR, -THD

vs. SAMPLING RATE

MAX12553 toc14

fCLK (MHz)

SFDR, -THD (dB)

604020

100

080

fIN ≈ 175MHz

SFDR

-THD

POWER DISSIPATION

vs. SAMPLING RATE

MAX12553 toc15

fCLK (MHz)

POWER DISSIPATION (mW)

604020

250

300

350

400

450

500

200

080

DIFFERENTIAL CLOCK

fIN ≈ 175MHz

CL ≈ 5pF

ANALOG + DIGITAL POWER

ANALOG POWER

SNR, SINAD

vs. ANALOG INPUT FREQUENCY

MAX12553 toc16

ANALOG INPUT FREQUENCY (MHz)

SNR, SINAD (dB)

300200100

0 400

fCLK ≈ 65MHz

SNR

SINAD

SFDR, -THD

vs. ANALOG INPUT FREQUENCY

MAX12553 toc17

ANALOG INPUT FREQUENCY (MHz)

SFDR, -THD (dBc)

300200100

0 400

fCLK ≈ 65MHz

SFDR

-THD

POWER DISSIPATION

vs. ANALOG INPUT FREQUENCY

MAX12553 toc18

ANALOG INPUT FREQUENCY (MHz)

POWER DISSIPATION (mW)

300200100

250

300

350

400

450

500

200

0 400

ANALOG + DIGITAL POWER

ANALOG POWER

DIFFERENTIAL CLOCK

fCLK ≈ 65MHz

CL ≈ 5pF

Typical Operating Characteristics (continued)

(VDD = 3.3V, OVDD = 2.0V, GND = 0, REFIN = REFOUT (internal reference), VIN = -0.5dBFS, CLKTYP = high, DCE = high, PD = low,

G/T= low, fCLK ≈65MHz (50% duty cycle), TA = +25°C, unless otherwise noted.)

MAX12553

14-Bit, 65Msps, 3.3V ADC

_______________________________________________________________________________________ 9

SNR, SINAD

vs. ANALOG INPUT AMPLITUDE

MAX12553 toc19

ANALOG INPUT AMPLITUDE (dBFS)

SNR, SINAD (dB)

-5-10-20 -15-30 -25-35

-40 0

fCLK = 64.96256MHz

fIN = 175.0071MHz

SNR

SINAD

SFDR, -THD

vs. ANALOG INPUT AMPLITUDE

MAX12553 toc20

ANALOG INPUT AMPLITUDE (dBFS)

SFDR, -THD (dBc)

-5-10-20 -15-30 -25-35

100

-40 0

fCLK = 64.96256MHz

fIN = 175.0071MHz

SFDR

-THD

POWER DISSIPATION

vs. ANALOG INPUT AMPLITUDE

MAX12553 toc21

ANALOG INPUT AMPLITUDE (dBFS)

POWER DISSIPATION (mW)

-5-10-20 -15-30 -25-35

250

300

350

400

450

500

200

-40 0

DIFFERENTIAL CLOCK

fCLK = 64.96256MHz

fIN = 175.0071MHz

CL ≈ 5pF

ANALOG + DIGITAL POWER

ANALOG POWER

SNR, SINAD

vs. ANALOG SUPPLY VOLTAGE

MAX12553 toc22

VDD (V)

SNR, SINAD (dB)

3.43.0 3.22.8

2.6 3.6

fCLK = 64.96256MHz

fIN = 175.00717MHz

SNR

SINAD

SFDR, -THD

vs. ANALOG SUPPLY VOLTAGE

MAX12553 toc23

VDD (V)

SFDR, -THD (dBc)

3.43.0 3.22.8

100

2.6 3.6

fCLK = 64.96256MHz

fIN = 175.00717MHz

SFDR

-THD

POWER DISSIPATION

vs. ANALOG SUPPLY VOLTAGE

MAX12553 toc24

VDD (V)

POWER DISSIPATION (mW)

3.43.0 3.22.8

250

300

350

400

450

500

200

2.6 3.6

DIFFERENTIAL CLOCK

fCLK = 64.96256MHz

fIN = 175.00717MHz

CL ≈ 5pF

ANALOG + DIGITAL POWER

ANALOG POWER

SNR, SINAD

vs. DIGITAL SUPPLY VOLTAGE

MAX12553 toc25

OVDD (V)

SNR, SINAD (dB)

3.43.02.62.21.8

1.4 3.8

fCLK = 65MHz

fIN = 174.9007416MHz

SNR

SINAD

SFDR, -THD

vs. DIGITAL SUPPLY VOLTAGE

MAX12553 toc26

OVDD (V)

SFDR, -THD (dBc)

3.43.02.62.21.8

100

1.4 3.8

fCLK = 65MHz

fIN = 174.9007416MHz

SFDR

-THD

POWER DISSIPATION

vs. DIGITAL SUPPLY VOLTAGE

MAX12553 toc27

OVDD (V)

POWER DISSIPATION (mW)

3.43.02.62.21.8

250

300

350

400

450

500

200

1.4 3.8

DIFFERENTIAL CLOCK

fCLK = 65MHz

fIN = 174.9007416MHz

CL ≈ 5pF

ANALOG + DIGITAL POWER

ANALOG POWER

Typical Operating Characteristics (continued)

(VDD = 3.3V, OVDD = 2.0V, GND = 0, REFIN = REFOUT (internal reference), VIN = -0.5dBFS, CLKTYP = high, DCE = high, PD = low,

G/T= low, fCLK ≈65MHz (50% duty cycle), TA = +25°C, unless otherwise noted.)

MAX12553

14-Bit, 65Msps, 3.3V ADC

10 ______________________________________________________________________________________

SNR, SINAD vs. TEMPERATURE

MAX12553 toc28

TEMPERATURE (°C)

SNR, SINAD (dB)

603510-15

-40 85

fCLK = 65MHz

fIN = 175MHz

SNR

SINAD

SFDR, -THD vs. TEMPERATURE

MAX12553 toc29

TEMPERATURE (°C)

SFDR, -THD (dBc)

603510-15

-40 85

fCLK = 65MHz

fIN = 175MHz

SFDR

-THD

ANALOG POWER DISSIPATION

vs. TEMPERATURE

MAX12553 toc30

TEMPERATURE (°C)

ANALOG POWER DISSIPATION (mW)

603510-15

250

300

350

400

450

500

200

-40 85

DIFFERENTIAL CLOCK

fCLK = 65MHz

fIN = 175MHz

OFFSET ERROR

vs. TEMPERATURE

MAX12553 toc31

TEMPERATURE (°C)

OFFSET ERROR (%FS)

603510-15

-0.2

-0.1

0.1

0.2

0.3

-0.3

-40 85

VREFIN = 2.048V

GAIN ERROR

vs. TEMPERATURE

MAX12553 toc32

TEMPERATURE (°C)

GAIN ERROR (%FS)

603510-15

-2

-1

-3

-40 85

VREFIN = 2.048V

Typical Operating Characteristics (continued)

(VDD = 3.3V, OVDD = 2.0V, GND = 0, REFIN = REFOUT (internal reference), VIN = -0.5dBFS, CLKTYP = high, DCE = high, PD = low,

G/T= low, fCLK ≈65MHz (50% duty cycle), TA = +25°C, unless otherwise noted.)

MAX12553

14-Bit, 65Msps, 3.3V ADC

______________________________________________________________________________________ 11

REFERENCE OUTPUT VOLTAGE

LOAD REGULATION

MAX12553 toc33

IREFOUT SINK CURRENT (mA)

VREFOUT (V)

0-0.5-1.0-1.5

1.96

1.97

1.98

1.99

2.00

2.01

2.02

2.03

2.04

2.05

1.95

-2.0 0.5

+85°C

+25°C

-40°C

REFERENCE OUTPUT VOLTAGE

SHORT-CIRCUIT PERFORMANCE

MAX12553 toc34

IREFOUT SINK CURRENT (mA)

VREFOUT (V)

0-1.0-2.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

-3.0 1.0

+85°C

+25°C

-40°C

REFERENCE OUTPUT VOLTAGE

vs. TEMPERATURE

MAX12553 toc35

TEMPERATURE (°C)

VREFOUT (V)

603510-15

2.031

2.033

2.035

2.037

2.039

2.029

-40 85

REFP, COM, REFN

LOAD REGULATION

MAX12553 toc36

SINK CURRENT (mA)

VOLTAGE (V)

10-1

0.5

1.0

1.5

2.0

2.5

3.0

-2 2

VREFP

VCOM

VREFN

INTERNAL REFERENCE

MODE AND BUFFERED EXTERNAL

REFERENCE MODE

REFP, COM, REFN

SHORT-CIRCUIT PERFORMACE

MAX12553 toc37

SINK CURRENT (mA)

VOLTAGE (V)

40-4

0.5

1.0

1.5

2.0

2.5

3.5

3.0

-8 128

VREFP

VCOM

VREFN

INTERNAL REFERENCE

MODE AND BUFFERED

EXTERNAL REFERENCE MODE

Typical Operating Characteristics (continued)

(VDD = 3.3V, OVDD = 2.0V, GND = 0, REFIN = REFOUT (internal reference), VIN = -0.5dBFS, CLKTYP = high, DCE = high, PD = low,

G/T= low, fCLK ≈65MHz (50% duty cycle), TA = +25°C, unless otherwise noted.)

MAX12553

14-Bit, 65Msps, 3.3V ADC

12 ______________________________________________________________________________________

PIN NAME FUNCTION

1REFP

Positive Reference I/O. The full-scale analog input range is ±(VREFP - VREFN) x 2/3. Bypass REFP to

GND with a 0.1µF capacitor. Connect a 1µF capacitor in parallel with a 10µF capacitor between REFP

and REFN. Place the 1µF REFP to REFN capacitor as close to the device as possible on the same

side of the PC board.

2REFN

Negative Reference I/O. The full-scale analog input range is ±(VREFP - VREFN) x 2/3. Bypass REFN to

GND with a 0.1µF capacitor. Connect a 1µF capacitor in parallel with a 10µF capacitor between REFP

and REFN. Place the 1µF REFP to REFN capacitor as close to the device as possible on the same

side of the PC board.

3COM

Common-Mode Voltage I/O. Bypass COM to GND with a 2.2µF capacitor. Place the 2.2µF COM to

GND capacitor as close to the device as possible. This 2.2µF capacitor can be placed on the

opposite side of the PC board and connected to the MAX12553 through a via.

4, 7, 16,

35 GND Ground. Connect all ground pins and EP together.

5INP Positive Analog Input

6INN Negative Analog Input

8DCE Duty-Cycle Equalizer Input. Connect DCE low (GND) to disable the internal duty-cycle equalizer.

Connect DCE high (OVDD or VDD) to enable the internal duty-cycle equalizer.

9CLKN

Negative Clock Input. In differential clock input mode (CLKTYP = OVDD or VDD), connect the differential

clock signal between CLKP and CLKN. In single-ended clock mode (CLKTYP = GND), apply the single-

ended clock signal to CLKP and connect CLKN to GND.

10 CLKP

Positive Clock Input. In differential clock input mode (CLKTYP = OVDD or VDD), connect the differential

clock signal between CLKP and CLKN. In single-ended clock mode (CLKTYP = GND), apply the single-

ended clock signal to CLKP and connect CLKN to GND.

11 CLKTYP Clock Type Definition Input. Connect CLKTYP to GND to define the single-ended clock input. Connect

CLKTYP to OVDD or VDD to define the differential clock input.

12–15, 36 VDD Analog Power Input. Connect VDD to a 3.15V to 3.60V power supply. Bypass VDD to GND with a parallel

capacitor combination of ≥2.2µF and 0.1µF. Connect all VDD pins to the same potential.

17, 34 OVDD Output-Driver Power Input. Connect OVDD to a 1.7V to VDD power supply. Bypass OVDD to GND with a

parallel capacitor combination of ≥2.2µF and 0.1µF.

18 DOR

Data Out-of-Range Indicator. The DOR digital output indicates when the analog input voltage is out of

range. When DOR is high, the analog input is beyond its full-scale range. When DOR is low, the analog

input is within its full-scale range (Figure 6).

19 D13 CMOS Digital Output, Bit 13 (MSB)

20 D12 CMOS Digital Output, Bit 12

21 D11 CMOS Digital Output, Bit 11

22 D10 CMOS Digital Output, Bit 10

23 D9 CMOS Digital Output, Bit 9

24 D8 CMOS Digital Output, Bit 8

25 D7 CMOS Digital Output, Bit 7

26 D6 CMOS Digital Output, Bit 6

27 D5 CMOS Digital Output, Bit 5

Pin Description

MAX12553

14-Bit, 65Msps, 3.3V ADC

______________________________________________________________________________________ 13

PIN NAME FUNCTION

28 D4 CMOS Digital Output, Bit 4

29 D3 CMOS Digital Output, Bit 3

30 D2 CMOS Digital Output, Bit 2

31 D1 CMOS Digital Output, Bit 1

32 D0 CMOS Digital Output, Bit 0 (LSB)

33 DAV

Data-Valid Output. DAV is a single-ended version of the input clock that is compensated to correct for

any input clock duty-cycle variations. DAV is typically used to latch the MAX12553 output data into an

external back-end digital circuit.

37 PD Power-Down Input. Force PD high for power-down mode. Force PD low for normal operation.

38 REFOUT

Internal Reference Voltage Output. For internal reference operation, connect REFOUT directly to REFIN

or use a resistive divider from REFOUT to set the voltage at REFIN. Bypass REFOUT to GND with a

≥0.1µF capacitor.

39 REFIN

Reference Input. In internal reference mode and buffered external reference mode, bypass REFIN to

GND with a ≥0.1µF capacitor. In these modes,VREFP - VREFN = VREFIN x 3/4. For unbuffered external

reference mode operation, connect REFIN to GND.

40 G/TOutput Format Select Input. Connect G/T to GND for the two’s complement digital output format.

Connect G/T to OVDD or VDD for the Gray code digital output format.

—EP

Exposed Paddle. The MAX12553 relies on the exposed paddle connection for a low-inductance ground

connection. Connect EP to GND to achieve specified performance. Use multiple vias to connect the

top-side PC board ground plane to the bottom-side PC board ground plane.

Pin Description (continued)

MAX12553 Σ

−

DIGITAL ERROR CORRECTION

FLASH

ADC

T/H

DAC

STAGE 2

D13–D0

INP

INN STAGE 1

T/H STAGE 9 STAGE 10

END OF PIPE

OUTPUT

DRIVERS

D13–D0

Figure 1. Pipeline Architecture—Stage Blocks

MAX12553

Detailed Description

The MAX12553 uses a 10-stage, fully differential,

pipelined architecture (Figure 1) that allows for high-

speed conversion while minimizing power consump-

tion. Samples taken at the inputs move progressively

through the pipeline stages every half-clock cycle.

From input to output, the total clock-cycle latency is 8.5

clock cycles.

Each pipeline converter stage converts its input voltage

into a digital output code. At every stage, except the

last, the error between the input voltage and the digital

output code is multiplied and passed along to the next

pipeline stage. Digital error correction compensates for

ADC comparator offsets in each pipeline stage and

ensures no missing codes. Figure 2 shows the

MAX12553 functional diagram.

Input Track-and-Hold (T/H) Circuit

Figure 3 displays a simplified functional diagram of the

input T/H circuit. This input T/H circuit allows for high

analog input frequencies of 175MHz and beyond and

supports a common-mode input voltage of VDD/2 ±0.5V.

The MAX12553 sampling clock controls the ADC’s

switched-capacitor T/H architecture (Figure 3) allowing

the analog input signal to be stored as charge on the

sampling capacitors. These switches are closed (track)

when the sampling clock is high and open (hold) when

the sampling clock is low (Figure 4). The analog input

signal source must be capable of providing the dynam-

ic current necessary to charge and discharge the sam-

pling capacitors. To avoid signal degradation, these

capacitors must be charged to one-half LSB accuracy

within one-half of a clock cycle.

The analog input of the MAX12553 supports differential

or single-ended input drive. For optimum performance

with differential inputs, balance the input impedance of

INP and INN and set the common-mode voltage to mid-

supply (VDD/2). The MAX12553 provides the optimum

common-mode voltage of VDD/2 through the COM out-

put when operating in internal reference mode and

buffered external reference mode. This COM output

voltage can be used to bias the input network as shown

in Figures 10, 11, and 12.

Reference Output (REFOUT)

An internal bandgap reference is the basis for all the

internal voltages and bias currents used in the

MAX12553. The power-down logic input (PD) enables

and disables the reference circuit. The reference circuit

requires 10ms to power up and settle when power is

applied to the MAX12553 or when PD transitions from

high to low. REFOUT has approximately 17kΩto GND

when the MAX12553 is in power-down.

The internal bandgap reference and its buffer generate

VREFOUT to be 2.048V. The reference temperature coeffi-

cient is typically +50ppm/°C. Connect an external ≥0.1µF

bypass capacitor from REFOUT to GND for stability.

14-Bit, 65Msps, 3.3V ADC

14 ______________________________________________________________________________________

MAX12553

INP

INN

14-BIT

PIPELINE

ADC DEC

REFERENCE

SYSTEM

COM

REFOUT

REFN

REFP

OVDD

DAV

OUTPUT

DRIVERS

D13–D0

DOR

REFIN

T/H

POWER CONTROL

AND

BIAS CIRCUITS

CLKP CLOCK

GENERATOR

AND

DUTY-CYCLE

EQUALIZER

CLKN

CLKTYP

VDD

GND

DCE

G/T

Figure 2. Simplified Functional Diagram

MAX12553

CPAR

2pF

VDD

BOND WIRE

INDUCTANCE

1.5nH

INP

SAMPLING

CLOCK

*THE EFFECTIVE RESISTANCE OF THE

SWITCHED SAMPLING CAPACITORS IS:

*CSAMPLE

4.5pF

CPAR

2pF

VDD

BOND WIRE

INDUCTANCE

1.5nH

INN *CSAMPLE

4.5pF

RSAMPLE = 1

fCLK x CSAMPLE

Figure 3. Simplified Input T/H Circuit

REFOUT sources up to 1.0mA and sinks up to 0.1mA

for external circuits with a load regulation of 35mV/mA.

Short-circuit protection limits IREFOUT to a 2.1mA

source current when shorted to GND and a 0.24mA

sink current when shorted to VDD.

Analog Inputs and Reference

Configurations

The MAX12553 full-scale analog input range is

adjustable from ±0.35V to ±1.10V with a common-

mode input range of VDD/2 ±0.5V. The MAX12553 pro-

vides three modes of reference operation. The voltage

at REFIN (VREFIN) sets the reference operation mode

(Table 1).

To operate the MAX12553 with the internal reference,

connect REFOUT to REFIN either with a direct short or

through a resistive divider. In this mode, COM, REFP,

and REFN are low-impedance outputs with VCOM =

VDD/2, VREFP = VDD/2 + VREFIN x 3/8, and VREFN =

VDD/2 - VREFIN x 3/8. The REFIN input impedance is

very large (>50MΩ). When driving REFIN through a

resistive divider, use resistances ≥10kΩto avoid load-

ing REFOUT.

Buffered external reference mode is virtually identical to

internal reference mode except that the reference

source is derived from an external reference and not

the MAX12553 REFOUT. In buffered external reference

mode, apply a stable 0.7V to 2.2V source at REFIN. In

this mode, COM, REFP, and REFN are low-impedance

outputs with VCOM = VDD/2, VREFP = VDD/2 + VREFIN x

3/8, and VREFN = VDD/2 - VREFIN x 3/8.

To operate the MAX12553 in unbuffered external refer-

ence mode, connect REFIN to GND. Connecting REFIN

to GND deactivates the on-chip reference buffers for

COM, REFP, and REFN. With the respective buffers

deactivated, COM, REFP, and REFN become high-

impedance inputs and must be driven through sepa-

rate, external reference sources. Drive VCOM to VDD/2

±5%, and drive REFP and REFN such that VCOM =

(VREFP + VREFN)/2. The full-scale analog input range is

±(VREFP - VREFN) x 2/3.

MAX12553

14-Bit, 65Msps, 3.3V ADC

______________________________________________________________________________________ 15

tAD

T/H

CLKN

CLKP

tAJ

TRACK HOLDTRACK HOLDTRACK HOLDTRACKHOLD

ANALOG

INPUT

SAMPLED

DATA

Figure 4. T/H Aperture Timing

VREFIN REFERENCE MODE

35% VREFOUT to 100%

VREFOUT

Internal Reference Mode. Drive REFIN with REFOUT either through a direct short or a resistive divider.

The full-scale analog input range is ±VREFIN/2:

VCOM = VDD/2

VREFP = VDD/2 + VREFIN x 3/8

VREFN = VDD/2 - VREFIN x 3/8

0.7V to 2.2V

Buffered External Reference Mode. Apply an external 0.7V to 2.2V reference voltage to REFIN.

The full-scale analog input range is ±VREFIN/2:

VCOM = VDD/2

VREFP = VDD/2 + VREFIN x 3/8

VREFN = VDD/2 - VREFIN x 3/8

<0.4V Unbuffered External Reference Mode. Drive REFP, REFN, and COM with external reference sources.

The full-scale analog input range is ±(VREFP - VREFN) x 2/3.

Table 1. Reference Modes

MAX12553

All three modes of reference operation require the

same bypass capacitor combinations. Bypass COM

with a 2.2µF capacitor to GND. Bypass REFP and

REFN each with a 0.1µF capacitor to GND. Bypass

REFP to REFN with a 1µF capacitor in parallel with a

10µF capacitor. Place the 1µF capacitor as close to

the device as possible on the same side of the PC

board. Bypass REFIN and REFOUT to GND with a

0.1µF capacitor.

For detailed circuit suggestions, see Figure 13 and

Figure 14.

Clock Input and Clock Control Lines

(CLKP, CLKN, CLKTYP)

The MAX12553 accepts both differential and single-

ended clock inputs. For single-ended clock input oper-

ation, connect CLKTYP to GND, CLKN to GND, and

drive CLKP with the external single-ended clock signal.

For differential clock input operation, connect CLKTYP

to OVDD or VDD, and drive CLKP and CLKN with the

external differential clock signal. To reduce clock jitter,

the external single-ended clock must have sharp falling

edges. Consider the clock input as an analog input and

route it away from any other analog inputs and digital

signal lines.

CLKP and CLKN are high impedance when the

MAX12553 is powered down (Figure 5).

Low clock jitter is required for the specified SNR perfor-

mance of the MAX12553. Analog input sampling

occurs on the falling edge of the clock signal, requiring

this edge to have the lowest possible jitter. Jitter limits

the maximum SNR performance of any ADC according

to the following relationship:

where fIN represents the analog input frequency and tJ

is the total system clock jitter. Clock jitter is especially

critical for undersampling applications. For example,

assuming that clock jitter is the only noise source, to

obtain the specified 71dB of SNR with an input frequen-

cy of 175MHz, the system must have less than 0.25ps

of clock jitter. In actuality, there are other noise sources

such as thermal noise and quantization noise that con-

tribute to the system noise, requiring the clock jitter to

be less than 0.2ps to obtain the specified 71dB of SNR

at 175MHz.

Clock Duty-Cycle Equalizer (DCE)

The clock duty-cycle equalizer uses a delay-locked

loop (DLL) to create internal timing signals that are

duty-cycle independent. Due to this DLL, the

MAX12553 requires approximately 100 clock cycles to

acquire and lock to new clock frequencies.

Disabling the clock duty-cycle equalizer reduces the

analog supply current by 1.5mA.

SNR ft

IN J

log

=×

×π×













20 1

14-Bit, 65Msps, 3.3V ADC

16 ______________________________________________________________________________________

MAX12553

CLKP

CLKN

VDD

GND

10kΩ

DUTY-CYCLE

EQUALIZER

SWITCHES S1_ AND S2_ ARE OPEN

DURING POWER-DOWN, MAKING

CLKP AND CLKN HIGH IMPEDANCE.

SWITCHES S2_ ARE OPEN IN

SINGLE-ENDED CLOCK MODE.

S1H

S2H

S1L

S2L

Figure 5. Simplified Clock-Input Circuit

System-Timing Requirements

Figure 6 shows the relationship between the clock, ana-

log inputs, DAV indicator, DOR indicator, and the result-

ing output data. The analog input is sampled on the

falling edge of the clock signal and the resulting data

appears at the digital outputs 8.5 clock cycles later.

The DAV indicator is synchronized with the digital out-

put and optimized for use in latching data into digital

back-end circuitry. Alternatively, digital back-end cir-

cuitry can be latched with the falling edge of the con-

version clock (CLKP-CLKN).

Data-Valid Output (DAV)

DAV is a single-ended version of the input clock (CLKP).

Output data changes on the falling edge of DAV, and

DAV rises once output data is valid (Figure 6).

The state of the duty-cycle equalizer input (DCE)

changes the waveform at DAV. With the duty-cycle

equalizer disabled (DCE = low), the DAV signal is the

inverse of the signal at CLKP delayed by 6.8ns (tDAV).

With the duty-cycle equalizer enabled (DCE = high), the

DAV signal has a fixed pulse width that is independent of

CLKP. In either case, with DCE high or low, output data

at D13–D0 and DOR are valid from 8.5ns before the ris-

ing edge of DAV to 6.3ns after the rising edge of DAV,

and the rising edge of DAV is synchronized to have a

6.9ns (tDAV) delay from the falling edge of CLKP.

DAV is high impedance when the MAX12553 is in

power-down (PD = high). DAV is capable of sinking

and sourcing 600µA and has three times the drive

strength of D13–D0 and DOR. DAV is typically used to

latch the MAX12553 output data into an external back-

end digital circuit.

Keep the capacitive load on DAV as low as possible

(<25pF) to avoid large digital currents feeding back

into the analog portion of the MAX12553 and degrading

its dynamic performance. An external buffer on DAV

isolates it from heavy capacitive loads. Refer to the

MAX12555 evaluation kit schematic for an example of

DAV driving back-end digital circuitry through an exter-

nal buffer.

MAX12553

14-Bit, 65Msps, 3.3V ADC

______________________________________________________________________________________ 17

DAV

D13–D0

NN+1 N+2

N+3

N+4 N+5

N+6

N+7

N+8

N+9

tDAV

tSETUP

tAD

N-1

N-2

N-3

tHOLD

tCL tCH

DOR

8.5 CLOCK-CYCLE DATA LATENCY

DIFFERENTIAL ANALOG INPUT (INP–INN)

tSETUP tHOLD

N N+1 N+2 N+3 N+5 N+6 N+7N-1N-2N-3 N+9N+4 N+8

CLKN

CLKP

(VREFP - VREFN) x 2/3

(VREFN - VREFP) x 2/3

Figure 6. System-Timing Diagram

MAX12553

Data Out-of-Range Indicator (DOR)

The DOR digital output indicates when the analog input

voltage is out of range. When DOR is high, the analog

input is out of range. When DOR is low, the analog

input is within range. The valid differential input range is

from (VREFP - VREFN) x 3/4 to (VREFN - VREFP) x 3/4.

Signals outside this valid differential range cause DOR

to assert high as shown in Table 2 and Figure 6.

DOR is synchronized with DAV and transitions along

with the output data D13–D0. There is an 8.5 clock-

cycle latency in the DOR function as is with the output

data (Figure 6).

DOR is high impedance when the MAX12553 is in

power-down (PD = high). DOR enters a high-imped-

ance state within 10ns after the rising edge of PD and

becomes active 10ns after PD’s falling edge.

Digital Output Data (D13–D0), Output Format (G/T)

The MAX12553 provides a 14-bit, parallel, tri-state out-

put bus. D13–D0 and DOR update on the falling edge

of DAV and are valid on the rising edge of DAV.

The MAX12553 output data format is either Gray code

or two’s complement, depending on the logic input G/T.

With G/Thigh, the output data format is Gray code.

With G/Tlow, the output data format is two’s comple-

ment. See Figure 8 for a binary-to-Gray and Gray-to-

binary code-conversion example.

The following equations, Table 2, Figure 7, and Figure 8

define the relationship between the digital output and

the analog input:

for Gray code (G/T= 1)

for two’s complement (G/T= 0)

where CODE10 is the decimal equivalent of the digital

output code as shown in Table 2.

Digital outputs D13–D0 are high impedance when the

MAX12553 is in power-down (PD = high). D13–D0 tran-

sition high 10ns after the rising edge of PD and

become active 10ns after PD’s falling edge.

Keep the capacitive load on the MAX12553 digital out-

puts D13–D0 as low as possible (<15pF) to avoid large

digital currents feeding back into the analog portion of

the MAX12553 and degrading its dynamic perfor-

mance. The addition of external digital buffers on the

digital outputs isolates the MAX12553 from heavy

capacitive loading. To improve the dynamic perfor-

mance of the MAX12553, add 220Ωresistors in series

with the digital outputs close to the MAX12553. Refer to

the MAX12555 evaluation kit schematic for an example

of the digital outputs driving a digital buffer through

220Ωseries resistors.

Power-Down Input (PD)

The MAX12553 has two power modes that are con-

trolled with the power-down digital input (PD). With PD

VV V V CODE

INP INN REFP REFN

−−=××()

3 16384

VV V V CODE

INP INN REFP REFN

−− −

=×× () 4

8192

16384

14-Bit, 65Msps, 3.3V ADC

18 ______________________________________________________________________________________

MAX12553

14-Bit, 65Msps, 3.3V ADC

______________________________________________________________________________________ 19

GRAY CODE

OUTPUT CODE

(G/T = 1)

TWO’S-COMPLEMENT

OUTPUT CODE

(G/T = 0)

BINARY

D13 ➝ D0 DOR

HEXADECIMAL

EQUIVALENT

D13 ➝ D0

DECIMAL

EQUIVALENT

D13 ➝ D0

(CODE10)

BINARY

D13 ➝ D0 DOR

HEXADECIMAL

EQUIVALENT

D13➝ D0

DECIMAL

EQUIVALENT

D13 ➝ D0

(CODE10)

VINP - VINN

VREFP = 2.418V

VREFN = 0.882V

10 0000 0000 0000 10x2000 +16383 01 1111 1111 1111 10x1FFF +8191

>+1.023875V

(DATA OUT OF

RANGE)

10 0000 0000 0000 0 0x2000 +16383 01 1111 1111 1111 0 0x1FFF +8191 +1.023875V

10 0000 0000 0001 0 0x2001 +16382 01 1111 1111 1110 0 0x1FFE +8190 +1.023750V

11 0000 0000 0011 0 0x3003 +8194 00 0000 0000 0010 0 0x0002 +2 +0.000250V

11 0000 0000 0001 0 0x3001 +8193 00 0000 0000 0001 0 0x0001 +1 +0.000125V

11 0000 0000 0000 0 0x3000 +8192 00 0000 0000 0000 0 0x0000 0 +0.000000V

01 0000 0000 0000 0 0x1000 +8191 11 1111 1111 1111 0 0x3FFF -1 -0.000125V

01 0000 0000 0001 0 0x1001 +8190 11 1111 1111 1110 0 0x3FFE -2 -0.000250V

00 0000 0000 0001 0 0x0001 +1 10 0000 0000 0001 0 0x2001 -8191 -1.023875V

00 0000 0000 0000 0 0x0000 0 10 0000 0000 0000 0 0x2000 -8192 -1.024000V

00 0000 0000 0000 10x0000 0 10 0000 0000 0000 10x2000 -8192

<-1.024000V

(DATA OUT OF

RANGE)

()

Table 2. Output Codes vs. Input Voltage

MAX12553

14-Bit, 65Msps, 3.3V ADC

20 ______________________________________________________________________________________

low, the MAX12553 is in normal operating mode. With

PD high, the MAX12553 is in power-down mode.

The power-down mode allows the MAX12553 to effi-

ciently use power by transitioning to a low-power state

when conversions are not required. Additionally, the

MAX12553 parallel output bus is high impedance in

power-down mode, allowing other devices on the bus

to be accessed.

In power-down mode, all internal circuits are off, the

analog supply current reduces to 0.045mA, and the

digital supply current reduces to 0.02mA. The following

list shows the state of the analog inputs and digital out-

puts in power-down mode:

•INP, INN analog inputs are disconnected from the

internal input amplifier (Figure 3).

•REFOUT has approximately 17kΩto GND.

•REFP, COM, REFN go high impedance with respect

to VDD and GND, but there is an internal 4kΩresistor

between REFP and COM, as well as an internal 4kΩ

resistor between REFN and COM.

•D13–D0, DOR, and DAV go high impedance.

•CLKP, CLKN go high impedance (Figure 5).

The wake-up time from power-down mode is dominat-

ed by the time required to charge the capacitors at

REFP, REFN, and COM. In internal reference mode and

buffered external reference mode, the wake-up time is

typically 10ms with the recommended capacitor array

(Figure 13). When operating in unbuffered external ref-

erence mode, the wake-up time is dependent on the

external reference drivers.

Applications Information

Using Transformer Coupling

In general, the MAX12553 provides better SFDR and THD

performance with fully differential input signals as

opposed to single-ended input drive. In differential input

mode, even-order harmonics are lower as both inputs are

balanced, and each of the ADC inputs only requires half

the signal swing compared to single-ended input mode.

An RF transformer (Figure 10) provides an excellent

solution to convert a single-ended input source signal

to a fully differential signal, required by the MAX12553

for optimum performance. Connecting the center tap of

the transformer to COM provides a VDD/2 DC level shift

to the input. Although a 1:1 transformer is shown, a

step-up transformer can be selected to reduce the

drive requirements. A reduced signal swing from the

input driver, such as an op amp, can also improve the

overall distortion. The configuration of Figure 10 is good

for frequencies up to Nyquist (fCLK/2).

The circuit of Figure 11 converts a single-ended input

signal to fully differential just as Figure 10. However,

Figure 11 utilizes an additional transformer to improve

DIFFERENTIAL INPUT VOLTAGE (LSB)

TWO'S COMPLEMENT OUTPUT CODE (LSB)

-8189 +8191+8189-1 0 +1-8191

0x2000

0x2001

0x2002

0x2003

0x1FFF

0x1FFE

0x1FFD

0x3FFF

0x0000

0x0001

(VREFP - VREFN) x 2/3 (VREFP - VREFN) x 2/3

1 LSB = VREFP - VREFN

16384

Figure 7. Two’s Complement Transfer Function (G/

= 0)

DIFFERENTIAL INPUT VOLTAGE (LSB)

GRAY OUTPUT CODE (LSB)

+1 +8191+8189-1 0

-8191 -8189

0x0000

0x0001

0x0003

0x0002

0x2000

0x2001

0x2003

0x1000

0x3000

0x3001

(VREFP - VREFN) x 2/3 (VREFP - VREFN) x 2/3

1 LSB = VREFP - VREFN

16384

Figure 8. Gray Code Transfer Function (G/

= 1)

MAX12553

14-Bit, 65Msps, 3.3V ADC

______________________________________________________________________________________ 21

BINARY-TO-GRAY CODE CONVERSION

1) THE MOST SIGNIFICANT GRAY CODE BIT IS THE SAME

AS THE MOST SIGNIFICANT BINARY BIT.

2) SUBSEQUENT GRAY CODE BITS ARE FOUND ACCORDING

TO THE FOLLOWING EQUATION:

BINARY

GRAY CODE

BIT POSITION

3) REPEAT STEP 2 UNTIL COMPLETE.

BINARY

GRAY CODE

BIT POSITION

4) THE FINAL GRAY CODE CONVERSION IS:

BINARY

BIT POSITION

GRAY-TO-BINARY CODE CONVERSION

WHERE IS THE EXCLUSIVE OR FUNCTION (SEE TRUTH

TABLE BELOW) AND X IS THE BIT POSITION.

011 101001100

D11 D7 D3 D0

D13

01 1 1 0100 1100

D11 D7 D3 D0

D13

01 1 1 0100 1100

D11 D7 D3 D0

D13

1011 0100 1100 BINARY

GRAY CODE0

D11 D7 D3 D0 BIT POSITION

D13

GRAY CODE0 101101110 1010

GRAYX = BINARYX BINARYX+1

GRAY12 = BINARY12 BINARY13

GRAY12 = 1 0

GRAY12 = 1

GRAY11 = BINARY11 BINARY12

GRAY11 = 1 1

GRAY11 = 0

1) THE MOST SIGNIFICANT BINARY BIT IS THE SAME

AS THE MOST SIGNIFICANT GRAY CODE BIT.

2) SUBSEQUENT BINARY BITS ARE FOUND ACCORDING

TO THE FOLLOWING EQUATION:

GRAY CODE

BINARY

BIT POSITION

3) REPEAT STEP 2 UNTIL COMPLETE.

GRAY CODE

BINARY

BIT POSITION

4) THE FINAL GRAY CODE CONVERSION IS:

GRAY CODE

BIT POSITION

WHERE IS THE EXCLUSIVE OR FUNCTION (SEE TRUTH

TABLE BELOW) AND X IS THE BIT POSITION.

01001110 1010

D11 D7 D3 D0

D13

01 0 1110 1010

D11 D7 D3 D0

D13

01 0 0 1110 1010

D11 D7 D3 D0

D13

0110 1110 1010 GRAY CODE

BINARY0

D11 D7 D3 D0 BIT POSITION

D13

BINARY0 111010100 1100

BINARYX = BINARYX+1 GRAYX

BINARY12 = BINARY13 GRAY12

BINARY12 = 0 1

BINARY12 = 1

BINARY11 = BINARY12 GRAY11

BINARY11 = 1 0

BINARY11 = 1

EXCULSIVE OR TRUTH TABLE

Y = A B

Figure 9. Binary-to-Gray and Gray-to-Binary Code Conversion

MAX12553

14-Bit, 65Msps, 3.3V ADC

22 ______________________________________________________________________________________

the common-mode rejection, allowing high-frequency

signals beyond the Nyquist frequency. The two sets of

termination resistors provide an equivalent 75Ωtermi-

nation to the signal source. The second set of termina-

tion resistors connects to COM, providing the correct

input common-mode voltage. Two 0Ωresistors in series

with the analog inputs allow high IF input frequencies.

These 0Ωresistors can be replaced with low-value

resistors to limit the input bandwidth.

Single-Ended, AC-Coupled Input Signal

Figure 12 shows an AC-coupled, single-ended input

application. The MAX4108 provides high speed, high

bandwidth, low noise, and low distortion to maintain the

input signal integrity.

MAX12553

N.C.

VIN

0.1µF

MINICIRCUITS

TT1-6 OR T1-1T

24.9Ω

12pF

2.2µF

INP

COM

INN

Figure 10. Transformer-Coupled Input Drive for Input

Frequencies Up to Nyquist

MAX12553

N.C. N.C.

MINICIRCUITS

ADT1-1WT

N.C.

VIN

0.1µF

MINICIRCUITS

ADT1-1WT

0Ω*

5.6pF

2.2µF

INP

COM

INN

110Ω

0.1%

110Ω

0.1%

75Ω

0.5%

75Ω

0.5%

*0Ω RESISTORS CAN BE REPLACED WITH LOW-VALUE

RESISTORS TO LIMIT THE BANDWIDTH.

Figure 11. Transformer-Coupled Input Drive for Input Frequencies Beyond Nyquist

MAX12553

5.6pF

2.2µF

INP

COM

INN

24.9Ω

100Ω

0.1µF

MAX4108

VIN

Figure 12. Single-Ended, AC-Coupled Input Drive

MAX12553

14-Bit, 65Msps, 3.3V ADC

______________________________________________________________________________________ 23

Buffered External Reference

Drives Multiple ADCs

The buffered external reference mode allows for more

control over the MAX12553 reference voltage and

allows multiple converters to use a common reference.

The REFIN input impedance is >50MΩ.

Figure 13 uses the MAX6029EUK21 precision 2.048V

reference as a common reference for multiple convert-

ers. The 2.048V output of the MAX6029 passes through

a one-pole 10Hz lowpass filter to the MAX4230. The

MAX4230 buffers the 2.048V reference and provides

additional 10Hz lowpass filtering before its output is

applied to the REFIN input of the MAX12553.

MAX12553

NOTE: ONE FRONT-END REFERENCE

CIRCUIT IS CAPABLE OF SOURCING 15mA

AND SINKING 30mA OF OUTPUT CURRENT.

*PLACE THE 1µF REFP-to-REFN BYPASS CAPACITOR AS CLOSE TO THE DEVICE AS POSSIBLE.

16.2kΩ

0.1µF

1µF

2.048V

+3.3V

547Ω

1.47kΩ

+3.3V

10µF

330µF

+3.3V

2.2µF

0.1µF

1µF* 10µF

0.1µF

REFP

REFN

COM

VDD

GND

REFIN

REFOUT

MAX12553

+3.3V

2.2µF

0.1µF

1µF* 10µF

0.1µF

REFP

REFN

COM

VDD

GND

REFIN

REFOUT

MAX6029EUK21

MAX4230

Figure 13. External Buffered Reference Driving Multiple ADCs

MAX12553

14-Bit, 65Msps, 3.3V ADC

24 ______________________________________________________________________________________

Unbuffered External

Reference Drives Multiple ADCs

The unbuffered external reference mode allows for pre-

cise control over the MAX12553 reference and allows

multiple converters to use a common reference.

Connecting REFIN to GND disables the internal refer-

ence, allowing REFP, REFN, and COM to be driven

directly by a set of external reference sources.

Figure 14 uses the MAX6029EUK30 precision 3.000V

reference as a common reference for multiple convert-

ers. A seven-component resistive divider chain follows

the MAX6029 voltage reference. The 0.47µF capacitor

along this chain creates a 10Hz lowpass filter. Three

MAX4230 operational amplifiers buffer taps along this

resistor chain providing 2.413V, 1.647V, and 0.880V to

the MAX12553’s REFP, COM, REFN reference inputs,

MAX12553

*PLACE THE 1µF REFP-TO-REFN BYPASS CAPACITOR AS CLOSE TO THE DEVICE AS POSSIBLE.

0.1µF

2.413V

+3.3V

547Ω

1.47kΩ

+3.3V

10µF

330µF

+3.3V

2.2µF

0.1µF

1µF*

10µF

0.1µF

REFOUT

REFN

REFIN 39

VDD

GND

COM

REFP 38

MAX6029EUK30

MAX4230

0.1µF

0.47µF

1.647V

547Ω

1.47kΩ

+3.3V

10µF

330µF

MAX4230

0.1µF

0.880V

547Ω

1.47kΩ

+3.3V

10µF

330µF

MAX4230

MAX12553

+3.3V

2.2µF

0.1µF

1µF*

10µF

0.1µF

REFOUT

REFN

REFIN 39

VDD

GND

COM

REFP 38

3.000V

20kΩ

52.3kΩ

20kΩ

0.1µF

2.2µF

Figure 14. External Unbuffered Reference Driving Multiple ADCs

MAX12553

14-Bit, 65Msps, 3.3V ADC

______________________________________________________________________________________ 25

respectively. The feedback around the MAX4230 op

amps provides additional 10Hz lowpass filtering. The

2.413V and 0.880V reference voltages set the full-scale

analog input range to ±1.022V = ±(VREFP - VREFN) x 2/3.

A common power source for all active components

removes any concern regarding power-supply

sequencing when powering up or down.

Grounding, Bypassing, and

Board Layout

The MAX12553 requires high-speed board layout

design techniques. Refer to the MAX12555 evaluation

kit. data sheet for a board layout reference. Locate all

bypass capacitors as close to the device as possible,

preferably on the same side of the board as the ADC,

using surface-mount devices for minimum inductance.

Bypass VDD to GND with a 0.1µF ceramic capacitor in

parallel with a 2.2µF ceramic capacitor. Bypass OVDD

to GND with a 0.1µF ceramic capacitor in parallel with a

2.2µF ceramic capacitor.

Multilayer boards with ample ground and power planes

produce the highest level of signal integrity. All

MAX12553 GNDs and the exposed backside paddle

must be connected to the same ground plane. The

MAX12553 relies on the exposed backside paddle con-

nection for a low-inductance ground connection. Use

multiple vias to connect the top-side ground to the bot-

tom-side ground. Isolate the ground plane from any

noisy digital system ground planes such as a DSP or

output buffer ground.

Route high-speed digital signal traces away from the

sensitive analog traces. Keep all signal lines short and

free of 90°turns.

Ensure that the differential analog input network layout

is symmetric and that all parasitics are balanced equal-

ly. Refer to the MAX12555 evaluation kit data sheet for

an example of symmetric input layout.

Parameter Definitions

Integral Nonlinearity (INL)

Integral nonlinearity is the deviation of the values on an

actual transfer function from a straight line. For the

MAX12553, this straight line is between the end points

of the transfer function, once offset and gain errors have

been nullified. INL deviations are measured at every

step of the transfer function and the worst-case devia-

tion is reported in the Electrical Characteristics table.

Differential Nonlinearity (DNL)

Differential nonlinearity is the difference between an

actual step width and the ideal value of 1 LSB. A DNL

error specification of less than 1 LSB guarantees no

missing codes and a monotonic transfer function. For

the MAX12553, DNL deviations are measured at every

step of the transfer function and the worst-case devia-

tion is reported in the Electrical Characteristics table.

Offset Error

Offset error is a figure of merit that indicates how well

the actual transfer function matches the ideal transfer

function at a single point. Ideally the midscale

MAX12553 transition occurs at 0.5 LSB above mid-

scale. The offset error is the amount of deviation

between the measured midscale transition point and

the ideal midscale transition point.

Gain Error

Gain error is a figure of merit that indicates how well the

slope of the actual transfer function matches the slope

of the ideal transfer function. The slope of the actual

transfer function is measured between two data points:

positive full scale and negative full scale. Ideally, the

positive full-scale MAX12553 transition occurs at 1.5

LSBs below positive full scale, and the negative full-

scale transition occurs at 0.5 LSB above negative full

scale. The gain error is the difference of the measured

transition points minus the difference of the ideal transi-

tion points.

Small-Signal Noise Floor (SSNF)

Small-signal noise floor is the integrated noise and dis-

tortion power in the Nyquist band for small-signal

inputs. The DC offset is excluded from this noise calcu-

lation. For this converter, a small signal is defined as a

single tone with an amplitude less than -35dBFS. This

parameter captures the thermal and quantization noise

characteristics of the converter and can be used to

help calculate the overall noise figure of a receive

channel. Go to www.maxim-ic.com for application

notes on thermal + quantization noise floor.

Signal-to-Noise Ratio (SNR)

For a waveform perfectly reconstructed from digital

samples, the theoretical maximum SNR is the ratio of

the full-scale analog input (RMS value) to the RMS

quantization error (residual error). The ideal, theoretical

minimum analog-to-digital noise is caused by quantiza-

tion error only and results directly from the ADC’s reso-

lution (N bits):

SNR[max] = 6.02 ×N + 1.76

In reality, there are other noise sources besides quanti-

zation noise: thermal noise, reference noise, clock jitter,

etc. SNR is computed by taking the ratio of the RMS

signal to the RMS noise. RMS noise includes all spec-

tral components to the Nyquist frequency excluding the

MAX12553

14-Bit, 65Msps, 3.3V ADC

26 ______________________________________________________________________________________

fundamental, the first six harmonics (HD2–HD7), and

the DC offset.

Signal-to-Noise Plus Distortion (SINAD)

SINAD is computed by taking the ratio of the RMS signal

to the RMS noise plus distortion. RMS noise plus distor-

tion includes all spectral components to the Nyquist fre-

quency excluding the fundamental and the DC offset.

Effective Number of Bits (ENOB)

ENOB specifies the dynamic performance of an ADC at

a specific input frequency and sampling rate. An ideal

ADC’s error consists of quantization noise only. ENOB for

a full-scale sinusoidal input waveform is computed from:

Single-Tone Spurious-Free Dynamic Range

(SFDR)

SFDR is the ratio expressed in decibels of the RMS

amplitude of the fundamental (maximum signal compo-

nent) to the RMS amplitude of the next-largest spurious

component, excluding DC offset.

Total Harmonic Distortion (THD)

THD is the ratio of the RMS sum of the first six harmon-

ics of the input signal to the fundamental itself. This is

expressed as:

where V1is the fundamental amplitude, and V2through

V7are the amplitudes of the 2nd- through 7th-order

harmonics (HD2–HD7).

Intermodulation Distortion (IMD)

IMD is the ratio of the RMS sum of the intermodulation

products to the RMS sum of the two fundamental input

tones. This is expressed as:

The fundamental input tone amplitudes (V1and V2) are

at -7dBFS. Fourteen intermodulation products (VIM_)

are used in the MAX12553 IMD calculation. The inter-

modulation products are the amplitudes of the output

spectrum at the following frequencies, where fIN1 and

fIN2 are the fundamental input tone frequencies:

•Second-order intermodulation products:

fIN1 + fIN2, fIN2 - fIN1

•Third-order intermodulation products:

2 x fIN1 - fIN2, 2 x fIN2 - fIN1, 2 x fIN1 + fIN2, 2 x fIN2 + fIN1

•Fourth-order intermodulation products:

3 x fIN1 - fIN2, 3 x fIN2 - fIN1, 3 x fIN1 + fIN2, 3 x fIN2 + fIN1

•Fifth-order intermodulation products:

3 x fIN1 - 2 x fIN2, 3 x fIN2 - 2 x fIN1, 3 x fIN1 + 2 x fIN2,

3 x fIN2 + 2 x fIN1

Third-Order Intermodulation (IM3)

IM3 is the total power of the third-order intermodulation

products to the Nyquist frequency relative to the total

input power of the two input tones fIN1 and fIN2. The

individual input tone levels are at -7dBFS. The third-

order intermodulation products are 2 x fIN1 - fIN2, 2 x

fIN2 - fIN1, 2 x fIN1 + fIN2, 2 x fIN2 + fIN1.

Two-Tone Spurious-Free Dynamic Range

(SFDRTT)

SFDRTT represents the ratio, expressed in decibels, of

the RMS amplitude of either input tone to the RMS

amplitude of the next-largest spurious component in the

spectrum, excluding DC offset. This spurious compo-

nent can occur anywhere in the spectrum up to Nyquist

and is usually an intermodulation product or a harmonic.

Aperture Delay

The MAX12553 samples data on the falling edge of its

sampling clock. In actuality, there is a small delay

between the falling edge of the sampling clock and the

IMD VV V V

IM IM IM IM

log .......

=× +++ +













20 1222132142

1222

THD VVVVVV

log

=× +++++













20 223242526272

ENOB SINAD

=









−176

602

MAX12553

14-Bit, 65Msps, 3.3V ADC

______________________________________________________________________________________ 27

actual sampling instant. Aperture delay (tAD) is the time

defined between the falling edge of the sampling clock

and the instant when an actual sample is taken (Figure 4).

Aperture Jitter

Figure 4 depicts the aperture jitter (tAJ), which is the

sample-to-sample variation in the aperture delay.

Output Noise (nOUT)

The output noise (nOUT) parameter is similar to the ther-

mal + quantization noise parameter and is an indication

of the ADC’s overall noise performance.

No fundamental input tone is used to test for nOUT; INP,

INN, and COM are connected together and 1024k data

points collected. nOUT is computed by taking the RMS

value of the collected data points.

Overdrive Recovery Time

Overdrive recovery time is the time required for the

ADC to recover from an input transient that exceeds the

full-scale limits. The MAX12553 specifies overdrive

recovery time using an input transient that exceeds the

full-scale limits by ±10%.

REFP 1

REFN 2

COM 3

GND 4

INP 5

INN 6

GND 7

DCE 8

CLKN 9

CLKP 10

D230

D329

D428

D527

D626

D725

D824

D923

D1022

D1121

REFIN

REFOUT

VDD

GND

OVDD

DAV

CLKTYP

VDD

GND

OVDD

DOR

D13

D12

G/T

TOP VIEW

MAX12553

EXPOSED PADDLE (GND)

THIN QFN

6mm x 6mm x 0.8mm

Pin Configuration

MAX12553

14-Bit, 65Msps, 3.3V ADC

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are

implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

28 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600

Package Information

(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,

go to www.maxim-ic.com/packages.)

QFN THIN 6x6x0.8.EPS

e e

A1 A2 A

E/2

D/2

E2/2

(NE-1) X e

(ND-1) X e

D2/2

21-0141

PACKAGE OUTLINE

36, 40, 48L THIN QFN, 6x6x0.8mm

8. COPLANARITY APPLIES TO THE EXPOSED HEAT SINK SLUG AS WELL AS THE TERMINALS.

6. ND AND NE REFER TO THE NUMBER OF TERMINALS ON EACH D AND E SIDE RESPECTIVELY.

5. DIMENSION b APPLIES TO METALLIZED TERMINAL AND IS MEASURED BETWEEN 0.25 mm AND 0.30 mm

FROM TERMINAL TIP.

4. THE TERMINAL #1 IDENTIFIER AND TERMINAL NUMBERING CONVENTION SHALL CONFORM TO JESD 95-1

SPP-012. DETAILS OF TERMINAL #1 IDENTIFIER ARE OPTIONAL, BUT MUST BE LOCATED WITHIN THE

ZONE INDICATED. THE TERMINAL #1 IDENTIFIER MAY BE EITHER A MOLD OR MARKED FEATURE.

9. DRAWING CONFORMS TO JEDEC MO220, EXCEPT FOR 0.4mm LEAD PITCH PACKAGE T4866-1.

7. DEPOPULATION IS POSSIBLE IN A SYMMETRICAL FASHION.

3. N IS THE TOTAL NUMBER OF TERMINALS.

2. ALL DIMENSIONS ARE IN MILLIMETERS. ANGLES ARE IN DEGREES.

1. DIMENSIONING & TOLERANCING CONFORM TO ASME Y14.5M-1994.

NOTES:

10. WARPAGE SHALL NOT EXCEED 0.10 mm.

21-0141

PACKAGE OUTLINE

36, 40, 48L THIN QFN, 6x6x0.8mm