MIC2582-MYM-TR - MICROCHIP TECHNOLOGY

MIC2582/MIC2583

Single Channel Hot Swap Controllers

Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com

April 2009 M9999-043009-C

General Description

The MIC2582 and MIC2583 are single channel positive

voltage hot swap controllers designed to allow the safe

insertion of boards into live system backplanes. The

MIC2582 and MIC2583 are available in 8-pin SOIC and

16-pin QSOP packages, respectively. Using a few external

components and by controlling the gate drive of an

external N-Channel MOSFET device, the MIC2582/83

provide inrush current limiting and output voltage slew rate

control in harsh, critical power supply environments.

Additionally, a circuit breaker function will latch the output

MOSFET off if the current limit threshold is exceeded for a

determined period. The MIC2583R option includes an

auto-restart function upon detecting an over current

condition.

Datasheets and support documentation can be found on

Micrel’s web site at www.micrel.com.

Features

• MIC2582:

Pin-for-pin functional equivalent to the LTC1422

• 2.3V to 13.2V supply voltage operation

• Surge voltage protection up to 20V

• Current regul ation limits inrush current regardle ss of load

capacitance

• Programmable inru sh current limiting

• Electronic circuit breaker

• Optional dual-level overcurrent thresh old detects

excessive load faults

• Fast response to short circuit conditions (<1µs)

• Programmable output under-voltage detection

• Under-voltage Lockout (UVLO) protection

• Auto-restart function (MIC2583R)

• Power-On Reset and Power-G ood status outputs

(Power-Good for the MIC2583 an d MIC2583R only)

• /FAULT status output (MIC2583 and MI C2583R)

Applications

• RAID systems

• Base stations

• PC board hot swap insertion and remova l

• +12V backplanes

• Network switches

___________________________________________________________________________________________________________

Typical Applications

Figure 1. MIC2583/83R Typical Application Circuit

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Ordering Information

Part Number

Standard Pb-Free

Fast Circuit Breaker Threshold Circuit Breaker Package

MIC2582-JBM MIC2582-xYM x = J, 100mV

x = J1, Off

x = M, Off

Latched off 8-Pin SOIC

MIC2583-xBQS MIC2583-xYQS x = J, 100mV

x = K*, 150mV

x = L*, 200mV

x = M*, Off

Latched off 16-pin QSOP

MIC2583R-xBQS MIC2583R-xYQS x = J, 100mV

x = K*, 150mV

x = L*, 200mV

x = M*, Off

Auto-retry 16-pin QSOP

Note:

* Contact factory for availability.

Pin Configur ation

8-Pin SOIC (M)

16-Pin QSOP (QS)

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Pin Description

Pin Number

8-Pin SOIC Pin Number

16-Pin QSOP Pin Name Pin Name

1 1 /POR

Power-On Reset Output: Open drain N-channel device, Active Low. This pin

remains asserted during start-up unti l a time perio d (tPOR) after the FB pin

voltage rises above the power-good threshold (VFB). T he timing capacitor CPOR

determines tPOR. When the output voltage monitored at the FB pin falls below

VFB, /POR is asserted for a minimum of one timing cycle (tPOR). The /POR pin

requires a pull-up resistor (10kΩ minimum) to VCC.

2 3 ON

ON Input: Active High. The ON pin, an input to a Schmitt-triggered comparator

used to enable/disable the controller, is compared to a 1.24V reference with

50mV of hysteresis. When a logic high is a pplied to the ON pin (VON > 1.24V), a

start-up sequence begins when the GATE pin starts ramping up towards its final

operating voltage. When the ON pin receives a logic lo w signal (VON < 1.19V),

the GATE pin is grounded and /FAULT remains high if VCC is above the UVLO

threshold. ON must be low for 20µs in order to initiate a start-up sequence.

Additionally, toggling the ON pin LOW to HIGH resets the circuit breaker.

3 4 CPOR

Power-On Reset Timer: A capacitor connected bet ween this pin a nd ground sets

the supply contact start-up delay (tSTART) and the power-on reset interval (tPOR).

When VCC rises above the UVLO threshold, the capacitor connected to CPOR

begins to charge. When the voltage at CPOR crosses 0.3V, the start-up

threshold (VSTART), a start cycle is initiated if ON is asserted while capacitor CPOR

is immediately dischar ged to ground. When the voltage at F B rises above VFB,

capacitor CPOR begins to charge again. Whe n the voltage at CPOR rises above

the power-on reset delay threshold (VTH), the timer resets by pulling CPOR to

ground, and /POR is de-asserted. If CPOR is left open, then tSTART defaults to

20µs.

4 7, 8 GND Ground Connection: T ie to analog ground.

5 12 FB

Power-Good Threshold Input (Under-voltage Detect): T his input is internally

compared to a 1.24V reference with 30mV of hysteresis. An external resistive

divider may be used to set the voltage at this pin. If this input momentarily goes

below 1.24V, then /POR is activated for one timing cycle, tPOR, indicating an

output under-voltage condition. T he /POR signal de-asserts one timing cycle

after the FB pin exceeds the power-good threshold by 30mV. A 5µs filter on this

pin prevents glitches from ina dvertently activating this signal.

6 14 GATE

Gate Drive Output: Connects to the gate of an external N-channel MOSFET. An

internal clamp ensures that no more than 9V is applied between the GATE pin

and the source of the external MOSFET . The GATE pin is immediately brought

low when either the circuit breaker trips or an under-voltage lockout condition

occurs.

7 15 SENSE

Circuit Breaker Sense Input: A resistor between this pin and VCC sets the

current limit threshold. Whenever the voltage across the sen s e resistor exceeds

the slow trip current limit threshold (VTRIPSLOW), the GATE voltage is adjusted to

ensure a constant load current. If VTRIPSLOW (50mV) is exceeded for longer than

time period tOCSLOW, then the circuit breaker is tripped and the GATE pin is

immediately pulled l ow. If the voltage across the sense resis t or exceeds the fast

trip circuit breaker threshold, VTRIPFAST, at any point due to fast, high amplit ude

power supply faults, then the GATE pin is immediately brou ght low without delay.

To disable the circuit breaker, the SENSE an d VCC pins can be tied together.

The default VTRIPFAST for either device is 100mV. Other fast trip thresholds are

available: 150mV, 200mV, or OFF (VTRIPFAST disabled). Please contact factory for

availability of other options.

8 16 VCC

Positive Supply Input: 2.3V to 13.2V. The GATE pin is held low by an internal

under-voltage lockout circuit until VCC exceeds a threshold of 2.2V. If VCC

exceeds 13.2V, an internal shunt regulator protects the chip from transient

voltages up to 20V at the VCC and SENSE pins.

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Pin Number

8-Pin SOIC Pin Number

16-Pin QSOP Pin Name Pin Name

n/a 2 PWRGD

Power-Good Output: Open drain N-channel device, Active High. When the

voltage at the FB pin is lower than 1.24V, PWRGD output is held low. When the

voltage at the FB pin exceeds 1.24V, then PWRGD is asserted immed iately. The

PWRGD pin requires a pull-up resistor (10kΩ minimum) to VCC.

n/a 5 CFILTER

Current Limit Response Timer: A capacitor connected to this pin defines th e

period of time (tOCSLOW) in which an over current event must last to signal a fault

condition and trip the circuit br eaker. If no capacitor is connected, then tOCSLOW

defaults to 5µs.

n/a 11 /FAULT

Circuit Breaker Fault Status Output: Open drain N-chan nel device, Active Low.

The /FAULT pin is asserted when the circuit breaker trips due to an over current

condition or when an under-vo ltage lockout condition exists. The/FAULT pin

requires a pull-up resistor (10kΩ minimum) to VCC.

n/a 13 DIS

Discharge Output: When the MIC2583/83R is turned off, a 500Ω internal resistor

at this output allows the discharging of any load capacitance to ground.

n/a 6, 9, 10 NC No internal connection.

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Absolute Maximum Ratings(1)

Supply Voltage (VCC).......................................–0.3V to 20V

/POR, /FAULT, PWRGD pins.......................... –0.3V to 15V

SENSE pin ............................................ –0.3V to VCC+0.3V

ON pin ............................................ –0.3V to VCC+0.3V

GATE pin ..................................................... –0.3V to 20V

FB Input pin s..................................................... –0.3V to 6V

Junction Temperature..............................................+125°C

Lead Temperature

Standard Package (-JBM and –xBQS)

(IR Reflow, Peak Temperature)..240°C + 0°C/-5°C

Pb-Free Package (-xYM or –xYQS)

(IR Reflow, Peak Temperature)..260°C + 0°C/-5°C

EDS Rating

Human body model..................................................2kV

Machine model ......................................................100V

Operating Ratings(2)

Supply Voltage (VCC)..................................+2.3V to +13.2V

Ambient Temperature (TA)..........................–40°C to +85°C

Junction Thermal Resistance

SOIC (θJA)........................................................163°C/W

QSOP (θJA) ......................................................112°C/W

Electrical Characteristics(3)

VCC = 5.0V, TA = 25°C unless noted. Bold values indicate –40°C ≤ TA ≤ +85°C.

Symbol Parameter Condition Min Typ Max Units

VCC Supply Voltage 2.3 13.2 V

ICC Supply Current VON = 2V 1.5 2.5 mA

VTRIPSLOW 42 50 59

VTRIPFAST

(MIC2582-Jxx)

100

VTRIP Circuit Breaker Trip Voltage

(Current Limit Threshold)

VTRIP = VCC − VSENSE

VTRIPFAST

(MIC2583/83R) X = J

X = K

X = L

130

175

100

150

200

110

170

225

VCC > 3V 7 8 9 V VGS External Gate Drive VGATE − VCC VCC = 2.3V 3.5 4.8 6.5 V

Start Cycle, VGATE = 0V, VCC = 13.2V −30 17 −8 µA IGATE GATE Pin Pull-Up Current VCC = 2.3V −26 17 −8 µA

VCC = 13.2V, Note 4 100 µA

VCC = 2.3V, Note 4 50 µA

IGATEOFF GATE Pink Sink Current VGATE > 1V

/FAULT = 0

(MIC2583/83R only) Turn Off 110 µA

VCC − VSENSE > VTRIPSLOW (timer on) −8.5 −6.5 −4.5 µA ITIMER Current Limit/Overcurrent Timer

(CFILTER) Current

(MIC2583/83R) VCC − VSENSE > VTRIPSLOW (timer off) 4.5 6.5 8.5 µA

Timer on −3.5 2.5 −1.5 µA ICPOR Power-On-Reset Timer Current Timer off 0.5 1.3 mA

VTH POR Delay and Overcurrent

Timer (CFILTER) Threshold VCPOR rising

VCFILTER rising (MIC2583/83R only)

1.19

1.245

1.30

VCC rising 2.1 2.2 2.3 V VUV Undervoltage Lockout Threshold VCC falling 1.90 2.05 2.20 V

VUVHYS Undervoltage Lockout Hysteresis 150 mV

ON rising 1.19 1.24 1.29 V VON ON Pin Threshold Voltage 2.3V ≤ VCC ≤ 13.2V ON falling 1.14 1.19 1.24 V

VONHYS ON Pin Hysteresis 50 mV

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Symbol Parameter Condition Min Typ Max Units

ΔVON ON Pin Threshold Line Regulation 2.3V ≤ VCC ≤ 13.2V 2 mV

ION ON Pin Input Current VON = VCC −0.5 µA

VSTART Start-Up Delay Timer Threshold VCPOR rising 0.26 0.31 0.36 V

Upper threshold 0.19 1.24 1.30 V VAUTO Auto-Restart Threshold Voltage

(MIC2583R only) Lower threshold 0.26 0.31 0.36 V

Charge current 10 13 16 µA IAUTO Auto-Restart Current

(MIC2583R only) Discharge current 1.4 2 µA

FB rising 1.19 1.24 1.29 V VFB Power-Good Threshold Voltage 2.3V = VCC = 13.2V FB falling 1.15 1.20 1.25 V

VFBHYS FB Hysteresis 40 mV

IFBLKG FB Pin Leakage Current 2.3V = VCC = 13.2V, VFB = 1.3V 1.5 µA

VOL /POR, /FAULT, PWRGD

Output Voltage

(/FAULT, PWRGD MIC2583/83R only)

IOUT = 1mA 0.4 V

RDIS Output Discharge Resistance

(MIC2583/83R only) 500

1000 Ω

AC Parameters(4)

tOCFAST Fast Overcurrent SENSE to GATE

Low Trip Time VCC = 5V, VCC − VSENSE = 100mV

CGATE = 10nF, Figure 2 1 µs

tOCSLOW Slow Over cu r rent SE NSE to GA TE

Low Trip Time VCC = 5V, VCC − VSENSE = 50mV

CFILTER = 0, Figure 2 5 µs

tONDLY ON Delay Filter 20 µs

tFBDLY FB Delay Filter 20 µs

Notes:

1. Exceeding the absolute maximum rating may damage the device.

2. The device is not guaranteed to function outside its operating rating.

3. Specification for packaged product only.

4. Not a tested parameter, guaranteed by design.

Timing Diagrams

Figure 2. Current-Limit Response

Figure 3. Power-On Reset Response

Figure 4. Power-On Start-Up Delay Timing

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Test Circuit

Figure 5. Applications Test Circuit

(not all pins shown for simplicity)

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Typical Characteristics

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Functional Characteristics (See Figure 5, Applicatio ns Test Circuit)

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Functional Characteristics (See Figure 5, Applicatio ns Test Circuit)

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Functional Diagram

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Functional Description

Hot Swap Insertion

When circuit boards are inserted into live system

backplanes and supply voltages, high inrush currents

can result due to the charging of bulk capacitance that

resides across the supply pins of the circuit board. This

inrush current, although transient in nature, may be high

enough to cause permanent damage to on board

components or may cause the system’s supply voltages

to go out of regulation during the transient period which

may result in system failures. The MIC2582 and

MIC2583 act as a controller for external N-Channel

MOSFET devices in which the gate drive is controlled to

provide inrush current limiting and output voltage slew

rate control during hot plug insertions.

Power Supply

VCC is the supply input to the MIC2582/83 controller

with a voltage range of 2.3V to 13.2V. The VCC input

can withstand transient spikes up to 20V. In order to

ensure stability of the supply voltage, a minimum 0.47µF

capacitor from VCC to ground is recommended.

Alternatively, a low pass filter, shown in the typical

application circuit (see Figure 1), can be used to

eliminate high frequency oscillations as well as help

suppress transient spikes.

Also, due to the existence of an undetermined amount of

parasitic inductance in the absence of bulk capacitance

along the supply path, placing a Zener diode at the VCC

of the controller to ground in order to provide external

supply transient protection is strongly recommended for

relatively high current applications (≥3A). See Figure 1.

Start-Up Cycle

Supply Contact Delay

During a hot insert of a PC board into a backplane or

when the supply (VCC) is powered up, as the voltage at

the ON pin rises above its threshold (1.24V typical), the

MIC2582/83 first checks that both supply voltages are

above their respective UVLO thresholds. If so, the

device is enabled and an internal 2.5µA current source

begins charging capacitor CPOR to 0.3V to initiate a start-

up sequence. Once the start-up delay (tSTART) elapses,

the CPOR pin is pulled immediately to ground and a

17µA current source begins charging the GATE output

to drive the external MOSFET that switches VIN to VOUT.

The programmed contact start-up delay is calculated

using the following equation:

()

FCx

Ct POR

CPOR

START

PORSTART

12.0≅

⎥

⎦

⎤

⎢

⎣

⎡

= (1)

Where the start-up delay timer thre shold (VSTART) is 0.3V,

and the Power-On Reset timer current (ICPOR) is 2.5µA.

See Table 2 for some typical supply contact start-up

delays using several standard value capacitors. As the

GATE voltage continues ramping toward its final value

(VCC + VGS) at a defined slew rate (See Load

Capacitance/Gate Capacitance Dominated Startup

sections), a second CPOR timing cycle begins if:

1)/FAULT is high and 2)CFILTER is low (i.e., not an

overvoltage, undervoltage lockout, or overcurrent state).

This second timing cycle (tPOR) begins when the voltage

at the FB pin exceeds its threshold (VFB). This condition

indicates that the output voltage is valid. See Figure 3 in

the Timing Diagrams. When the power supply is already

present (i.e., not a “hot swapping” condition) and the

MIC2582/83 device is enabled by applying a logic high

signal at the ON pin, the GATE output begins ramping

immediately as the first CPOR timing cycle is bypassed.

Active current regulation is employed to limit the inrush

current transient response during start-up by regulating

the load current at the programmed current limit value

(See Current Limiting and Dual-Level Circuit Breaker

section). The following equation is used to determine the

nominal current limit value:

SENSESENSE

TRIPSLOW

LIM RmV

I50

== (2)

where VTRIPSLOW is the current limit slow trip threshold

found in the electrical table and RSENSE is the selected

value that will set the desired current limit. There are two

basic start-up modes for the MIC2582/83: 1) Start-up

dominated by load capacitance and 2) start-up

dominated by total gate capacitance. The magnitude of

the inrush current delivered to the load will determine the

dominant mode. If the inrush current is greater than the

programmed current limit (ILIM), then load capacitance is

dominant. Otherwise, gate capacitance is dominant. The

expected inrush current may be calculated using the

following equation:

GATE

LOAD

GATE

LOAD

GATE C

xIINRUSH

17=≅ (3)

where IGATE is the GATE pin pull-up current, CLOAD is the

load capacitance, and CGATE is the total GATE

capacitance (CISS of the external MOSFET and any

external capacitor connected from the MIC2582/83

GATE pin to ground).

Load Capacitance Dominated Start-Up

In this case, the load capacitance (CLOAD) is large

enough to cause the inrush current to exceed the

programmed current limit but is less than the fast-trip

threshold (or the fast-trip threshold is disabled, ‘M’

option). During start-up under this condition, the load

current is regulated at the programmed current limit

value (ILIM) and held constant until the output voltage

rises to its final value. The output slew rate and

equivalent GATE voltage slew rate is computed by the

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following equation:

Output Voltage Slew Rate, dVOUT/dt =

LOAD

LIM

CI (4)

where ILIM is the programmed current limit value.

Consequently, the value of CFILTER must be selected to

ensure that the overcurrent response time, tOCSLOW,

exceeds the time needed for the output to reach its final

value. For example, given a MOSFET with an input

capacitance CISS = CGATE = 4700pF, CLOAD is 2200µF,

and ILIM is set to 6A with a 12V input, then the load

capacitance dominates as determined by the calculated

INRUSH > ILIM. Therefore, the output voltage slew rate

determined from Equation 4 is:

Output Voltage Slew Rate, dVOUT/dt = ms

A73.2

2200

and the resulting tOCSLOW needed to achieve a 12V

output is approximately 4.5ms. (See Power-On Reset

and Overcurrent Timer Delays section to calculate

tOCSLOW).

GATE Capacitance Dominated Start-Up

In this case, the value of the load capacitance relative to

the GATE capacitance is small enough such that the

load current during start-up never exceeds the current

limit threshold as determined by Equation 3. The

minimum value of CGATE that will ensure that the current

limit is never exceeded is given by the equation below:

LOAD

LIM

GATE

GATE C

C×=(min) (5)

where CGATE is the summation of the MOSFET input

capacitance (CISS) and the value of the external

capacitor connected to the GATE pin of the MIC2582/83

to ground. Once CGATE is determined, use the following

equation to determine the output slew rate for gate

capacitance dominated start-u p.

dVOUT/dt = GATE

GATE

I (6)

Table 1 depicts the output slew rate for various values of

CGATE.

IGATE = 17µA

CGATE dVOUT/dt

0.001µF 17V/ms

0.01µF 1.7V/ms

0.1µF 0.17V/ms

1µF 0.017V/ms

Table 1. Output Slew Rate Selection for GATE

Capacitance Dominated Start-Up

Current Limiting and Dual-Level Circuit Breaker

Many applications will require that the inrush and steady

state supply current be limited at a specific value in order

to protect critical components within the system.

Connecting a sense resistor between the VCC and

SENSE pins sets the nominal current limit value of the

MIC2582/83 and the current limit is calculated using

Equation 2.

The MIC2582/83 also features a dual-level circuit

breaker triggered via 50mV and 100mV current-limit

thresholds sensed across the VCC and SENSE pins.

The first level of the circuit breaker functions as follows.

For the MIC2583/83R, once the voltage sensed across

these two pins exceeds 50mV, the overcurrent timer, its

duration set by capacitor CFILTER, starts to ramp the

voltage at CFILTER using a 6.5µA constant current source.

If the voltage at CFILTER reaches the overcurrent timer

threshold (VTH) of 1.24V, then CFILTER immediately

returns to ground as the circuit breaker trips and the

GATE output is immediately shut down. The default

overcurrent time period for the MIC2582/83 is 5µs. For

the second level, if the voltage sensed across VCC and

SENSE exceeds 100mV at any time, the circuit breaker

trips and the GATE shuts down immediately, bypassing

the overcurrent time period. The MIC2582-MYM option

is equipped with only a single circuit breaker threshold

(50mV). To disable current limit and circuit breaker

operation, tie the SENSE and VCC pins together and the

CFILTER (MIC2583/83R) pin to ground.

Output Undervoltage De tection

The MIC2582/83 employ output undervoltage detection

by monitoring the output voltage through a resistive

divider connected at the FB pin. During turn on, while the

voltage at the FB pin is below the threshold (VFB), the

/POR pin is asserted low.

Once the FB pin voltage crosses VFB, a 2.5µA current

source charges capacitor CPOR. Once the CPOR pin

voltage reaches 1.24V, the time period tPOR elapses as

the CPOR pin is pulled to ground and the /POR pin goes

HIGH. If the voltage at FB drops below VFB for more than

10µs, the /POR pin resets for at least one timing cycle

defined by tPOR (See Applications Information for an

example).

Power-On Reset and Overcurrent Timer Delays

The Power-On Reset delay, tPOR, is the time period for

the /POR pin to go HIGH once the voltage at the FB pin

exceeds the power-good threshold (VFB). A capacitor

connected to CPOR sets the interval and is determined

by using Equation 1 with VTH substituted for VSTART. The

resulting equation becomes:

()

Ct POR

CPOR

PORPOR

×≅×= 5.0 (7)

where the Power-On Reset threshold (VTH) and timer

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current (ICPOR) are typically 1.24V and 2.5µA,

respectively.

For the MIC2583/83R, a capacitor connected to

CFILTER is used to set the timer which activates the

circuit breaker during overcurrent conditions. When the

voltage across the sense resistor exceeds the slow trip

current limit threshold of 50mV, the overcurrent timer

begins to charge for a time period (tOCSLOW), determined

by CFILTER. When no capacitor is connected to CFILTER

and for the MIC2582, tOCSLOW defaults to 5µs. If tOCSLOW

elapses, then the circuit breaker is activated and the

GATE output is immediately pulled to ground. For the

MIC2583/83R, the following equation is used to

determine the overcurrent timer period, tOCSLOW.

)(0.19 FC

Ct FILTER

TIMER

FILTEROCSLOW

×≅×= (8)

where VTH, the CFILTER timer threshold, is 1.24V and

ITIMER, the overcurrent timer current, is 6.5µA. Tables 2

and 3 provide a quick reference for several timer

calculations using sele ct standard value capacitors.

CPOR t

START t

POR

0.01µF 1.2ms 5ms

0.02µF 2.4ms 10ms

0.033µF 4ms 16.5ms

0.05µF 6ms 25ms

0.1µF 12ms 50ms

0.33µF 40ms 165ms

0.47µF 56ms 235ms

1µF 120ms 500ms

Table 2. Selected Power-On Reset and Start-Up Delays

CFILTER t

OCSLOW

680pF 130µs

2200pF 420µs

4700pF 900µs

8200pF 1.5ms

0.033µF 6ms

0.1µF 19ms

0.22µF 42ms

0.47µF 90ms

Table 3. Selected Overcurrent Timer Delays

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Application Information

Design Consideration for Output Unde rvoltage

Detection

For output undervoltage detection, the first consideration

is to establish the output voltage level that indicates

“power is good.” For this example, the output value for

which a 12V supply will signal “good” is 11V. Next,

consider the tolerances of the input supply and FB

threshold (VFB). For this example, the 12V supply varies

±5%, thus the resulting output voltage may be as low as

11.4V and as high as 12.6V. Additionally, the FB

threshold has ±50mV tolerance and may be as low as

1.19V and as high as 1.29V. Thus, to determine the

values of the resistive divider network (R5 and R6) at the

FB pin, shown in the typical application circuit on page 1,

use the following iterative design proced ure.

1) Choose R6 to allow 100µA or more in the FB

resistive divider branch.

kΩ.

R912

100

1.29V

100

6FB(MAX) ===

μμ

R6 is chosen as 12.4kΩ ±1%.

2) Next, determine R5 using the output “good”

voltage of 11V and the following equation:

()

⎥

⎦

⎤

⎢

⎣

⎡+

=R6R6R5

FBOUT(Good) VV (9)

Using some basic alge bra and simplifying Equation 9 to

isolate R5, yields:

⎥

⎦

⎤

⎢

⎣

⎡−

⎟

⎠

⎞

⎜

⎝

⎛

=165 FB(MAX)

OUT(Good)

RR (9.1)

where VFB(MAX) = 1.29V, VOUT(Good) = 11V, and R6 is

12.4kΩ. Substituting these values into Equation 9.1 now

yields R5 = 93.33kΩ. A standard 93.1kΩ ±% is selected.

Now, consider the 11.4V minimum output voltage, the

lower tolerance for R6 and higher tolerance for R5,

12.28kΩ and 94.03kΩ, respectively. With only 11.4V

available, the voltage sensed at the FB pin exceeds

VFB(MAX), thus the /POR and PWRGD (MIC2583/83R)

signals will transition from LOW to HIGH, indicating

“power is good” given the worse case tolerances of this

example. Lastly, in giving consideration to the leakage

current associated with the FB input, it is recommended

to either: 1) provide ample design margin (20mV to

30mV) to allow for loss in the potential (∆V) at the FB

pin, or 2) allow >>100µA to flow in the FB resistor

network.

PCB Connection Sense

There are several configuration options for the

MIC2582/83’s ON pin to detect if the PCB has been fully

seated in the backplane before initiating a start-up cycle.

In the typical applications circuit, the MIC2582/83 is

mounted on the PCB with a resistive divider network

connected to the ON pin. R2 is connected to a short pin

on the PCB edge connector. Until the connectors mate,

the ON pin is held low which keeps the GATE output

charge pump off. Once the connectors mate, the resistor

network is pulled up to the input supply,

Figure 6. PCB Connection Sense with ON/OFF Control

Micrel, Inc. MIC2582/MIC2583

April 2009 17 M9999-043009-C

12V in this example, and the ON pin voltage exceeds its

threshold (VON) of 1.24V and the MIC2582/83 initiates a

start-up cycle. In Figure 6, the connection sense consisting

of a discrete logic-level MOSFET and a few resistors allows

for interrupt control from the processor or other signal

controller to shut off the output of the MIC2582/83. R4 pulls

the GATE of Q2 to VIN and the ON pin is held low until the

connectors are fully mated.

Once the connectors fully mate, a logic LOW at the

/ON_OFF signal turns Q2 off and allows the ON pin to pull

up above its threshold and initiate a start-up cycle. Applying

a logic HIGH at the /ON_OFF signal will turn Q2 on and

short the ON pin of the MIC2582/83 to ground which turns

off the GATE output charge pump.

Higher UVLO Setting

Once a PCB is inserted into a backplane (power supply ), the

internal UVLO circuit of the MIC2582/83 holds the GATE

output charge pump off until VCC ex ceeds 2.2V. If VCC falls

below 2.1V, the UVLO circuit pulls the GATE output to

ground and clears the overvoltage and/or current limit faults.

A typical 12V application, for example, should implement a

higher UVLO than the internal 2.1V threshold of MIC2582 to

avoid delivering power to downstream modules/loads while

the input is below tolerance. For a higher UVLO threshold,

the circuit in Figure 7 can be used to delay the output

MOSFET from switching on until the desired input voltage is

achieved. The circuit allows the charge pump to remain off

until VIN exceeds .24.1

R×

⎟

⎠

⎞

⎜

⎝

⎛+ The GATE drive output

will be shut down when VIN falls below .19.1

R×

⎟

⎠

⎞

⎜

⎝

⎛+ In

the example circuit (Figure 7), the rising UVLO threshold is

set at approximately 9.5V and the falling UVLO threshold is

established as 9.1V. The circuit consists of an external

resistor divider at the ON pin that keeps the GATE output

charge pump off until the voltage at the ON pin exceeds its

threshold (VON) and after the start-up timer elapse s.

5V Switch with 3.3V Supply Generation

The MIC2582/83 can be configured to switch a primary

supply while generating a secondary regulated voltage rail.

The circuit in Figure 8 enables the MIC2582 to switch a 5V

supply while also providing a 3.3V low dropout regulated

supply with only a few added external components. Upon

enabling the MIC2582, the GATE output voltage increases

and thus the 3.3V supply also begins to ramp. As the 3.3V

output supply crosses 3.3V, the FB pin threshold is also

exceeded which triggers the power-on reset comparator.

The /POR pin goes HIGH, turning on transistor Q3 which

lowers the voltage on the gate of MOSFET Q2. The result is

a regulated 3.3V supply with the gate feedback loop of Q2

compensated by capacitor C3 and resistors R4 and R5. For

MOSFET Q2, special consideration must be given to the

power dissipation capability of the selected MOSFET as

1.5V to 2V will drop across the device during normal

operation in this application. Therefore, the device is

susceptible to overheating dependent upon the current

requirements for the regulated output. In this example, the

power dissipated by Q2 is approximately = 1W. However, a

substantial amount of power will be generated with higher

current requirements and/or conditions. As a general

guideline, expect the ambient temperature within the power

supply box to exceed the maximum operating ambient

temperature of the system environment by approximately

20ºC. Given the MOSFET’s Rθ(JA) and the expected power

dissipated by the MOSFET, an approximation for the

junction temperature at which the device will operate is

obtained as follows:

TJ = (PD x Rθ(JA)) + TA (10)

where TA = TA (MAX OPERATING) + 20ºC. As a precaution, the

implementation of additional copper heat sinking is highly

recommended for the area under/around the MOSFET.

Figure 7. Higher UVLO Setting

Micrel, Inc. MIC2582/MIC2583

April 2009 18 M9999-043009-C

For additional information on MOSFET thermal

considerations, please see MOSFET Selection text and

subsequent sections.

Auto-Restart - MIC2583R

The MIC2583R provides an auto-restart function. Upon

an overcurrent fault condition such as a short circuit, the

MIC2583R initially shuts off the GATE output. The

MIC2583R attempts to restart with a 12µA charge

current at a preset 10% duty cycle until the fault

condition is removed. The interval between auto-retry

attempts is set by capacitor CFILTER.

Sense Resistor Selection

The MIC2582 and MIC2583 use a low-value sense

resistor to measure the current flowing through the

MOSFET switch (and therefore the load). This sense

resistor is nominally set at 50mV/ILOAD(CONT). To

accommodate worst-case tolerances for both the sense

resistor (allow ±3% over time and temperature for a

resistor with ±1% initial tolerance) and still supply the

maximum required steady-state load current, a slightly

more detailed calculation must be used.

The current limit threshold voltage (i.e., the “trip point”)

for the MIC2582/83 may be as low as 42mV, which

would equate to a sense resistor value of

42mV/ILOAD(CONT). Carrying the numbers through for the

case where the value of the sense resistor is 3% high

yields:

()

)()(

)( 8.40

03.1 42

CONTLOADCONTLOAD

MAXSENSE ImV

ImV

R== (11)

Once the value of RSENSE has been chosen in this

manner, it is good practice to check the maximum

ILOAD(CONT) which the circuit may let through in the case of

tolerance buildup in the opposite direction. Here, the

worst-case maximum current is found using a 59mV trip

voltage and a sense resistor that is 3% low in value. The

resulting equation is:

()

)()(

),( 8.60

)97.0( 59

NOMSENSENOMSENSE

MAXCONTLOAD RmV

RmV

I== (12)

As an example, if an output must carry a continuous 2A

without nuisance trips o ccurring, Equation 11

yields: .4.20

8.40

)( Ω== m

RMAXSENSE The next

lowest standard value is 20mΩ. At the other set of

tolerance extremes for the output in question,

,04.3

0.20 8.60

),( A

IMAXCONTLOAD =

= approximately 3A.

Knowing this final data, we can determine the necessary

wattage of the sense resistor using P = I2R, where I will

be ILOAD(CONT, MAX), and R will be (0.97)(RSENSE(NOM)).

These numbers yield the following: PMAX = (3A)2

(19.4mΩ) = 0.175W.

In this example, a ¼W sense resist or is sufficient.

Figure 8. 5V Switch/3.3V LDO Application

Micrel, Inc. MIC2582/MIC2583

April 2009 19 M9999-043009-C

MOSFET Selection

Selecting the proper external MOSFET for use with the

MIC2582/83 involves three straightfo rward tasks:

• Choice of a MOSFET which meets minimum voltage

requirements.

• Selection of a device to handle the maximum

continuous current (steady-state thermal issues).

• Verify the selected part’s ability to withstand any peak

currents (transient thermal issues).

MOSFET Voltage Requirements

The first voltage requirement for the MOSFET is easily

stated: the drain-source breakdown voltage of the

MOSFET must be greater than VIN(MAX). For instance, a

12V input may reasonably be expected to see high-

frequency transients as high as 18V. Therefore, the

drain-source breakdown voltage of the MOSFET must

be at least 19V. For ample safety margin and standard

availability, the closest value will be 20V.

The second breakdown voltage criterion that must be

met is a bit subtler than simple drain-source breakdown

voltage, but is not hard to meet. In MIC2582/83

applications, the gate of the external MOSFET is driven

up to approximately 19.5V by the internal output

MOSFET (again, assuming 12V operation).

At the same time, if the output of the external MOSFET

(its source) is suddenly subjected to a short, the gate-

source voltage will go to (19.5V – 0V) = 19.5V. This

means that the external MOSFET must be chosen to

have a gate-source breakdown voltage of 20V or more,

which is an available standard maximum value.

However, if operation is at or above 13V, the 20V gate-

source maximum will likely be exceeded. As a result, an

external Zener diode clamp should be used to prevent

breakdown of the external MOSFET when operating at

voltages above 8V. A Zener diode with 10V rating is

recommended as shown in Figure 9. At the present time,

most power MOSFETs with a 20V gate-source voltage

rating have a 30V drain-source breakdown rating or

higher.

As a general tip, choose surface-mount devices with a

drain-source rating of 30V as a starting point.

Finally, the external gate drive of the MIC2582/83

requires a low-voltage logic level MOSFET when

operating at voltages lower than 3V. There are 2.5V

logic level MOSFETs available. Please see Table 4

“MOSFET and Sense Resistor Vendors” for

suggested manufacturers.

Figure 9. Zener Clamped MOSFET Gate

Micrel, Inc. MIC2582/MIC2583

April 2009 20 M9999-043009-C

MOSFET Steady-State Thermal Issues

The selection of a MOSFET to meet the maximum

continuous current is a fairly straightforward exercise.

First, arm yourself with the following data:

• The value of ILOAD(CONT, MAX.) for the output in

question (see Sense Resistor Selection).

• The manufacturer’s data sheet for the

candidate MOSFET.

• The maximum ambient temperature in which

the device will be require d to operate.

• Any knowledge you can get about the heat

sinking available to the device (e.g., can heat

be dissipated into the ground plane or power

plane, if using a surface-mount part? Is any

airflow available?).

The data sheet will almost always give a value of on

resistance given for the MOSFET at a gate-source

voltage of 4.5V, and another value at a gate-source

voltage of 10V. As a first approximation, add the two

values together and divide by two to get the on-

resistance of the part with 8V of enhancement.

Call this value RON. Since a heavily enhanced MOSFET

acts as an ohmic (resistive) device, almost all that’s

required to determine steady-state power dissipation is

to calculate I2R.

The one addendum to this is that MOSFETs have a

slight increase in RON with increasing die temperature. A

good approximation for this value is 0.5% increase in

RON per ºC rise in junction temperature above the point

at which RON was initially specified by the manufacturer.

For instance, if the selected MOSFET has a calculated

RON of 10mΩ at a TJ = 25ºC, and the actual junction

temperature ends up at 110ºC, a good first cut at the

operating value for RON would be:

()()

[]

Ω≅−+Ω≅ mmRON 3.14005.025110110 (13)

The final step is to make sure that the heat sinking

available to the MOSFET is capable of dissipating at

least as much power (rated in ºC/W) as that with which

the MOSFETs performance was specified by the

manufacturer. Here are a few practical tips:

1. The heat from a surface-mount device such as

an SOIC-8 MOSFET flows almost entirely out

of the drain leads. If the drain leads can be

soldered down to one square inch or more, the

copper will act as the heat sink for the part.

This copper must be on the same layer of the

board as the MOSFET drain.

2. Airflow works. Even a few LFM (linear feet per

minute) of air will cool a MOSFET down

substantially.

If you can, position the MOSFET(s) near the

inlet of a power supply’s fan, or the outlet of a

processor’s cooling fan.

3. The best test of a surface-mount MOSFET for

an application (assuming the above tips show

it to be a likely fit) is an empirical one. Check

the MOSFETs temperature in the actual layout

of the expected final circuit, at full operating

current. The use of a thermocouple on the

drain leads, or infrared pyrometer on the

package, will then give a reasonable idea of

the device’s junction temperature.

MOSFET Transient Thermal Issues

Having chosen a MOSFET that will withstand the

imposed voltage stresses, and the worse case

continuous I2R power dissipation which it will see, it

remains only to verify the MOSFETs ability to handle

short-term overload power dissipation without

overheating. A MOSFET can handle a much higher

pulsed power without damage than its continuous

dissipation ratings would imply. The reason for this is

that, like everything else, thermal devices (silicon die,

lead frames, etc.) have thermal inertia.

In terms related directly to the specification and use of

power MOSFETs, this is known as “transient thermal

impedance,” or Zθ(JA). Almost all power MOSFET data

sheets give a Transient Thermal Impedance Curve. For

example, take the following case: VIN = 12V, tOCSLOW has

been set to 100msec, ILOAD(CONT. MAX) is 2.5A, the slow-

trip threshold is 50mV nominal, and the fast-trip

threshold is 100mV. If the output is accidentally

connected to a 3Ω load, the output current from the

MOSFET will be regulated to 2.5A for 100ms (tOCSLOW)

before the part trips. During that time, the dissipation in

the MOSFET is given by:

P = E x I; EMOSFET = [12V-(2.5A)(3Ω)] = 4.5V

MOSFET = (4.5V x 2.5A) = 11.25W for 100msec.

At first glance, it would appear that a really hefty

MOSFET is required to withstand this sort of fault

condition. This is where the transient thermal impedance

curves become very useful. Figure 10 shows the curve

for the Vishay (Siliconix) Si4410DY, a commonly used

SOIC-8 power MOSFET.

Taking the simplest case first, we’ll assume that once a

fault event such as the one in question occurs, it will be

a long time– 10 minutes or more– before the fault is

isolated and the channel is reset. In such a case, we can

approximate this as a “single pulse” event, that is to say,

there’s no significant duty cycle. Then, reading up from

the X-axis at the point where “Square Wave Pulse

Duration” is equal to 0.1sec (=100msec), we see that the

Zθ(JA) of this MOSFET to a highly infrequent event of this

duration is only 8% of its continuous Rθ(JA).

This particular part is specified as having an Rθ(JA) of

50°C/W for inte rvals of 10 secon ds or less.

Micrel, Inc. MIC2582/MIC2583

April 2009 21 M9999-043009-C

Thus:

Assume TA = 55°C maximum, 1 square inch of copper at

the drain leads, no airflow.

Recalling from our previous approximation hint, the part

has an RON of (0.0335/2) = 17mΩ at 25°C.

Assume it has been carrying just about 2.5A for some

time.

When performing this calculation, be sure to use the

highest anticipated ambient temperature (TA(MAX)) in

which the MOSFET will be operating as the starting

temperature, and find the operating junction temperature

increase (∆TJ) from that point. Then, as shown next, the

final junction temperature is found by adding TA(MAX) and

∆TJ. Since this is not a closed-form equation, getting a

close approximation may take one or two iterations, and

the calculation tends to converge quickly.

Then the starting (steady-state) TJ is:

J ≅ TA(MAX) + ∆TJ

≅ TA(MAX) + [RON + TA(MAX) – TA)(0.005/ºC)(RON)]

x I2 x Rθ(JA)

TJ ≅ 55ºC + [17mΩ + (55º C-25ºC)(0.005)(17mΩ)]

x (2.5A)2 x (50ºC/W)

TJ ≅ (55ºC + (0.122W)(50ºC/W)

≅ 61.1ºC

Iterate the calculation once to see if this value is within a

few percent of the expected final value. For this iteration

we will start with TJ equal to the already calculated value

of 61.1°C:

TJ ≅ TA + [17mΩ + (61.1ºC-25ºC)(0.005)(17mΩ)]

x (2.5A)2 x (50ºC/W)

TJ ≅ (55ºC + (0.125W)(50ºC/W) ≅ 61.27ºC

So our original approximation of 61.1ºC was very close

to the correct value. We will use TJ = 61ºC.

Finally, add the temperature increase due to the

maximum power dissipation calculated from a “single

event”, (11.25W)(50ºC/W)(0.08) = 45ºC to the steady-

state TJ to get TJ(TRANSIENT MAX.) = 106ºC. This is an

acceptable maximum junction temperature for this part.

Figure 10. Transient Thermal Impedance

Micrel, Inc. MIC2582/MIC2583

April 2009 22 M9999-043009-C

PCB Layout Considera tions

Because of the low values of the sense resistors used

with the MIC2582/83 controllers, special attention to the

layout must be used in order for the device’s circuit

breaker function to operate properly. Specifically, the

use of a 4-wire Kelvin connection to accurately measure

the voltage across RSENSE is highly recommended. Kelvin

sensing is simply a means of making sure that any

voltage drops in the power traces connecting to the

resistors does not get picked up by the traces

themselves. Additionally, these Kelvin connections

should be isolated from all other signal traces to avoid

introducing noise onto these sensitive nodes. Figure 11

illustrates a recommended, single layer layout for the

RSENSE, Power MOSFET, timer(s), and feedback network

connections. The feedback network resistor values are

selected for a 12V application. Many hot swap

applications will require load currents of several

amperes. Therefore, the power (VCC and Return) trace

widths (W) need to be wide enough to allow the current

to flow while the rise in temperature for a given copper

plate (e.g., 1oz. or 2oz.) is kept to a maximum of

10ºC~25ºC. Also, these traces should be as short as

possible in order to minimize the IR drops between the

input and the load.

Finally, the use of plated-through vias will be needed to

make circuit connections to power and ground planes

when utilizing multi-layer PC boards.

MOSFET and Sense Resistor Vendors

Device types and manufacturer contact information for

power MOSFETs and sense resistors are provided in

Table 4. Some of the recommended MOSFETs include a

metal heat sink on the bottom side of the package. The

recommended trace for the MOSFET Gate of Figure 11

must be redirected when using MOSFETs packaged in

this style. Contact the device manufacturer for package

information.

Figure 11. Recommended PCB Layout for Sense Resistor,

Power MOSFET, and F eedback Network

Micrel, Inc. MIC2582/MIC2583

April 2009 23 M9999-043009-C

MOSFET Vendor Key MOSFET Type(s) Applications(1) Contact Information

Vishay (Siliconix) Si44 20DY (SOIC-8) package

Si4442DY (SOIC-8) package

Si4876DY (SOIC-8) package

Si7892DY (PowerPAK™ SOIC-8)

IOUT ≤ 10A

IOUT = 10-15A, VCC < 3V

IOUT ≤ 5A, VCC ≤ 5V

IOUT ≤ 15A

www.siliconix.com

(203) 452-5664

International Rectifier IRF7413 (SOIC-8 package)

IRF7457 (SOIC-8 package)

IRF7601 (SOIC-8 package)

IOUT ≤ 10A

IOUT = 10-15A

IOUT ≤ 5A, VCC ≤ 3V

www.irf.com

(310) 322-3331

Fairchild Semiconductor F DS6680A (SOIC-8 package) IOUT ≤ 10A www.fairchildsemi.com

(207) 775-8100

Philips PH3230 (SOT669-LFPAK) IOUT ≥ 20A www.philips.com

Hitachi HAT2099H (LFPAK) IOUT ≥ 20A www.halsp.hitachi.com

(408) 433-1990

Note:

1. These devices are not limited to these conditions in many cases, but these conditions are provided as a helpful reference for customer applications.

Resistor Vendors Sense Resistors Contact Information

Vishay (Dale) “WSL” Series www.vishay.com/docswsl_30100.pdf

(203) 452-5664

IRC “OARS” Series

”LR” Series

(second source to “WSL”)

www.irctt.com/pdf_files/OARS.pdf

www.irctt.com/pdf_files/LRC.pdf

(828) 264-8861

Table 4. MOSFET and Sense Resistor Vendors

Micrel, Inc. MIC2582/MIC2583

April 2009 24 M9999-043009-C

Package Information

8-Pin SOIC (M)

Micrel, Inc. MIC2582/MIC2583

April 2009 25 M9999-043009-C

16-Pin QSOP (QS)

MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA

TEL +1 (408) 944-0800 FAX + 1 (408) 474-1000 WEB http://www.micrel.com

The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its

use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.

Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product

can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant

into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A

Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully

indemnify Micrel for any damages resulting from such use or sale.