General Description
The MAX668/MAX669 constant-frequency, pulse-width-
modulating (PWM), current-mode DC-DC controllers are
designed for a wide range of DC-DC conversion applications
including step-up, SEPIC, flyback, and isolated-output
configurations. Power levels of 20W or more can be
controlled with conversion efficiencies of over 90%. The
1.8V to 28V input voltage range supports a wide range of
battery and AC-powered inputs. An advanced BiCMOS
design features low operating current (220μA), adjustable
operating frequency (100kHz to 500kHz), soft-start, and
a SYNC input allowing the MAX668/MAX669 oscillator to
be locked to an external clock.
DC-DC conversion efficiency is optimized with a low
100mV current-sense voltage as well as with Maxim’s
proprietary Idle Mode™ control scheme. The controller
operates in PWM mode at medium and heavy loads for
lowest noise and optimum efficiency, then pulses only as
needed (with reduced inductor current) to reduce operating
current and maximize efficiency under light loads. A logic-
level shutdown input is also included, reducing supply
current to 3.5μA.
The MAX669, optimized for low input voltages with a
guaranteed start-up voltage of 1.8V, requires bootstrapped
operation (IC powered from boosted output). It supports
output voltages up to 28V. The MAX668 operates with
inputs as low as 3V and can be connected in either a boot-
strapped or non-bootstrapped (IC powered from input sup-
ply or other source) configuration. When not bootstrapped,
it has no restriction on output voltage. Both ICs are avail-
able in an extremely compact 10-pin μMAX package.
Benets and Features
●1.8V Minimum Start-Up Voltage (MAX669)
●Wide Input Voltage Range (1.8V to 28V)
● Tiny 10-Pin μMAX Package
●Current-Mode PWM and Idle Mode™ Operation
●Efficiency over 90%
● Adjustable 100kHz to 500kHz Oscillator or
SYNC Input
● 220μA Quiescent Current
●Logic-Level Shutdown
●Soft-Start
Applications
●Cellular Telephones
●Telecom Hardware
●LANs and Network Systems
●POS Systems
19-4778; Rev 3; 6/16
Idle Mode is a trademark of Maxim Integrated Products.
+ Denotes a lead(Pb)-free/RoHS-compliant package.
T = Tape and reel.
/V Denotes an automotive qualified part.
Note: Devices are also available in a lead(Pb)-free/RoHS-
compliant package. Specify lead-free by adding “+” to the part
number when ordering.
PART TEMP RANGE PIN-PACKAGE
MAX668EUB -40°C to +85°C 10 µMAX
MAX669EUB -40°C to +85°C 10 µMAX
MAX669EUB/V+T -40°C to +85°C 10 µMAX
MAX669
FREQ CS+
SYNC/
SHDN
PGND
FB
GND
VCC EXT
LDO
REF
VOUT = 28V
VIN = 1.8V to 28V
1
2
3
4
5
10
9
8
7
6
SYNC/SHDN
VCC
EXT
PGNDREF
GND
FREQ
LDO
MAX668
MAX669
µMAX
TOP VIEW
CS+FB
MAX668/MAX669 1.8V to 28V Input, PWM Step-Up
Controllers in μMAX
Typical Operating Circuit Pin Conguration
Ordering Information
VCC to GND ..........................................................-0.3V to +30V
PGND to GND ....................................................................±0.3V
SYNC/SHDN to GND ............................................-0.3V to +30V
EXT, REF to GND .................................. -0.3V to (VLDO + 0.3V)
LDO, FREQ, FB, CS+ to GND ................................-0.3V to +6V
LDO Output Current ........................................... -1mA to +20mA
REF Output Current ............................................. -1mA to +1mA
LDO Short Circuit to GND ......................................... Momentary
REF Short Circuit to GND .........................................Continuous
Continuous Power Dissipation (TA = +70°C)
10-Pin μMAX (derate 5.6mW/°C above +70°C) .........444mW
Operating Temperature Range ........................... -40°C to +85°C
Junction Temperature ...................................................... +150°C
Storage Temperature Range ............................ -65°C to +150°C
Lead Temperature (soldering,10sec) ...............................+300°C
Soldering Temperature (Reflow) ......................................+300°C
Lead(Pb)-Free Packages ................................................+260°C
Packages Containing Lead(Pb).......................................+240°C
(VCC = VLDO = +5V, ROSC = 200kΩ, TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER CONDITIONS MIN TYP MAX UNITS
PWM CONTROLLER
Input Voltage Range, VCC
MAX668
328 V
MAX669
1.8 28
Input Voltage Range with VCC Tied to LDO 2.7 5.5 V
FB Threshold 1.225 1.250 1.275 V
FB Threshold Load Regulation
Typically 0.013% per mV on CS+;
VCS+ range is 0 to 100mV for 0 to full load
current.
0.013 %/mV
FB Threshold Line Regulation
Typically 0.012% per % duty factor
on
EXT; EXT
duty factor for a step-up is:
100% (1 – VIN/VOUT)
0.012
%/%
FB Input Current VFB =
1.30V
120
nA
Current Limit Threshold 85 100 115 mV
Idle Mode Current-Sense Threshold 5 15 25 mV
CS+ Input Current CS+ forced to
GND
0.2 1 µA
VCC Supply Current (Note 1) VFB =
1.30V,
VCC = 3V to 28V 220 350 µA
Shutdown Supply Current (VCC)SYNC/SHDN = GND, VCC = 28V 3.5 6 µA
REFERENCE AND LDO REGULATORS
LDO Output Voltage LDO load =
∞ to 400Ω
5V ≤ VCC ≤ 28V
(includes LDO dropout) 4.50 5.00 5.50
V
3V ≤ VCC ≤ 28V
(includes LDO dropout) 2.65 5.50
Undervoltage Lockout Threshold Sensed at LDO, falling edge,
hysteresis = 1%, MAX668 only 2.40 2.50 2.60 V
REF Output Voltage No load, CREF = 0.22μF 1.225 1.250 1.275 V
REF Load Regulation REF load = 0 to 50μA -2 -10 mV
REF Undervoltage Lockout Threshold Rising edge, 1% hysteresis 1.0 1.1 1.2 V
OSCILLATOR
Oscillator Frequency
ROSC = 200kΩ ±1% 225 250 275
kHzROSC = 100kΩ ±1% 425 500 575
ROSC = 500kΩ ±1% 85 100 115
MAX668/MAX669 1.8V to 28V Input, PWM Step-Up
Controllers in μMAX
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Absolute Maximum Ratings
Electrical Characteristics
(VCC = VLDO = +5V, ROSC = 200kΩ, TA = -40°C to +85°C, unless otherwise noted.) (Note 2)
(VCC = VLDO = +5V, ROSC = 200kΩ, TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER CONDITIONS MIN TYP MAX UNITS
Maximum Duty Cycle
ROSC = 200kΩ ±1% 87 90 93
%ROSC = 100kΩ ±1% 86 90 94
ROSC = 500kΩ ±1% 86 90 94
Minimum EXT Pulse Width 290 ns
Minimum SYNC Input-Pulse Duty Cycle 20 45 %
Minimum SYNC Input Low Pulse Width 50 200 ns
SYNC Input Rise/Fall Time Not tested 200 ns
SYNC Input Frequency Range 100 500 kHz
SYNC/SHDN Falling Edge to Shutdown
Delay 70 µs
SYNC/SHDN Input High Voltage 3.0V < VCC < 28V 2.0 V
1.8V < VCC < 3.0V (MAX669) 1.5
SYNC/SHDN Input Low Voltage 3.0V < VCC < 28V 0.45 V
1.8V < VCC < 3.0V (MAX669) 0.30
SYNC/SHDN Input Current VSYNC/SHDN = 5V 0.5 3.0 µA
VSYNC/SHDN = 28V 1.5 6.5
EXT Sink/Source Current EXT forced to 2V 1 A
EXT On-Resistance EXT high or low 2 5 Ω
PARAMETER CONDITIONS MIN MAX UNITS
PWM CONTROLLER
Input Voltage Range, VCC
MAX668
328 V
MAX669
1.8 28
Input Voltage Range with VCC Tied to LDO 2.7 5.5 V
FB Threshold 1.22 1.28 V
FB Input Current VFB =
1.30V
20
nA
Current-Limit Threshold 85 115 mV
Idle Mode Current-Sense Threshold 3 27 mV
CS+ Input Current CS+ forced to
GND
1 µA
VCC Supply Current (Note 1) VFB =
1.30V,
VCC = 3V to 28V 350 µA
Shutdown Supply Current (VCC)SYNC/SHDN = GND, VCC = 28V 6 µA
REFERENCE AND LDO REGULATORS
LDO Output Voltage LDO load =
∞ to 400Ω
5V ≤ VCC ≤ 28V
(includes LDO dropout) 4.50 5.50
V
3V ≤ VCC ≤ 28V
(includes LDO dropout) 2.65 5.50
LDO Undervoltage Lockout Threshold Sensed at LDO, falling edge,
hysteresis = 1%, MAX669 only 2.40 2.60 V
MAX668/MAX669 1.8V to 28V Input, PWM Step-Up
Controllers in μMAX
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Electrical Characteristics
Electrical Characteristics (continued)
(VCC = VLDO = +5V, ROSC = 200kΩ, TA = -40°C to +85°C, unless otherwise noted.) (Note 2)
Note 1: This is the VCC current consumed when active but not switching. Does not include gate-drive current.
Note 2: Limits at TA = -40°C are guaranteed by design.
PARAMETER CONDITIONS MIN MAX UNITS
REF Output Voltage No load, CREF = 0.22μF 1.22 1.28 V
REF Load Regulation REF load = 0 to 50μA -10 mV
REF Undervoltage Lockout Threshold Rising edge, 1% hysteresis 1.0 1.2 V
OSCILLATOR
Oscillator Frequency
ROSC = 200kΩ ±1% 222 278
kHz
ROSC = 100kΩ ±1% 425 575
ROSC = 500kΩ ±1% 85 115
Maximum Duty Cycle
ROSC = 200kΩ ±1% 87 93
%
ROSC = 100kΩ ±1% 86 94
ROSC = 500kΩ ±1% 86 94
Minimum SYNC Input-Pulse Duty Cycle 45 %
Minimum SYNC Input Low Pulse Width 200 ns
SYNC Input Rise/Fall Time Not tested 200 ns
SYNC Input Frequency Range 100 500 kHz
SYNC/SHDN Input High Voltage 3.0V < VCC < 28V 2.0 V
1.8V < VCC < 3.0V (MAX669) 1.5
SYNC/SHDN Input Low Voltage 3.0V < VCC < 28V 0.45 V
1.8V < VCC < 3.0V (MAX669) 0.30
SYNC/SHDN Input Current VSYNC/SHDN = 5V 3.0 µA
VSYNC/SHDN = 28V 6.5
EXT On-Resistance EXT high or low 5 Ω
MAX668/MAX669 1.8V to 28V Input, PWM Step-Up
Controllers in μMAX
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Electrical Characteristics (continued)
(Circuits of Figures 2, 3, 4, and 5; TA = +25°C; unless otherwise noted.)
70
1 10,000100010 100
MAX668 EFFICIENCY vs.
LOAD CURRENT (VOUT = 12V)
85
75
95
80
90
MAX668 toc02
LOAD CURRENT (mA)
EFFICIENCY (%)
VIN = 5V
NON-BOOTSTRAPPED
FIGURE 4
R4 = 200kΩ
MAX668 EFFICIENCY vs.
LOAD CURRENT (VOUT = 24V)
MAX668 toc03
LOAD CURRENT (mA)
EFFICIENCY (%)
70
1 10,000100010 100
85
75
95
80
90 VIN = 8V
VIN = 5V
VIN = 12V
NON-BOOTSTRAPPED
FIGURE 4
R4 = 200kΩ
0
1.5
1.0
0.5
2.0
2.5
3.0
0 400300100 200 500 600 700 800 900 1000
MAX669 MINIMUM START-UP VOLTAGE
vs. LOAD CURRENT
MAX668 toc04
LOAD CURRENT (mA)
MINIMUM START-UP VOLTAGE (V)
VOUT = 5V
VOUT = 12V
BOOTSTRAPPED
FIGURE 2
0
400
200
800
600
1000
1200
0 10 155 20 25 30
SUPPLY CURRENT vs.
SUPPLY VOLTAGE
MAX668 toc05
SUPPLY VOLTAGE (V)
SUPPLY CURRENT (µA)
MAX669
MAX668
CURRENT INTO VCC PIN
ROSC = 500kΩ
0
500
1000
2000
2500
3000
3500
4000
0 42 6 8 10 12
NO-LOAD SUPPLY CURRENT vs.
SUPPLY VOLTAGE
MAX668 toc06
SUPPLY VOLTAGE (V)
NO-LOAD SUPPLY CURRENT (µA)
1500
VOUT = 12V
BOOTSTRAPPED
FIGURE 2
R4 = 200kΩ
0
0.5
1.5
2.0
2.5
3.0
3.5
0 105 15 20 25 30
SHUTDOWN CURRENT vs.
SUPPLY VOLTAGE
MAX668 toc07
SUPPLY VOLTAGE (V)
SHUTDOWN CURRENT (µA)
1.0
CURRENT INTO VCC PIN
MAX668
MAX669
50
60
65
70
75
80
85
90
95
1 10 100 1000 10,000
EFFICIENCY vs. LOAD CURRENT
(VOUT = 5V)
MAX668 toc01
LOAD CURRENT (mA)
EFFICIENCY (%)
55
VIN = 3.3V
VIN = 3.6V
VIN = 2V
VIN = 2.7V
BOOTSTRAPPED
FIGURE 3
R4 = 200kΩ
150
190
210
170
250
230
270
290
-40 -20 0 20 40 60 80 100
SUPPLY CURRENT vs.
TEMPERATURE
MAX668 toc08
TEMPERATURE (°C)
SUPPLY CURRENT (µA)
ROSC = 100kΩ
ROSC = 200kΩ
ROSC = 500kΩ
0.1 1 10 20
LDO DROPOUT VOLTAGE vs.
LDO CURRENT
MAX668 toc09
LDO CURRENT (mA)
LDO DROPOUT VOLTAGE (mV)
300
0
50
100
150
200
250
VIN = 3V
VIN = 4.5V
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MAX668/MAX669 1.8V to 28V Input, PWM Step-Up
Controllers in μMAX
Typical Operating Characteristics
(Circuits of Figures 2, 3, 4, and 5; TA = +25°C; unless otherwise noted.)
1.240
1.242
1.243
1.241
1.245
1.246
1.244
1.248
1.249
1.247
1.250
-40 -20 0 20 40 60 80 100
REFERENCE VOLTAGE vs.
TEMPERATURE
MAX668 toc10
TEMPERATURE (°C)
REFERENCE VOLTAGE (V)
VCC = 5V
0
100
150
50
250
300
200
400
450
350
500
0 100 200 300 400 500
SWITCHING FREQUENCY vs. ROSC
MAX668 toc11
ROSC (kΩ)
SWITCHING FREQUENCY (kHz)
VCC = 5V
0
100
300
400
500
600
-40 0-20 20 40 60 80 100
SWITCHING FREQUENCY vs.
TEMPERATURE
MAX668 toc12
TEMPERATURE (°C)
SWITCHING FREQUENCY (kHz)
200
100kΩ
165kΩ
499kΩ
VIN = 5V
100 1000 10,000
EXT RISE/FALL TIME vs.
CAPACITANCE
MAX668 toc13
CAPACITANCE (pF)
EXT RISE/FALL TIME (ns)
60
0
10
20
30
40
50
tR, VCC = 3.3V
tF, VCC = 3.3V
tR, VCC = 5V
tF, VCC = 5V
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MAX668/MAX669 1.8V to 28V Input, PWM Step-Up
Controllers in μMAX
Typical Operating Characteristics (continued)
(Circuits of Figures 2, 3, 4, and 5; TA = +25°C; unless otherwise noted.)
EXITING SHUTDOWN
MAX668 toc14
OUTPUT
VOLTAGE
5V/div
INDUCTOR
CURRENT
2A/div
SHUTDOWN
VOLTAGE
5V/div
MAX668, VIN = 5V, VOUT = 12V, LOAD = 1.0A, ROSC = 100kW,
LOW VOLTAGE, NON-BOOTSTRAPPED
500ms/div
0V
0V
0A
ENTERING SHUTDOWN
MAX668 toc15
OUTPUT
VOLTAGE
5V/div
SHUTDOWN
VOLTAGE
5V/div
MAX668, VIN = 5V, VOUT = 12V, LOAD = 1.0A,
LOW VOLTAGE, NON-BOOTSTRAPPED
200µs/div
0V
0V
HEAVY-LOAD SWITCHING WAVEFORM
MAX668 toc16
VOUT
200mV/div
AC-COUPLED
IL
1A/div
Q1, DRAIN
5V/div
MAX668, VIN = 5V, VOUT = 12V, ILOAD = 1.0A,
LOW VOLTAGE, NON-BOOTSTRAPPED
1µs/div
0V
0A
LIGHT-LOAD SWITCHING WAVEFORM
MAX668 toc17
VOUT
100mV/div
AC-COUPLED
IL
1A/div
Q1, DRAIN
5V/div
MAX668, VIN = 5V, VOUT = 12V, ILOAD = 0.1A,
LOW VOLTAGE, NON-BOOTSTRAPPED
1µs/div
0V
0A
LOAD-TRANSIENT RESPONSE
MAX668 toc18
OUTPUT
VOLTAGE
AC-COUPLED
100mV/div
LOAD
CURRENT
1A/div
MAX668, VIN = 5V, VOUT = 12V, ILOAD = 0.1A TO 1.0A,
LOW VOLTAGE, NON-BOOTSTRAPPED
1ms/div
LINE-TRANSIENT RESPONSE
MAX668 toc19
INPUT
VOLTAGE
5V/div
0V
OUTPUT
VOLTAGE
100mV/div
AC-COUPLED
MAX668, VIN = 5V TO 8V, VOUT = 12V, LOAD = 1.0A,
HIGH VOLTAGE, NON-BOOTSTRAPPED
20ms/div
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MAX668/MAX669 1.8V to 28V Input, PWM Step-Up
Controllers in μMAX
Typical Operating Characteristics (continued)
Detailed Description
The MAX668/MAX669 current-mode PWM controllers
operate in a wide range of DC-DC conversion applications,
including boost, SEPIC, flyback, and isolated output
configurations. Optimum conversion efficiency is maintained
over a wide range of loads by employing both PWM
operation and Maxim’s proprietary Idle Mode control to
minimize operating current at light loads. Other features
include shutdown, adjustable internal operating frequency
or synchronization to an external clock, soft start, adjustable
current limit, and a wide (1.8V to 28V) input range.
MAX668 vs. MAX669 Differences
Differences between the MAX668 and MAX669 relate
to their use in bootstrapped or non-bootstrapped circuits
(Table 1). The MAX668 operates with inputs as low
as 3V and can be connected in either a bootstrapped
or non-bootstrapped (IC powered from input supply or
other source) configuration. When not bootstrapped, the
MAX668 has no restriction on output voltage. When boot-
strapped, the output cannot exceed 28V.
The MAX669 is optimized for low input voltages (down to
1.8V) and requires bootstrapped operation (IC powered
from VOUT) with output voltages no greater than 28V.
Bootstrapping is required because the MAX669 does not
have undervoltage lockout, but instead drives EXT with
an open-loop, 50% duty-cycle start-up oscillator when
LDO is below 2.5V. It switches to closed-loop operation
only when LDO exceeds 2.5V. If a non-bootstrapped
connection is used with the MAX669 and if VCC (the
input voltage) remains below 2.7V, the output voltage will
soar above the regulation point. Table 2 recommends the
appropriate device for each biasing option.
Table 1. MAX668/MAX669 Comparison
PIN NAME FUNCTION
1 LDO 5V On-Chip Regulator Output. This regulator powers all internal circuitry including the EXT gate driver.
Bypass LDO to GND with a 1µF or greater ceramic capacitor.
2FREQ Oscillator Frequency Set Input. A resistor from FREQ to GND sets the oscillator from 100kHz (ROSC =
500kΩ) to 500kHz (ROSC = 100kΩ). fOSC = 5 x 1010 / ROSC. ROSC is still required if an external clock is used
at SYNC/SHDN. (See SYNC/SHDN and FREQ Inputs section.)
3 GND Analog Ground
4 REF 1.25V Reference Output. REF can source 50µA. Bypass to GND with a 0.22µF ceramic capacitor.
5 FB Feedback Input. The FB threshold is 1.25V.
6 CS+ Positive Current-Sense Input. Connect a current-sense resistor, RCS, between CS+ and PGND.
7 PGND Power Ground for EXT Gate Driver and Negative Current-Sense Input
8 EXT External MOSFET Gate-Driver Output. EXT swings from LDO to PGND.
9VCC Input Supply to On-Chip LDO Regulator. VCC accepts inputs up to 28V. Bypass to GND with a 0.1µF ceramic
capacitor.
10 SYNC/
SHDN
Shutdown control and Synchronization Input. There are three operating modes:
• SYNC/SHDN low: DC-DC off.
• SYNC/SHDN high: DC-DC on with oscillator frequency set at FREQ by ROSC.
• SYNC/SHDN clocked: DC-DC on with operating frequency set by SYNC clock input. DC-DC conversion
cycles initiate on rising edge of input clock.
FEATURE MAX668 MAX669
VCC Input
Range 3V to 28V 1.8V to 28V
Operation
Bootstrapped or nonboot-
strapped. VCC can be
connected to input,
output, or other voltage
source such as a logic
supply.
Must be boot-
strapped (VCC must
be connected to
boosted output
voltage, VOUT).
Under-
voltage
Lockout
IC stops switching for
LDO below 2.5V. No
Soft-Start Yes When LDO is
above 2.5V
MAX668/MAX669 1.8V to 28V Input, PWM Step-Up
Controllers in μMAX
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Pin Description
PWM Controller
The heart of the MAX668/MAX669 current-mode PWM
controller is a BiCMOS multi-input comparator that
simultaneously processes the output-error signal, the
current-sense signal, and a slope-compensation ramp
(Figure 1). The main PWM comparator is direct summing,
lacking a traditional error amplifier and its associated
phase shift. The direct summing configuration approaches
ideal cycle-by-cycle control over the output voltage since
there is no conventional error amp in the feedback path.
In PWM mode, the controller uses fixed-frequency,
current-mode operation where the duty ratio is set by the
input/output voltage ratio (duty ratio = (VOUT - VIN)/VIN
in the boost configuration). The current-mode feedback
loop regulates peak inductor current as a function of the
output error signal.
At light loads the controller enters Idle mode. During Idle
mode, switching pulses are provided only as needed to
service the load, and operating current is minimized to
provide best light-load efficiency. The minimum-current
comparator threshold is 15mV, or 15% of the full-load
value (IMAX) of 100mV. When the controller is synchro-
nized to an external clock, Idle Mode occurs only at very
light loads.
Bootstrapped/Non-Bootstrapped Operation
Low-Dropout Regulator (LDO)
Several IC biasing options, including bootstrapped and
non-bootstrapped operation, are made possible by an
on-chip, low-dropout 5V regulator. The regulator input is
at VCC, while its output is at LDO. All MAX668/MAX669
functions, including EXT, are internally powered from
LDO. The VCC-to-LDO dropout voltage is typically 200mV
(300mV max at 12mA), so that when VCC is less than
5.2V, LDO is typically VCC - 200mV. When LDO is in drop-
out, the MAX668/MAX669 still operate with VCC as low
as 3V (as long as LDO exceeds 2.7V), but with reduced
amplitude FET drive at EXT. The maximum VCC input
voltage is 28V.
LDO can supply up to 12mA to power the IC, supply
gate charge through EXT to the external FET, and supply
small external loads. When driving particularly large FETs
at high switching rates, little or no LDO current may be
available for external loads. For example, when switched
at 500kHz, a large FET with 20nC gate charge requires
20nC x 500kHz, or 10mA.
VCC and LDO allow a variety of biasing connections to
optimize efficiency, circuit quiescent current, and full-load
start-up behavior for different input and output voltage
ranges. Connections are shown in Figure 2, Figure 3,
Figure 4, and Figure 5. The characteristics of each are
outlined in Table 1.
Figure 1. MAX668/MAX669 Functional Diagram
ANTISAT MUX LOW-VOLTAGE
START-UP
OSCILLATOR
(MAX669 ONLY)
+A
-A X6
+C
-C X1
+S
-S X1
SLOPE COMPENSATION
S Q
BIAS
OSC OSC
FREQ
SYNC/SHDN
0
1
LDO
PGND
1.25V
REF
EXT
UVLO
VCC
R1
552k
R2
276k
R3
276k
100mV
15mV
IMAX
IMIN
MAIN PWM
COMPARATOR
1.25V
FB
CURRENT SENSE
CS+
MAX668
MAX669
LDO
MAX669 ONLY
R
MAX668/MAX669 1.8V to 28V Input, PWM Step-Up
Controllers in μMAX
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Bootstrapped Operation
With bootstrapped operation, the IC is powered from
the circuit output (VOUT). This improves efficiency when
the input voltage is low, since EXT drives the FET with
a higher gate voltage than would be available from the
low-voltage input. Higher gate voltage reduces the FET
on-resistance, increasing efficiency. Other (undesirable)
characteristics of bootstrapped operation are increased
IC operating power (since it has a higher operating
voltage) and reduced ability to start up with high load
current at low input voltages. If the input voltage range
extends below 2.7V, then bootstrapped operation with the
MAX669 is the only option.
With VCC connected to VOUT, as in Figure 2, EXT voltage
swing is 5V when VCC is 5.2V or more, and VCC - 0.2V
when VCC is less than 5.2V. If the output voltage does
not exceed 5.5V, the on-chip regulator can be disabled
by connecting VCC to LDO (Figure 3). This eliminates the
LDO forward drop and supplies maximum gate drive to
the external FET.
Figure 2. MAX669 High-Voltage Bootstrapped Configuration
Figure 3. MAX669 Low-Voltage Bootstrapped Configuration
MAX669
LDO
CS+
REF
FREQ
V+
SYNC/
SHDN
PGND
FB
GND
N1
EXT
VIN = 1.8V to 12V
C3
C2
C1
C4
R4
R1 R2
R3
C7
D1 C5 C6 C8
3
5
7
6
8
2
4
9
1
10
VOUT = 12V @ 0.5A
L1
MAX669
LDO
CS+
REF
FREQ
VCC
SYNC/
SHDN
PGND
FB
GND
N1
EXT
VIN = 1.8V to 5V
C3
0.221µF
C2
1µF
R4
100k
1%
R1
0.02Ω
R2
75k
1%
R3
24.9k
1%
C7
220pF
D1
MBRS340T3 C4
68µF
10V
C5
68µF
10V
C6
0.1µF
3
5
7
6
8
2
4
9
1
10
VOUT = 5V @ 1A
C1
68µF
10V
L1
4.7µH
FDS6680
IRF7401
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Non-Bootstrapped Operation
With non-bootstrapped operation, the IC is powered from the
input voltage (VIN) or another source, such as a logic sup-
ply. Non-bootstrapped operation (Figure 4) is recommended
(but not required) for input voltages above 5V, since the
EXT amplitude (limited to 5V by LDO) at this voltage range
is no higher than it would be with bootstrapped operation.
Note that non-bootstrapped operation is required if the
output voltage exceeds 28V, since this level is too high to
safely connect to VCC. Also note that only the MAX668
can be used with non-bootstrapped operation.
If the input voltage does not exceed 5.5V, the on-chip
regulator can be disabled by connecting VCC to LDO
(Figure 5). This eliminates the regulator forward drop
and supplies the maximum gate drive to the external
FET for lowest on-resistance. Disabling the regulator also
reduces the non-bootstrapped minimum input voltage
from 3V to 2.7V.
Figure 4. MAX668 High-Voltage Non-Bootstrapped Configuration
Figure 5. MAX668 Low-Voltage Non-Bootstrapped Configuration
MAX668
LDO
CS+
REF
FREQ
VCC
PGND
FB
GND
N1
EXT
VIN = 3V to 12V
C3
0.22µF
C4
1µF
C2
0.1µF
R4
100k
1%
R1
0.02W
R2
218k
1%
R3
24.9k
1%
C7
220pF
D1
MBRS340T3 C5
68µF
20V
C6
68µF
20V
C8
0.1µF
3
5
7
6
8
2
4
9
1
10
VOUT = 12V @ 1A
C1
68µF
20V
L1
4.7µH
FDS6680
SYNC/
SHDN
MAX668
LDO
CS+
REF
FREQ
VCC
SYNC/
SHDN
PGND
FB
GND
N1
EXT
VIN = 2.7V to 5.5V
C3
0.22µF
C2
1µF
R4
100k
1%
R1
0.02W
R2
218k
1%
R3
24.9k
1%
C7
220pF
D1
MBRS340T3 C4
68µF
20V
C5
68µF
20V
C6
0.1µF
3
5
7
6
8
2
4
9
1
10
VOUT = 12V @ 1A
C1
68µF
10V
L1
4.7µH
FDS6680
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In addition to the configurations shown in Table 2, the following
guidelines may help when selecting a configuration:
1) If VIN is ever below 2.7V, VCC must be bootstrapped to
VOUT and the MAX669 must be used. If VOUT never
exceeds 5.5V, LDO may be shorted to VCC and VOUT
to eliminate the dropout voltage of the LDO regulator.
2) If VIN is greater than 3.0V, VCC can be powered from
VIN, rather than from VOUT (non-bootstrapped). This
can save quiescent power consumption, especially
when VOUT is large. If VIN never exceeds 5.5V, LDO
may be shorted to VCC and VIN to eliminate the drop-
out voltage of the LDO regulator.
3) If VIN is in the 3V to 4.5V range (i.e., 1-cell Li-Ion or
3-cell NiMH battery range), bootstrapping VCC from
VOUT, although not required, may increase overall
efficiency by increasing gate drive (and reducing
FET resistance) at the expense of quiescent power
consumption.
4) If VIN always exceeds 4.5V, VCC should be tied to VIN,
since bootstrapping from VOUT does not increase gate
drive from EXT but does increase quiescent power
dissipation.
Table 2. Bootstrapped and Non-Bootstrapped Configurations
* For standard step-up DC-DC circuits (as in Figures 2, 3, 4, and 5), regulation cannot be maintained if VIN exceeds VOUT. SEPIC
and transformer-based circuits do not have this limitation.
CONFIGURATION FIGURE USE
WITH:
INPUT
VOLTAGE
RANGE* (V)
OUTPUT
VOLTAGE
RANGE (V)
COMMENTS
High-Voltage,
Bootstrapped Figure2 MAX669 1.8 to 28 3V to 28
Connect VCC to VOUT. Provides maximum external
FET gate drive for low-voltage (Input <3V) to high-
voltage (output >5.5V) boost circuits. VOUT cannot
exceed 28V.
Low-Voltage,
Bootstrapped Figure3 MAX669 1.8 to 5.5 2.7 to 5.5
Connect VOUT to VCC and LDO. Provides maxi-
mum possible external FET gate drive for low-volt-
age designs, but limits VOUT to 5.5V or less.
High-Voltage,
Non-Bootstrapped Figure4 MAX668 3 to 28 VIN to ∞ Connect VIN to VCC. Provides widest input and out-
put range, but external FET gate drive is reduced for
VIN below 5V.
Low-Voltage,
Non-Bootstrapped Figure5 MAX668 2.7 to 5.5 VIN to ∞
Connect VIN to VCC and LDO. FET gate-drive
amplitude = VIN for logic-supply (input 3V to 5.5V) to
high-voltage (output >5.5V) boost circuits. IC oper-
ating power is less than in Figure 4, since IC current
does not pass through the LDO regulator.
Extra IC supply,
Non-Bootstrapped None MAX668 Not
Restricted VIN to ∞
Connect VCC and LDO to a separate supply
(VBIAS) that powers only the IC. FET gate-drive
amplitude = VBIAS. Input power source (VIN) and
output voltage range (VOUT) are not restricted,
except that VOUT must exceed VIN.
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SYNC/SHDN and FREQ Inputs
The SYNC/SHDN pin provides both external-clock
synchronization (if desired) and shutdown control. When
SYNC/SHDN is low, all IC functions are shut down. A logic
high at SYNC/SHDN selects operation at a frequency set
by ROSC, connected from FREQ to GND. The relationship
between fOSC and ROSC is:
ROSC = 5 x 1010/fOSC
So a 500kHz operating frequency, for example, is set with
ROSC = 100kΩ.
Rising clock edges on SYNC/SHDN are interpreted as
synchronization inputs. If the sync signal is lost while
SYNC/SHDN is high, the internal oscillator takes over at
the end of the last cycle and the frequency is returned
to the rate set by ROSC. If sync is lost with SYNC/
SHDN low, the IC waits for 70μs before shutting down.
This maintains output regulation even with intermittent
sync signals. When an external sync signal is used, Idle
mode switchover at the 15mV current-sense threshold is
disabled so that Idle mode only occurs at very light loads.
Also, ROSC should be set for a frequency 15% below the
SYNC clock rate:
ROSC(SYNC) = 5 x 1010 / (0.85 x fSYNC)
Soft-Start
The MAX668/MAX669 feature a “digital” soft start which is
preset and requires no external capacitor. Upon startup,
the peak inductor increments from 1/5 of the value set by
RCS, to the full current-limit value, in five steps over 1024
cycles of fOSC or fSYNC. For example, with an fOSC of
200kHz, the complete soft-start sequence takes 5ms. See
the Typical Operating Characteristics for a photo of soft-
start operation. Soft-start is implemented: 1) when power
is first applied to the IC, 2) when exiting shutdown with
power already applied, and 3) when exiting undervoltage
lockout. The MAX669’s soft-start sequence does not start
until LDO reaches 2.5V.
Design Procedure
The MAX668/MAX669 can operate in a number of
DCDC converter configurations including step-up, SEPIC
(single-ended primary inductance converter), and flyback.
The following design discussions are limited to step-up,
although SEPIC and flyback examples are shown in the
Application Circuits section.
Setting the Operating Frequency
The MAX668/MAX669 can be set to operate from 100kHz
to 500kHz. Choice of operating frequency will depend on
number of factors:
1) Noise considerations may dictate setting (or synchronizing)
fOSC above or below a certain frequency or band of
frequencies, particularly in RF applications.
2) Higher frequencies allow the use of smaller value
(hence smaller size) inductors and capacitors.
3) Higher frequencies consume more operating power
both to operate the IC and to charge and discharge
the gate of the external FET. This tends to reduce
efficiency at light loads; however, the MAX668/
MAX669’s Idle mode feature substantially increases
light-load efficiency.
4) Higher frequencies may exhibit poorer overall efficiency
due to more transition losses in the FET; however, this
shortcoming can often be nullified by trading some
of the inductor and capacitor size benefits for lower-
resistance components.
The oscillator frequency is set by a resistor, ROSC,
connected from FREQ to GND. ROSC must be connected
whether or not the part is externally synchronized ROSC
is in each case:
ROSC = 5 x 1010 / fOSC
when not using an external clock.
ROSC(SYNC) = 5 x 1010 / (0.85 x fSYNC)
when using an external clock, fSYNC.
Setting the Output Voltage
The output voltage is set by two external resistors (R2
and R3, Figure 2, Figure 3, Figure 4, and Figure 5). First
select a value for R3 in the 10kΩ to 1MΩ range. R2 is
then given by:
R2 = R3 [(VOUT/VREF) – 1]
where VREF is 1.25V.
Determining Inductance Value
For most MAX668/MAX669 boost designs, the inductor
value (LIDEAL) can be derived from the following equation,
which picks the optimum value for stability based on the
MAX668/MAX669’s internally set slope compensation:
LIDEAL = VOUT / (4 x IOUT x fOSC)
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The MAX668/MAX669 allow significant latitude in inductor
selection if LIDEAL is not a convenient value. This may
happen if LIDEAL is a not a standard inductance (such as
10μH, 22μH, etc.), or if LIDEAL is too large to be obtained
with suitable resistance and saturation-current rating in
the desired size. Inductance values smaller than LIDEAL
may be used with no adverse stability effects; however,
the peak-to-peak inductor current (ILPP) will rise as L is
reduced. This has the effect of raising the required ILPK
for a given output power and also requiring larger output
capacitance to maintain a given output ripple. An induc-
tance value larger than LIDEAL may also be used, but
output-filter capacitance must be increased by the same
proportion that L has to LIDEAL. See the Capacitor
Selection section for more information on determining
output filter values.
Due to the MAX668/MAX669’s high switching frequencies,
inductors with any core materials that exhibit low core
loss (ferrite, or equivalent) are recommended for best
efficiency performance.
Determining Peak Inductor Current
The peak inductor current required for a particular output is:
ILPEAK = ILDC + (ILPP / 2)
where ILDC is the average DC input current and ILPP is
the inductor peak-to-peak ripple current. The ILDC and
ILPP terms are determined as follows:
OUT OUT D
LDC IN SW
I (V + V )
I = (V – V )
where VD is the forward voltage drop across the Schottky
rectifier diode (D1), and VSW is the drop across the external
FET, when on.
IN SW OUT D IN
LPP
OSC OUT D
(V – V ) (V + V – V )
I = L x f (V + V )
where L is the inductor value. The saturation rating
of the selected inductor should meet or exceed the
calculated value for ILPEAK, although most coil types can
be operated up to 20% over their saturation rating without
difficulty. In addition to the saturation criteria, the inductor
should have as low a series resistance as possible. For
continuous inductor current, the power loss in the inductor
resistance, PLR, is approximated by:
PLR ≅ (IOUT x VOUT / VIN)2 x RL
where RL is the inductor series resistance.
Once the peak inductor current is selected, the current-
sense resistor (RCS) is determined by:
RCS = 85mV / ILPEAK
For high peak inductor currents (>1A), Kelvin sensing
connections should be used to connect CS+ and PGND
to RCS. PGND and GND should be tied together at the
ground side of RCS.
Power MOSFET Selection
The MAX668/MAX669 drive a wide variety of N-channel
power MOSFETs (NFETs). Since LDO limits the EXT output
gate drive to no more than 5V, a logic-level NFET is
required. Best performance, especially at low input voltag-
es (below 5V), is achieved with low-threshold NFETs that
specify on-resistance with a gate-source voltage (VGS) of
2.7V or less. When selecting an NFET, key parameters
can include:
1) Total gate charge (Qg)
2) Reverse transfer capacitance or charge (CRSS)
3) On-resistance (RDS(ON))
4) Maximum drain-to-source voltage (VDS(MAX))
5) Minimum threshold voltage (VTH(MIN))
At high switching rates, dynamic characteristics (parameters
1 and 2 above) that predict switching losses may have more
impact on efficiency than RDS(ON), which predicts DC losses.
Qg includes all capacitances associated with charging the
gate. In addition, this parameter helps predict the current
needed to drive the gate at the selected operating frequency.
The continuous LDO current for the FET gate is:
IGATE = Qg x fOSC
For example, the MMFT3055L has a typical Q
g
of 7nC
(at VGS = 5V); therefore, the IGATE current at 500kHz is
3.5mA. Use the FET manufacturer’s typical value for Qg in
the above equation, since a maximum value (if supplied) is
usually too conservative to be of use in estimating IGATE.
Diode Selection
The MAX668/MAX669’s high switching frequency
demands a high-speed rectifier. Schottky diodes are
recommended for most applications because of their fast
recovery time and low forward voltage. Ensure that the
diode’s average current rating is adequate using the diode
manufacturer’s data, or approximate it with the following
formula:
= + LPEAK OUT
DIODE OUT
I - I
II
3
Also, the diode reverse breakdown voltage must exceed
VOUT. For high output voltages (50V or above), Schottky
diodes may not be practical because of this voltage
requirement. In these cases, use a high-speed silicon
rectifier with adequate reverse voltage.
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Capacitor Selection
Output Filter Capacitor
The minimum output filter capacitance that ensures stability is:
IDEAL
OUT(MIN) CS IN(MIN) OSC
(7.5V L / L )
C(2 R V f )
×
=
π× ×
where VIN(MIN) is the minimum expected input voltage.
Typically COUT(MIN), though sufficient for stability, will not
be adequate for low output voltage ripple. Since output
ripple in boost DC-DC designs is dominated by capacitor
equivalent series resistance (ESR), a capacitance value 2
or 3 times larger than COUT(MIN) is typically needed. Low-
ESR types must be used. Output ripple due to ESR is:
VRIPPLE(ESR) = ILPEAK x ESRCOUT
Input Capacitor
The input capacitor (CIN) in boost designs reduces the
current peaks drawn from the input supply and reduces
noise injection. The value of CIN is largely determined by
the source impedance of the input supply. High source
impedance requires high input capacitance, particularly
as the input voltage falls. Since step-up DC-DC converters
act as “constant-power” loads to their input supply, input
current rises as input voltage falls. Consequently, in low-
input-voltage designs, increasing CIN and/or lowering its
ESR can add as many as five percentage points to
conversion efficiency. A good starting point is to use the
same capacitance value for CIN as for COUT.
Bypass Capacitors
In addition to CIN and COUT, three ceramic bypass
capacitors are also required with the MAX668/MAX669.
Bypass REF to GND with 0.22μF or more. Bypass LDO
to GND with 1μF or more. And bypass VCC to GND with
0.1μF or more. All bypass capacitors should be located as
close to their respective pins as possible.
Compensation Capacitor
Output ripple voltage due to COUT ESR affects loop
stability by introducing a left half-plane zero. A small
capacitor connected from FB to GND forms a pole with
the feedback resistance that cancels the ESR zero. The
optimum compensation value is:
=
COUT
FB OUT
ESR
C C x (R2 x R3) / (R2 + R3)
where R2 and R3 are the feedback resistors (Figure 2,
Figure 3, Figure 4, and Figure 5). If the calculated value
for CFB results in a non-standard capacitance value,
values from 0.5CFB to 1.5CFB will also provide sufficient
compensation.
Applications Information
Starting Under Load
In non-bootstrapped configurations (Figure 4, and Figure 5),
the MAX668 can start up with any combination of
output load and input voltage at which it can operate
when already started. In other words, there are no special
limitations to start up in non-bootstrapped circuits.
In bootstrapped configurations with the MAX668 or
MAX669, there may be circumstances where full load
current can only be applied after the circuit has
started and the output is near its set value. As the input
voltage drops, this limitation becomes more severe. This
characteristic of all bootstrapped designs occurs when the
MOSFET gate is not fully driven until the output voltage
rises. This is problematic because a heavily loaded output
cannot rise until the MOSFET has low on-resistance. In
such situations, low-threshold FETs (VTH < VIN(MIN))
are the most effective solution. The Typical Operating
Characteristics section shows plots of startup voltage
versus load current for a typical bootstrapped design.
Layout Considerations
Due to high current levels and fast switching waveforms
that radiate noise, proper PC board layout is essential.
Protect sensitive analog grounds by using a star ground
configuration. Minimize ground noise by connecting GND,
PGND, the input bypass-capacitor ground lead, and the
output-filter ground lead to a single point (star ground
configuration). Also, minimize trace lengths to reduce
stray capacitance, trace resistance, and radiated noise.
The trace between the external gain-setting resistors and
the FB pin must be extremely short, as must the trace
between GND and PGND.
Application Circuits
Low-Voltage Boost Circuit
Figure 3 shows the MAX669 operating in a low-voltage
boost application. The MAX669 is configured in the boot-
strapped mode to improve low input voltage performance.
The IRF7401 nMOSFET was selected for Q1 in this appli-
cation because of its very low 0.7V gate threshold voltage
(VGS). This circuit provides a 5V output at greater than 2A
of output current and operates with input voltages as low
as 1.8V. Efficiency is typically in the 85% to 90% range.
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+12V Boost Application
Figure 5 shows the MAX668 operating in a 5V to
12V boost application. This circuit provides output
currents of greater than 1A at a typical efficiency of 92%.
The MAX668 is operated in non-bootstrapped mode to
minimize the input supply current. This achieves
maximum light-load efficiency. If input voltages below
5V are used, the IC should be operated in bootstrapped
mode to achieve best low-voltage performance.
4-Cell to +5V SEPIC Power Supply
Figure 6 shows the MAX668 in a SEPIC (single-ended
primary inductance converter) configuration. This configuration
is useful when the input voltage can be either larger or
smaller than the output voltage, such as when converting
four NiMH, NiCd, or Alkaline cells to a 5V output. The
SEPIC configuration is often a good choice for combined
step-up/step-down applications.
The nMOSFET (Q1) must be selected to withstand a
drain-to-source voltage (VDS) greater than the sum of the
input and output voltages. The coupling capacitor (C2)
must be a low-ESR type to achieve maximum efficiency.
C2 must also be able to handle high ripple currents;
ordinary tantalum capacitors should not be used for high-
current designs.
The circuit in Figure 6 provides greater than 1A output
current at 5V when operating with an input voltage from 3V
to 25V. Efficiency will typically be between 70% and 85%,
depending upon the input voltage and output current.
Isolated +5V to +5V Power Supply
The circuit of Figure 7 provides a 5V isolated output at
400mA from a 5V input power supply. Transformer T1
provides electrical isolation for the forward path of the
converter, while the TLV431 shunt regulator and MOC211
opto-isolator provide an isolated feedback error voltage
for the converter. The output voltage is set by resistors
R2 and R3 such that the mid-point of the divider is 1.24V
(threshold of TLV431). Output voltage can be adjusted
from 1.24V to 6V by selecting the proper ratio for R2 and
R3. For output voltages greater than 6V, substitute the
TL431 for the TLV431, and use 2.5V as the voltage at the
midpoint of the voltage-divider.
Figure 6. MAX668 in SEPIC Configuration
R3
100k
R4
0.02Ω
R1
75k
R2
25k
C4
520pF
VIN
3V to 25V
30V
FDS6680
Q1
L1
CTX5-4
MAX668
SHDNVCC
9 10
LDO
FREQ
D1: MBR5340T3, 3A, 40V SCHOTTKY DIODE
R4: WSL-2512-R020F, 0.02Ω
C3: AVX TPSZ686M020R0150, 68µF, 150mΩ ESR
REF
EXT
CS+
8
6
PGNDGND
73
FB
1
2
4
5
1µF
22µF x 3
@ 35V
C3
68µF x 3
VOUT
5V @ 1A
4.9µH
C2
10µF @ 35V
D1
40V
0.22µF
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Figure 7. Isolated +5V to +5V at 400mA Power Supply
PACKAGE TYPE PACKAGE CODE OUTLINE NO. LAND PATTERN NO.
10 µMAX U10-2 21-0061 90-0330
MAX668
LDO
CS+
FB
SHDN
PGND
FREQREF
VCC
GND
T1
2:1
EXT
VIN = +5V
T1: COILTRONICS CTX03-14232
+5V @ 400mA
+5V RETURN
0.1Ω
R2
301k
1%
510Ω
TLV431
610Ω0.068µF
R3
100k
1%
MBR0540L
MBR0540L
47µH
220µF
10V
0.22µF
1µF
100k
10k
MOC211
IRF7603
220µF
10V
0.1µF
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Package Information
For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a “+”,
“#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing
pertains to the package regardless of RoHS status.
REVISION
NUMBER
REVISION
DATE DESCRIPTION PAGES
CHANGED
2 1/12 Added automotive qualied part and updated lead-free and leaded soldering
temperatures 1, 2
3 6/16 Updated verbiage in Determining Inductance Value section 14
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc. © 2016 Maxim Integrated Products, Inc.
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MAX668/MAX669 1.8V to 28V Input, PWM Step-Up
Controllers in μMAX
Revision History
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