LT1513-2IR#PBF - Linear Technology

LT1513/LT1513-2

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SEPIC Constant- or

Programmable-Current/

Constant-Voltage Battery Charger

The LT

1513 is a 500kHz current mode switching regula-

tor specially configured to create a constant- or program-

mable-current/constant-voltage battery charger. In addition

to the usual voltage feedback node, it has a current sense

feedback circuit for accurately controlling output current

of a flyback or SEPIC (Single-Ended Primary Inductance

Converter) topology charger. These topologies allow the

current sense circuit to be ground referred and completely

separated from the battery itself, simplifying battery switch-

ing and system grounding problems. In addition, these

topologies allow charging even when the input voltage is

lower than the battery voltage. The LT1513 can also drive

a CCFL Royer converter with high efficiency in floating or

grounded mode.

Maximum switch current on the LT1513 is 3A. This allows

battery charging currents up to 2A for a single lithium-ion

cell. Accuracy of 1% in constant-voltage mode is perfect

for lithium battery applications. Charging current can be

easily programmed for all battery types.

DESCRIPTION

■

Charger Input Voltage May Be Higher, Equal to or

Lower Than Battery Voltage

■

Charges Any Number of Cells Up to 20V

■

1% Voltage Accuracy for Rechargeable Lithium

Batteries

■

100mV Current Sense Voltage for High Efficiency

(LT1513)

■

0mV Current Sense Voltage for Easy Current

Programming (LT1513-2)

■

Battery Can Be Directly Grounded

■

500kHz Switching Frequency Minimizes

Inductor Size

■

Charging Current Easily Programmable or Shut Down

FEATURES

INPUT VOLTAGE (V)

CURRENT (A)

2.4

2.2

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4 25

LT1513 • TA02

10 15 20 30

12V

INDUCTOR = 10µH

ACTUAL PROGRAMMED CHARGING CURRENT WILL BE

INDEPENDENT OF INPUT VOLTAGE IF IT DOES NOT

EXCEED VALUES SHOWN

SINGLE Li-Ion CELL

(4.1V)

DOUBLE Li-Ion

CELL (8.2V)

16V

20V

BATTERY

VOLTAGE

Maximum Charging Current

, LTC and LT are registered trademarks of Linear Technology Corporation.

■

Charging of NiCd, NiMH, Lead-Acid or Lithium

Rechargeable Cells

■

Precision Current Limited Power Supply

■

Constant-Voltage/Constant-Current Supply

■

Transducer Excitation

■

Universal Input CCFL Driver

APPLICATIONS

Figure 1. SEPIC Charger with 1.25A Output Current

LT1513

L1A*

L1B*

1.25A

•

TAB4

GND

LT1513 • TA01

SYNC

AND/OR

SHUTDOWN

WALL

ADAPTER

INPUT

S/S

22µF

25V

C2**

4.7µF

0.1µF

†

L1A, L1B ARE TWO 10µH WINDINGS ON A

COMMON CORE: COILTRONICS CTX10-4

CERAMIC MARCON THCR40EIE475Z OR TOKIN 1E475ZY5U-C304

MBRD340 OR MBRS340T3. MBRD340 HAS 5µA TYPICAL

LEAKAGE, MBRS340T3 50

A TYPICAL

0.22µF

39Ω

270ΩR3

0.08Ω

22µF

25V

× 2

†

CHARGE

SHUTDOWN

TYPICAL APPLICATION

LT1513/LT1513-2

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ARBSOLUTEXI T

Supply Voltage ....................................................... 30V

Switch Voltage........................................................ 40V

S/S Pin Voltage....................................................... 30V

FB Pin Voltage (Transient, 10ms) ......................... ±10V

Pin Current .................................................... 10mA

Pin Voltage (Transient, 10ms) ......................... ±10V

Operating Junction Temperature Range

LT1513C............................................... 0°C to 125°C

LT1513I ............................................ –40°C to 125°C

Short Circuit ......................................... 0°C to 150°C

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec)................. 300°C

ELECTRICAL C CHARA TERISTICS

VIN = 5V, VC = 0.6V, VFB = VREF, IFB = 0V, VSW and S/S pins open, unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

REF

FB Reference Voltage Measured at FB Pin 1.233 1.245 1.257 V

= 0.8V ●1.228 1.245 1.262 V

FB Input Current V

= V

REF

300 550 nA

●600 nA

FB Reference Voltage Line Regulation 2.7V ≤ V

≤ 25V, V

= 0.8V ●0.01 0.03 %/V

IREF

Reference Voltage (LT1513) Measured at I

Pin –107 –100 –93 mV

= 0V, V

= 0.8V ●–110 –100 –90 mV

Input Current V

IFB

= V

IREF

(Note 2) ●10 25 35 µA

Reference Voltage Line Regulation 2.7V ≤ V

≤ 25V, V

= 0.8V ●0.01 0.05 %/V

FBVOS

Voltage Offset (LT1513-2) (Note 3) I

VFB

= 60µA (Note 4) ●–7.5 2.5 12.5 mV

Input Current V

IFB

= V

IREF

●–200 –10 0 nA

Source Current V

IREF

= –10mV, V

= 1.2V ●–700 –300 –100 µA

Error Amplifier Transconductance ∆I

= ±25µA 1100 1500 1900 µmho

●700 2300 µmho

Error Amplifier Source Current V

= V

REF

– 150mV, V

= 1.5V ●120 200 350 µA

Error Amplifier Sink Current V

= V

REF

+ 150mV, V

= 1.5V ●1400 2400 µA

PACKAGE/ORDER I FOR ATIO

Consult factory for Military grade parts.

ORDER PART

NUMBER

LT1513CR

LT1513-2CR

LT1513IR

LT1513-2IR

JMAX

= 125°C, θ

= 30°C/W

WITH PACKAGE SOLDERED TO 0.5INCH

COPPER

AREA OVER BACKSIDE GROUND PLANE OR INTERNAL

POWER PLANE, θ

CAN VARY FROM 20°C/W TO

> 40°C/W DEPENDING ON MOUNTING TECHNIQUE

R PACKAGE

7-LEAD PLASTIC DD

FRONT VIEW

TAB

GND

S/S

GND

ORDER PART

NUMBER

LT1513-2CT7

LT1513-2IT7

JMAX

= 125°C, θ

= 50°C/W, θ

= 4°C/W

T7 PACKAGE

7-LEAD TO-220

S/S

GND

FRONT VIEW

LT1513/LT1513-2

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ELECTRICAL C CHARA TERISTICS

VIN = 5V, VC = 0.6V, VFB = VREF, IFB = 0V, VSW and S/S pins open, unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

Error Amplifier Clamp Voltage High Clamp, V

= 1V 1.70 1.95 2.30 V

Low Clamp, V

= 1.5V 0.25 0.40 0.52 V

Error Amplifier Voltage Gain 500 V/V

Pin Threshold Duty Cycle = 0% 0.8 1 1.25 V

f Switching Frequency 2.7V ≤ V

≤ 25V 450 500 550 kHz

0°C ≤ T

≤ 125°C 430 500 580 kHz

< 0°C 400 580 kHz

Maximum Switch Duty Cycle ●85 95 %

Switch Current Limit Blanking Time 130 260 ns

BV Output Switch Breakdown Voltage 0°C ≤ T

≤ 125°C4047V

< 0°C35V

SAT

Output Switch ON Resistance I

= 2A ●0.25 0.45 Ω

LIM

Switch Current Limit Duty Cycle = 50% ●3.0 3.8 5.4 A

Duty Cycle = 80% (Note 1) ●2.6 3.4 5.0 A

∆I

/∆I

Supply Current Increase During Switch ON Time 15 25 mA/A

Control Voltage to Switch Current 4A/V

Transconductance

Minimum Input Voltage ●2.4 2.7 V

Supply Current 2.7V ≤ V

≤ 25V ●4 5.5 mA

Shutdown Supply Current 2.7V ≤ V

≤ 25V, V

S/S

≤ 0.6V, T

≥ 0°C●12 30 µA

< 0°C50µA

Shutdown Threshold 2.7V ≤ V

≤ 25V ●0.6 1.3 2 V

Shutdown Delay ●51225µs

S/S Pin Input Current 0V ≤ V

S/S

≤ 5V ●–10 15 µA

Synchronization Frequency Range ●600 800 kHz

The ● denotes specifications which apply over the full operating

temperature range.

Note 1: For duty cycles (DC) between 50% and 85%, minimum

guaranteed switch current is given by I

LIM

= 1.33 (2.75 – DC).

Note 2: The I

pin is servoed to its regulating state with V

= 0.8V.

Note 3: Consult factory for grade selected parts.

Note 4: The I

pin is sevoed to regulate FB to 1.245V

LT1513/LT1513-2

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TEMPERATURE (°C)

–50

NEGATIVE FEEDBACK INPUT CURRENT (µA)

–30

050 75

LT1513 • G06

–40

–10

–20

–25 25 100 125 150

CHARGING CURRENT (A)

BATTERY VOLTAGE (V)

00.4 0.8 1.2 1.6

1513 G07

2.00.2

00.6 1.0 1.4 1.8

(A) 8.4V BATTERY

CHRG

= 0.5A

(B) 8.4V BATTERY

CHRG

= 1A

CHRG

= 1.5A

= 12VMAXIMUM AVAILABLE

CHARGING CURRENT

WITH 12V INPUT

(A) (B)

(C)

Negative Feedback Input Current

vs Temperature

Output Charging Characteristics

Showing Constant-Current and

Constant-Voltage Operation

TEMPERATURE (°C)

–50

MINIMUM SYNCHRONIZATION VOLTAGE (V

P-P

)

0.5

1.0

1.5

2.0

050

100 150

LT1513 • G04

2.5

3.0

–25 25 75 125

SYNC

= 700kHz

Minimum Peak-to-Peak

Synchronization Voltage vs Temperature

TEMPERATURE (°C)

–50

FEEDBACK INPUT CURRENT (nA)

400

500

600

150

LT1513 • G05

300

200

0050 100

100

800

700

–25 25 75 125

= V

REF

Feedback Input Current

vs Temperature

TYPICAL PERFORMANCE CHARACTERISTICS

Switch Saturation Voltage

vs Switch Current Minimum Input Voltage

vs Temperature

Switch Current Limit

vs Duty Cycle

TEMPERATURE (°C)

–50

1.8

INPUT VOLTAGE (V)

2.0

2.2

2.4

2.6

050

100 150

LT1513 • G03

2.8

3.0

–25 25 75 125

DUTY CYCLE (%)

SWITCH CURRENT LIMIT (A)

20 40 60 80

LT1513 • G02

10010

030 50 70 90

25°C AND

125°C

–55°C

SWITCH CURRENT (A)

SWITCH SATURATION VOLTAGE (V)

0.6

0.8

1.0

3.2

LT1513 • G01

0.4

0.2

0.5

0.7

0.9

0.3

0.1

00.8 1.6 2.4 4.0

2.8

0.4 1.2 2.0 3.6

100°C

150°C

25°C

–55°C

LT1513/LT1513-2

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PIN FUNCTIONS

UUU

(Pin 1): The compensation pin is primarily used for

frequency compensation, but it can also be used for soft

starting and current limiting. It is the output of the error

amplifier and the input of the current comparator. Peak

switch current increases from 0A to 3.6A as the V

voltage

varies from 1V to 1.9V. Current out of the V

pin is about

200µA when the pin is externally clamped below the

internal 1.9V clamp level. Loop frequency compensation

is performed with a capacitor or series RC network from

the V

directly to the ground pin

(avoid ground loops).

FB (Pin 2): The feedback pin is used for positive output

voltage sensing. The R1/R2 voltage divider connected to

FB defines Li-Ion float voltage at full charge, or acts as a

voltage limiter for NiCd or NiMH applications. FB is the

inverting input to the voltage error amplifier. Input bias

current is typically 300nA, so divider current is normally

set to 100µA to swamp out any output voltage errors due

to bias current. The noninverting input of this amplifier is

tied internally to a 1.245V reference. The grounded end of

the output voltage divider should be connected directly to

the LT1513 ground pin (avoid ground loops).

(Pin 3): The current feedback pin is used to sense

charging current. It is the input to a current sense amplifier

that controls charging current when the battery voltage is

below a programmed limit. During constant-current

operation, the LT1513 I

pin regulates at –100mV. Input

resistance of this pin is 5kΩ, so filter resistance (R4,

Figure 1) should be less than 50Ω. The 39Ω, 0.22µF filter

shown in Figure 1 is used to convert the pulsating current

in the sense resistor to a smooth DC current feedback

signal. The LT1513-2 I

pin regulates at 0mV to provide

programmable current limit. The current through R5,

Figure 5, is balanced by the current through R4, program-

ming the maximum voltage across R3.

GND (Pin 4): The ground pin is common to both control

circuitry and switch current. V

, FB and S/S signals must

be Kelvin and connected as close as possible to this pin.

The TAB of the R package should also be connected to the

power ground.

(Pin 5): The switch pin is the collector of the power

switch, carrying up to 3A of current with fast rise and fall

times. Keep the traces on this pin as short as possible to

minimize radiation and voltage spikes. In particular, the

path in Figure 1 which includes SW to C2, D1, C1 and

around to the LT1513 ground pin should be as short as

possible to minimize voltage spikes at switch turn-off.

S/S (Pin 6): This pin can be used for shutdown and/or

synchronization. It is logic level compatible, but can be

tied to V

if desired. It defaults to a high ON state when

floated. A logic low state will shut down the charger to a

micropower state. Driving the S/S pin with a continuous

logic signal of 600kHz to 800kHz will synchronize switch-

ing frequency to the external signal. Shutdown is avoided

in this mode with an internal timer.

(Pin 7): The input supply pin should be bypassed with

a low ESR capacitor located right next to the IC chip. The

grounded end of the capacitor must be connected directly

to the ground plane to which the TAB is connected.

TAB: The TAB on the surface mount R package is electri-

cally connected to the ground pin, but a low inductance

connection must be made to both the TAB and the pin for

proper circuit operation. See suggested PC layout in

Figure 4.

LT1513/LT1513-2

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BLOCK DIAGRAM

–

FBA

S/S

50k*

*REMOVE ON LT1513-2

0.04Ω

–

LT1513 • BD

1.245V

REF

500kHz

OSC

SYNC

SHUTDOWN

DELAY AND RESET LOW DROPOUT

2.3V REG ANTISAT

LOGIC DRIVER

SWITCH

–

≈ 6

COMP

The LT1513 is a current mode switcher. This means that

switch duty cycle is directly controlled by switch current

rather than by output voltage or current. Referring to the

Block Diagram, the switch is turned “on” at the start of each

oscillator cycle. It is turned “off” when switch current

reaches a predetermined level. Control of output voltage

and current is obtained by using the output of a dual

feedback voltage sensing error amplifier to set switch

current trip level. This technique has the advantage of

simplified loop frequency compensation. A low dropout

internal regulator provides a 2.3V supply for all internal

circuitry on the LT1513. This low dropout design allows

input voltage to vary from 2.7V to 25V. A 500kHz oscillator

is the basic clock for all internal timing. It turns “on” the

output switch via the logic and driver circuitry. Special

adaptive antisat circuitry detects onset of saturation in the

power switch and adjusts driver current instantaneously to

limit switch saturation. This minimizes driver dissipation

and provides very rapid turn-off of the switch.

A unique error amplifier design has two inverting inputs

which allow for sensing both output voltage and current. A

1.245V bandgap reference biases the noninverting input.

The first inverting input of the error amplifier is brought out

for positive output voltage sensing. The second inverting

input is driven by a “current” amplifier which is sensing

output current via an external current sense resistor. The

current amplifier is set to a fixed gain of –12.5 which

provides a –100mV current limit sense voltage.

The LT1513-2 option removes the feedback resistors

around the I

amplifier and connects its output to the FB

signal. This provides a ground referenced current sense

voltage suitable for external current programming and

makes amplifier input and output available for external

loop compensation.

The error signal developed at the amplifier output is

brought out externally and is used for frequency compen-

sation. During normal regulator operation this pin sits at a

voltage between 1V (low output current) and 1.9V (high

output current). Switch duty cycle goes to zero if the V

is pulled below the V

pin threshold, placing the LT1513 in

an idle mode.

OPERATION

Figure 2

LT1513/LT1513-2

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APPLICATIONS INFORMATION

WUUU

The LT1513 is an IC battery charger chip specifically opti-

mized to use the SEPIC converter topology. A complete

charger schematic is shown in Figure 1. The SEPIC topology

has unique advantages for battery charging. It will operate

with input voltages above, equal to or below the battery

voltage, has no path for battery discharge when turned off,

and eliminates the snubber losses of flyback designs. It also

has a current sense point that is ground referred and need

not be connected directly to the battery. The two inductors

shown are actually just two identical windings on one

inductor core, although two separate inductors can be used.

A current sense voltage is generated with respect to ground

across R3 in Figure 1. The average current through R3 is

always identical to the current delivered to the battery. The

LT1513 current limit loop will servo the voltage across R3

to –100mV when the battery voltage is below the voltage

limit set by the output divider R1/R2. Constant-current

charging is therefore set at 100mV/R3. R4 and C4 filter the

current signal to deliver a smooth feedback voltage to the I

pin. R1 and R2 form a divider for battery voltage sensing and

set the battery float voltage. The suggested value for R2 is

12.4k. R1 is calculated from:

RRV

BAT

12 1 245

1 245 2 0 3

=+µ

(–.)

.(.)

BAT

= battery float voltage

0.3µA = typical FB pin bias current

A value of 12.4k for R2 sets divider current at 100µA. This is

a constant drain on the battery when power to the charger is

off. If this drain is too high, R2 can be increased to 41.2k,

reducing divider current to 30µA. This introduces an addi-

tional uncorrectable error to the constant voltage float mode

of about ±0.5% as calculated by:

V Error = 0.15 A(R1)(R2)

1.245(R1+R2)

BAT ±µ

±0.15µA = expected variation in FB bias current around the

nominal 0.3µA typical value.

With R2 = 41.2k and R1 = 228k, (V

BAT

= 8.2V), the error due

to variations in bias current would be ±0.42%.

A second option is to disconnect the divider when charger

power is off. This can be done with a small NFET as shown in

Figure 3. D2, C6 and R6 form a peak detector to drive the gate

of the FET to about the same as the battery voltage. If power

is turned off, the gate will drop to 0V and the only drain on the

battery will be the reverse leakage of the catch diode D1. See

Diode Selection for a discussion of diode leakage.

LT1513

L1A

L1B

GND

1513 F03

ADAPTER

INPUT

SCHEMATIC SIMPLIFIED FOR CLARITY

D2 = 1N914, 1N4148 OR EQUIVALENT

470pF R6

470k

Figure 3. Eliminating Divider Current

Maximum Input Voltage

Maximum input voltage for the LT1513 is partly determined

by battery voltage. A SEPIC converter has a maximum

switch voltage equal to input voltage plus output voltage.

The LT1513 has a maximum input voltage of 30V and a

maximum switch voltage of 40V, so this limits maximum

input voltage to 30V, or 40V – V

BAT

, whichever is less.

Shutdown and Synchronization

The dual function S/S pin provides easy shutdown and

synchronization. It is logic level compatible and can be

pulled high or left floating for normal operation. A logic low

on the S/S pin activates shutdown, reducing input supply

current to 12µA. To synchronize switching, drive the S/S pin

between 600kHz and 800kHz.

Inductor Selection

L1A and L1B are normally just two identical windings on one

core, although two separate inductors can be used. A typical

value is 10µH, which gives about 0.5A peak-to-peak induc-

tor current. Lower values will give higher ripple current,

which reduces maximum charging current. 5µH can be used

if charging currents are at least 20% lower than the values

LT1513/LT1513-2

sn1513 1513fas

APPLICATIONS INFORMATION

WUUU

shown in the maximum charging current graph. Higher

inductance values give slightly higher maximum charging

current, but are larger and more expensive. A low loss toroid

core such as Kool Mµ

, Molypermalloy or Metglas

recommended. Series resistance should be less than 0.04Ω

for each winding. “Open core” inductors, such as rods or

barrels are not recommended because they generate large

magnetic fields which may interfere with other electronics

close to the charger.

Input Capacitor

The SEPIC topology has relatively low input ripple current

compared to other topologies and higher harmonics are

especially low. RMS ripple current in the input capacitor is

less than 0.25A with L = 10µH and less than 0.5A with

L = 5µH. A low ESR 22µF, 25V solid tantalum capacitor (AVX

type TPS or Sprague type 593D) is adequate for most

applications with the following caveat. Solid tantalum

capacitors can be destroyed with a very high turn-on surge

current such as would be generated if a low impedance input

source were “hot switched” to the charger input. If this

condition can occur, the input capacitor should have the

highest possible voltage rating, at least twice the surge input

voltage if possible. Consult with the capacitor manufacturer

before a final choice is made. A 4.7µF ceramic capacitor such

as the one used for the coupling capacitor can also be used.

These capacitors do not have a turn-on surge limitation. The

input capacitor must be connected directly to the V

pin and

the ground plane close to the LT1513.

Output Capacitor

It is assumed as a worst case that all the switching output

ripple current from the battery charger could flow in the

output capacitor. This is a desirable situation if it is neces-

sary to have very low switching ripple current in the battery

itself. Ferrite beads or line chokes are often inserted in series

with the battery leads to eliminate high frequency currents

that could create EMI problems. This forces all the ripple

current into the output capacitor. Total RMS current into the

capacitor has a maximum value of about 1A, and this is

handled with the two paralleled 22µF, 25V capacitors shown

in Figure 1. These are AVX type TPS or Sprague type 593D

surface mount solid tantalum units intended for switching

applications. Do not substitute other types without ensuring

that they have adequate ripple current ratings. See Input

Capacitor section for details of surge limitation on solid

tantalum capacitors if the battery may be “hot switched” to

the output of the charger.

Coupling Capacitor

C2 in Figure 1 is the coupling capacitor that allows a SEPIC

converter topology to work with input voltages either higher

or lower than the battery voltage. DC bias on the capacitor is

equal to input voltage. RMS ripple current in the coupling

capacitor has a maximum value of about 1A at full charging

current. A conservative formula to calculate this is:

IIVV

COUP RMS CHRG IN BAT

()

()(.)

()

=+11

(1.1 is a fudge factor to account for inductor ripple current

and other losses)

With I

CHRG

= 1.2A, V

= 15V and V

BAT

= 8.2V, I

COUP

= 1.02A.

The recommended capacitor is a 4.7µF ceramic type from

Marcon or Tokin. These capacitors have extremely low ESR

and high ripple current ratings in a small package. Solid

tantalum units can be substituted if their ripple current rating

is adequate, but typical values will increase to 22µF or more

to meet the ripple current requirements.

Diode Selection

The switching diode should be a Schottky type to minimize

both forward and reverse recovery losses. Average diode

current is the same as output charging current, so this will be

under 2A. A 3A diode is recommended for most applications,

although smaller devices could be used at reduced charging

current.

Maximum diode reverse voltage will be equal to

input voltage plus battery voltage.

Diode reverse leakage current will be of some concern

during charger shutdown. This leakage current is a direct

drain on the battery when the charger is not powered. High

Kool Mµ is a registered trademark of Magnetics, Inc.

Metglas is a registered trademark of AlliedSignal Inc.

LT1513/LT1513-2

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APPLICATIONS INFORMATION

WUUU

current Schottky diodes have relatively high leakage cur-

rents (5µA to 500µA) even at room temperature. The latest

very-low-forward devices have especially high leakage cur-

rents. It has been noted that surface mount versions of some

Schottky diodes have as much as ten times the leakage of

their through-hole counterparts. This may be because a low

forward voltage process is used to reduce power dissipation

in the surface mount package. In any case, check leakage

specifications carefully before making a final choice for the

switching diode. Be aware that diode manufacturers want

to specify a maximum leakage current that is ten times

higher than the typical leakage. It is very difficult to get them

to specify a low leakage current in high volume production.

This is an on going problem for all battery charger circuits

and most customers have to settle for a diode whose typical

leakage is adequate, but theoretically has a worst-case

condition of higher than desired battery drain.

Thermal Considerations

Care should be taken to ensure that worst-case conditions

do not cause excessive die temperatures. Typical thermal

resistance is 30°C/W for the R package but this number will

C1,C3,C5 AND R3

TIED DIRECTLY TO

GROUND PLANE

LT1513 TAB AND GROUND

PIN SOLDERED TO

GROUND PLANE

BAT

C1 C1 D1

R3 C3

2 WINDING

INDUCTOR

L1A L1B

LT1513 • F04

Figure 4. LT1513 Suggested Partial Layout for Critical Thermal and Electrical Paths

vary depending on the mounting technique (copper area,

airflow, etc.).

Average supply current (including driver current) is:

ImA

IN BAT CHRG

=+40 024()( )(.)

Switch power dissipation is given by:

PIRVVV

SW CHRG SW BAT IN BAT

=+()()( )()

()

= Output switch ON resistance

Total power dissipation of the die is equal to supply current

times supply voltage, plus switch power:

D(TOTAL)

= (I

)(V

) + P

For V

= 10V, V

BAT

= 8.2V, I

CHRG

= 1.2A, R

= 0.3Ω,

= 4mA + 24mA = 28mA

= 0.64W

= (10)(0.028) + 0.64 = 0.92W

LT1513/LT1513-2

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APPLICATIONS INFORMATION

WUUU

Programmed Charging Current

LT1513-2 charging current can be programmed with a DC

voltage source or equivalent PWM signal, as shown in

Figure 5. In constant-current mode, I

acts as a virtual

ground. The I

SET

voltage across R5 is balanced by the

voltage across R4 in the ratio R4/R5.

Charging current is given by:

IVRRI

CHARGE ISET FBVOS

=()(/)–45

input current is small and can normally be ignored, but

offset voltage must be considered if operating over a

wide range of program currents. The voltage across R3 at

maximum charge current can be increased to reduce

offset errors at lower charge currents. In Figure 5, I

SET

from 0V to 5V corresponds to an I

CHARGE

of 0A to 1A

+37/–62mA. C4 and R4 smooth the switch current wave-

form. During constant-current operation, the voltage feed-

back network loads the FB pin, which is held at V

REF

by the

amplifier. It is recommended that this load does not

exceed 60µA to maintain a sharp constant voltage to

constant current crossover characteristic. I

CHARGE

can

also be controlled by a PWM input. Assuming the signal is

a CMOS rail-to-rail output with a source impedance of less

than a few hundred ohms, effective I

SET

is V

multiplied

by the PWM ratio. I

CHARGE

has good linearity over the

entire 0% to 100% range.

Voltage Mode Loop Stability

The LT1513 operates in constant-voltage mode during the

final phase of charging lithium-ion and lead-acid batteries.

This feedback loop is stabilized with a series resistor and

capacitor on the V

pin of the chip. Figure 6 shows the

simplified model for the voltage loop. The error amplifier is

modeled as a transconductance stage with g

= 1500µmho

Figure 6. Constant-Voltage Small-Signal Model

–

1500µmho

MODULATOR SECTION

= =

= DC INPUT VOLTAGE

BAT

= DC BATTERY VOLTAGE

4(V

)

+ V

BAT

R1*

71.5k

CAP

≈0.15Ω

EACH

BAT

0.1Ω

C1C1

BATTERY

1513 F06

22µF

EACH

12.5k

1.245V

330k

330Ω

* FOR 8.4V BATTERY. ADJUST VALUE OF R1 FOR ACTUAL BATTERY VOLTAGE

** R

AND C

MODEL PHASE DELAY IN THE MODULATOR

0.1µF

3pF

THIS IS A SIMPLIFIED AC MODEL FOR THE LT1513 IN CONSTANT-

VOLTAGE MODE. RESISTOR AND CAPACITOR NUMBERS

CORRESPOND TO THOSE USED IN FIGURE 1. R

AND C

MODEL

THE PHASE DELAY IN THE MODULATOR. C3 IS 3pF FOR A 10µH

INDUCTOR. IT SHOULD BE SCALED PROPORTIONALLY FOR OTHER

INDUCTOR VALUES (6pF FOR 20µH). THE MODULATOR IS A

TRANSCONDUCTANCE WHOSE GAIN IS A FUNCTION OF INPUT AND

BATTERY VOLTAGE AS SHOWN.

AS SHOWN, THIS LOOP HAS A UNITY-GAIN FREQUENCY OF

ABOUT 250Hz. UNITY-GAIN WILL MOVE OUT TO SEVERAL

KILOHERTZ IF BATTERY RESISTANCE INCREASES TO SEVERAL

OHMS. R5 IS NOT USED IN ALL APPLICATIONS, BUT IT GIVES

BETTER PHASE MARGIN IN CONSTANT-VOLTAGE MODE WITH

HIGH BATTERY RESISTANCE.

Figure 5

0.1µFR3

0.2Ω

1513 F05

L1B

SET

249k R4

10k

LT1513-2

LT1513/LT1513-2

sn1513 1513fas

APPLICATIONS INFORMATION

WUUU

Figure 7. Constant-Current Small-Signal Model

problem, and indeed small signal loop stability can be

excellent even in the presence of subharmonic switching.

The primary issue with subharmonics is the presence of EMI

at frequencies below 500kHz.

Constant-Current Mode Loop Stability

The LT1513 is normally very stable when operating in con-

stant-current mode (see Figure 7), but there are certain con-

ditions which may create instabilities. The combination of

higher value current sense resistors (low programmed charg-

ing current), higher input voltages, and the addition of a loop

compensation resistor (R5) on the V

pin may create an un-

stable current mode loop. (A resistor is sometimes added in

series with C5 to improve loop phase margin when the loop

is operating in voltage mode.) Instability results

because loop gain is too high in the 50kHz to 150kHz region

where excess phase occurs in the current sensing amplifier

and the modulator. The I

FBA

amplifier (gain of –12.5) has a

pole at approximately 150kHz. The modulator section con-

sisting of the current comparator, the power switch and the

magnetics, has a pole at approximately 50kHz when the

coupled inductor value is 10µH. Higher inductance will reduce

the pole frequency proportionally. The design procedure pre-

sented here is to roll off the loop to unity-gain at a frequency

of 25kHz or lower to avoid these excess phase regions.

(from the Electrical Characteristics). Amplifier output resis-

tance is modeled with a 330k resistor. The power stage

(modulator section) of the LT1513 is modeled as a transcon-

ductance whose value is 4(V

)/(V

+ V

BAT

). This is a very

simplified model of the actual power stage, but it is sufficient

when the unity-gain frequency of the loop is low compared

to the switching frequency. The output filter capacitor model

includes its ESR (R

CAP

). A series resistance (R

BAT

) is also

assigned to the battery model.

Analysis of this loop normally shows an extremely stable

system for all conditions, even with 0Ω for R5. The one

condition which can cause reduced phase margin is with a

very large battery resistance (>5Ω), or with the battery

replaced with a resistive load. The addition of R5 gives good

phase margin even under these unusual conditions. R5

should not be increased above 330Ω without checking for

two possible problems. The first is instability in the constant

current region (see Constant-Current Mode Loop Stability),

and the second is subharmonic switching where switch duty

cycle varies from cycle to cycle. This duty cycle instability is

caused by excess switching frequency ripple voltage on the

pin. Normally this ripple is very low because of the

filtering effect of C5, but large values of R5 can allow high

ripple on the V

pin. Normal loop analysis does not show this

–

1500µmho

MODULATOR SECTION

= 4(V1)(V

)

+ V

BAT

1513 F07

1.245V

FBA

VOLTAGE

GAIN = 12

330k

330Ω

0.1µF

10pF

100k

24Ω

0.1Ω

3pF

0.22µF

THIS IS A SIMPLIFIED AC MODEL FOR THE LT1513 IN

CONSTANT-CURRENT MODE. RESISTOR AND CAPACITOR

NUMBERS CORRESPOND TO THOSE USED IN FIGURE 1.

AND C

MODEL THE PHASE DELAY IN THE PowerPath.

C3 IS 3pF FOR A 10µH INDUCTOR. IT SHOULD BE SCALED

PROPORTIONALLY FOR OTHER INDUCTOR VALUES (6pF

FOR 20µH). THE PowerPath IS A TRANSCONDUCTANCE

WHOSE GAIN IS A FUNCTION OF INPUT AND BATTERY

VOLTAGE AS SHOWN.

THE CURRENT AMPLIFIER HAS A FIXED VOLTAGE GAIN OF 12.

ITS PHASE DELAY IS MODELED WITH R

AND C

THE ERROR AMPLIFIER HAS A TRANSCONDUCTANCE OF

1500µmho AND AN INTERNAL OUTPUT SHUNT RESISTANCE OF

330k.

AS SHOWN, THIS LOOP HAS A UNITY-GAIN FREQUENCY OF

ABOUT 27kHz. R5 IS NOT USED IN ALL APPLICATIONS, BUT IT

GIVES BETTER PHASE MARGIN IN CONSTANT VOLTAGE MODE.

LT1513/LT1513-2

sn1513 1513fas

APPLICATIONS INFORMATION

WUUU

The suggested way to control unity loop frequency is to

increase the filter time constant on the I

pin (R4/C4 in

Figures 1 and 7). The filter resistor cannot be arbitrarily

increased because high values will affect charging current

accuracy. Charging current will increase by 1% for each

40Ω increase in R4. There is no inherent limitation on the

value of C4, but if this capacitor is ceramic, it should be an

X7R type to maintain its value over temperature. X7R

dielectric requires a larger footprint.

The formula for calculating the minimum value for the filter

capacitor C4 is:

CRV R

fR V V

IN BAT

43 4 12 1500 5

()()()()( )()

()( )( )

= Highest input voltage

1500µ = Transconductance of error amplifier

(EA)f = Desired unity-gain frequency

BAT

= Battery voltage

For example, assume V

IN(MAX)

= 15V, R3 = 0.4Ω (charging

current set to 0.25A), R4 = 24Ω, R5 = 330Ω and V

BAT

= 8V,

CF40 4 4 15 12 0 0015 330

15 8 1=+=

. ( )( )( )( . )( )

)6.3(25000)(39)( µ

The value for C4 could be reduced to a more manageable size

by increasing R4 to 75Ω and reducing R5 to 300Ω, yielding

0.47µF for C4. The 2% increase in charging current can be

ignored or factored into the value for R3.

More Help

Linear Technology Field Application Engineers have a CAD

spreadsheet program for detailed calculations of circuit

operating conditions. In addition, our Applications Depart-

ment is always ready to lend a helping hand. The LT1371

data sheet may also be helpful. The LT1513 is identical

except for the current amplifier circuitry.

LT1513/LT1513-2

sn1513 1513fas

TYPICAL APPLICATIONS

Lithium-Ion Battery Charger with

Switchable Charge Current

Many battery chemistries require several constant-current

settings during the charging cycle. The circuit shown in

Figure 8 uses the LT1513-2 to provide switchable 1.35A and

0.13A constant-current modes. The circuit is based on a

standard SEPIC battery charger circuit set to a single

lithium-ion cell charge voltage of 4.1V. The LT1513-2 has I

referenced to ground allowing a simple resistor network to

set the charging current values. In constant-current mode,

the I

error amplifier drives the FB pin, increasing charging

current, until I

is balanced by I

IIRI

CHARGE R FBVOS

=()()–

There are several ways to control I

including DAC, PWM or

resistor network as shown here. If the lithium cell requires

precharging, Q1 is turned on, setting a constant current of

0.13A. When charge voltage is reached, Q1 is turned off,

programming the full charge current of 1.35A. As the cell

voltage approaches 4.1V, the voltage sensing network (R1,

R2) starts driving the V

pin, changing the LT1513-2 to

constant-voltage mode. As charging current falls, the output

remains in constant-voltage mode for the remainder of the

charging cycle. When charging is complete, the LT1513-2

can be shut down with the S/S pin.

LT1513-2

L1A

CTX10-4

L1B

GND

1513 F08

S/S

330Ω

36k

4.7k

0.1µF

0.22µF

4.7µFD1

MBRS330T3

22µF

25V

×2

25µF

25V

910Ω

10k

3.3V

0.25Ω

78.7k

0.5%

Li-Ion

RECHARGABLE

CELL

GND

34k

0.5%

CHARGE

SHUTDOWN

PRECHARGE

CHARGE

•

Figure 8. Lithium-Ion Battery Charger

LT1513/LT1513-2

sn1513 1513fas

TYPICAL APPLICATIONS

This Cold Cathode Fluorescent Lamp driver uses a Royer

class self-oscillating sine wave converter to driver a high

voltage lamp with an AC waveform. CCFL Royer converters

have significantly degraded efficiency if they must operate at

low input voltages, and this circuit was designed to handle

input voltages as low as 2.7V. Therefore, the LT1513 is

connected to generate a negative current through L2 that

allows the Royer to operate as if it were connected to a

constant higher voltage input.

The Royer output winding and the bulb are allowed to float in

this circuit. This can yield significantly higher efficiency in

situations where the stray bulb capacitance to surrounding

enclosure is high. To regulate bulb current in Figure 9, Royer

input

current is sensed with R2 and filtered with R3 and C6.

This negative feedback signal is applied to the I

pin of the

LT1513. For more information on this circuit contact the LTC

Applications Department and see Design Note 133. Consid-

erable written application literature on Royer CCFL circuits is

also available from other LTC Application and Design Notes.

2.7V

TO 20V L1

20µH

4.7µF

CERAMIC

47µF

ELECT

20k

10k

330k

PWM DIMMING

(≈1kHz)

10 6

•

4.7nF

1µF

C10

10nF

470Ω

0.082µF

WIMA C4

27pF

3kV

LAMP CURRENT

5.6mA

NEGATIVE VOLTAGE

IS GENERATED HERE

0.25Ω

1513 F09

15V

4, TAB

20µH

CCFL

0.1µF

•

Q2 Q1

C2: TOKIN MULTILAYER CERAMIC

C3: MUST BE A LOW LOSS CAPACITOR, WIMA MKP-20 OR EQUIVALENT

L1, L2: COILTRONICS CTX20-4 (MUST BE SEPARATE INDUCTORS)

Q1, Q2: ZETEX ZTX849 OR FZT849

T1: COILTRONICS CTX110605 (67:1)

LT1513-2

GND

S/S

Figure 9. CCFL Power Supply for Floaing Lamp Configuration Operates on 2.7V

LT1513/LT1513-2

sn1513 1513fas

PACKAGE DESCRIPTION

Dimensions in inches (millimeters) unless otherwise noted.

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

R Package

7-Lead Plastic DD Pak

(LTC DWG # 05-08-1462)

R (DD7) 0396

0.026 – 0.036

(0.660 – 0.914)

0.143+0.012

–0.020

()

3.632+0.305

–0.508

0.040 – 0.060

(1.016 – 1.524) 0.013 – 0.023

(0.330 – 0.584)

0.095 – 0.115

(2.413 – 2.921)

0.004+0.008

–0.004

()

0.102+0.203

–0.102

0.050 ± 0.012

(1.270 ± 0.305)

0.059

(1.499)

TYP

0.045 – 0.055

(1.143 – 1.397)

0.165 – 0.180

(4.191 – 4.572)

0.330 – 0.370

(8.382 – 9.398)

0.060

(1.524)

TYP

0.390 – 0.415

(9.906 – 10.541)

15° TYP

0.300

(7.620)

0.075

(1.905)

0.183

(4.648)

0.060

(1.524)

0.060

(1.524)

0.256

(6.502)

BOTTOM VIEW OF DD PAK

HATCHED AREA IS SOLDER PLATED

COPPER HEAT SINK

T7 Package

7-Lead Plastic TO-220 (Standard)

(LTC DWG # 05-08-1422)

0.040 – 0.060

(1.016 – 1.524)

0.026 – 0.036

(0.660 – 0.914)

(

TO-220

)

(

FORMED

)

1197

0.135 – 0.165

(3.429 – 4.191)

0.700 – 0.728

(17.780 – 18.491)

0.045 – 0.055

(1.143 – 1.397)

0.165 – 0.180

(4.191 – 4.572)

0.095 – 0.115

(2.413 – 2.921)

0.013 – 0.023

(0.330 – 0.584)

0.620

(15.75)

TYP

0.155 – 0.195

(3.937 – 4.953)

0.152 – 0.202

(3.860 – 5.130)

0.260 – 0.320

(6.604 – 8.128)

0.147 – 0.155

(3.734 – 3.937)

DIA

0.390 – 0.415

(9.906 – 10.541)

0.330 – 0.370

(8.382 – 9.398)

0.460 – 0.500

(11.684 – 12.700)

0.570 – 0.620

(14.478 – 15.748)

0.230 – 0.270

(5.842 – 6.858)

LT1513/LT1513-2

sn1513 1513fas

 LINEAR TECHNOLOGY CORPORATION 1996

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

●

(408) 432-1900

FAX: (408) 434-0507

●

TELEX: 499-3977

●

www.linear-tech.com

LT/TP 0198 REV A 4K • PRINTED IN THE USA

PART NUMBER DESCRIPTION COMMENTS

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Main Battery Removed

LTC®1325 Microprocessor Controlled Battery Management System Can Charge, Discharge and Gas Gauge NiCd, NiMH and Pb-Acid Batteries

with Software Charging Profiles

LT1510 1.5A Constant-Current/Constant-Voltage Battery Charger Step-Down Charger for Li-Ion, NiCd and NiMH

LT1511 3.0A Constant-Current/Constant-Voltage Battery Charger Step-Down Charger that Allows Charging During Computer Operation and

with Input Current Limiting Prevents Wall-Adapter Overload

LT1512 SEPIC Constant-Current/Constant-Voltage Battery Charger Step-Up/Step-Down Charger for Up to 1A Current

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