MIC2182YSMTR - Micrel - Product.Network

April 22, 2004 1 M9999-042204

MIC2182 Micrel

MIC2182

High-Efficiency Synchronous Buck Controller

General Description

Micrel’s MIC2182 is a synchronous buck (step-down) switch-

ing regulator controller. An all N-channel synchronous archi-

tecture and powerful output drivers allow up to a 20A output

current capabilty. The PWM and skip-mode control scheme

allows efficiency to exceed 95% over a wide range of load

current, making it ideal for battery powered applications, as

well as high current distributed power supplies.

The MIC2182 operates from a 4.5V to 32V input and can

operate with a maximum duty cycle of 86% for use in low-

dropout conditions. It also features a shutdown mode that

reduces quiescent current to 0.1µA.

The MIC2182 achieves high efficiency over a wide output

current range by automatically switching between PWM and

skip mode. Skip-mode operation enables the converter to

maintain high efficiency at light loads by turning off circuitry

pertaining to PWM operation, reducing the no-load supply

current from 1.6mA to 600µA. The operating mode is inter-

nally selected according to the output load conditions. Skip

mode can be defeated by pulling the PWM pin low which

reduces noise and RF interference.

The MIC2182 is available in a 16-pin SOP (small-outline

package) and SSOP (shrink small-outline package) with an

operating range from –40°C to +85°C.

Typical Application

VDD

BST C6

0.1µF

100k

4.5V to 30V*

OUT

3.3V/4A

GND

2.2nF

1nF

0.1µF

4.7µF

16V

C11

22uf

35V

SD103BWS

Q2*

Si4884

B140 C7

220uf

10V ×2

10µHR2

0.02Ω

Q1*

Si4884

HSD

VSW

LSD

PGND

CSH

VOUT

VREF

SGND

GND

MIC2182-3.3BSM

VIN

EN/UVLO

PWM

COMP

SYNC C13, 1nF

* 30V maximum input voltage limit is due

to standard 30V MOSFET selection.

See “Application Information” section for

5V to 3.3V/10A and other circuits.

4.5V–30V* to 3.3V/4A Converter

Micrel, Inc. • 1849 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com

Features

•4.5V to 32V Input voltage range

•1.25V to 6V Output voltage range

•95% efficiency

•300kHz oscillator frequency

•Current sense blanking

•5Ω impedance MOSFET Drivers

•Drives N-channel MOSFETs

•600µA typical quiescent current (skip-mode)

•Logic controlled micropower shutdown (IQ < 0.1µA)

•Current-mode control

•Cycle-by-cycle current limiting

•Built-in undervoltage protection

•Adjustable undervoltage lockout

•Easily synchronizable

•Precision 1.245V reference output

•0.6% total regulation

•16-pin SOP and SSOP packages

•Frequency foldback overcurrent protection

•Sustained short-circuit protection at any input voltage

•20A output current capability

Applications

•DC power distribution systems

•Notebook and subnotebook computers

•PDAs and mobile communicators

•Wireless modems

•Battery-operated equipment

MIC2182 Micrel

M9999-042204 2 April 22, 2004

Pin Configuration

HSD

VSW

BST

LSD

PWM

COMP

SGND

MIC2182

PGND

VDD

VIN

VOUT

SYNC

EN/UVLO

CSH

Adjustable

16-pin SOP (M)

16-Pin SSOP (SM)

HSD

VSW

BST

LSD

PWM

COMP

SGND

MIC2182-x.x

PGND

VDD

VIN

VOUT

SYNC

EN/UVLO

VREF

CSH

Fixed

16-pin SOP (M)

16-Pin SSOP (SM)

Ordering Information

Part Number Voltage Temperature Range Package Lead Finish

MIC2182BM Adjustable –40°C to +85°C 16-pin narrow SOP Standard

MIC2182-3.3BM 3.3V –40°C to +85°C 16-pin narrow SOP Standard

MIC2182-5.0BM 5.0V –40°C to +85°C 16-pin narrow SOP Standard

MIC2182BSM Adjustable –40°C to +85°C 16-pin narrow SSOP Standard

MIC2182-3.3BSM 3.3V –40°C to +85°C 16-pin narrow SSOP Standard

MIC2182-5.0BSM 5.0V –40°C to +85°C 16-pin narrow SSOP Standard

MIC2182YM Adjustable –40°C to +85°C 16-pin narrow SOP Pb-Free

MIC2182-3.3YM 3.3V –40°C to +85°C 16-pin narrow SOP Pb-Free

MIC2182-5.0YM 5.0V –40°C to +85°C 16-pin narrow SOP Pb-Free

MIC2182YSM Adjustable –40°C to +85°C 16-pin narrow SSOP Pb-Free

MIC2182-3.3YSM 3.3V –40°C to +85°C 16-pin narrow SSOP Pb-Free

MIC2182-5.0YSM 5.0V –40°C to +85°C 16-pin narrow SSOP Pb-Free

April 22, 2004 3 M9999-042204

MIC2182 Micrel

Pin Description

Pin Number Pin Name Pin Function

1 SS Soft-Start (External Component): Connect external capacitor to ground to

reduce inrush current by delaying and slowing the output voltage rise time.

Rise time is controlled by an internal 5µA current source that charges an

external capacitor to VDD.

2 PWM PWM/Skip-Mode Select (Input): Low sets PWM-mode operation. 1nF

capacitor to ground sets automatic PWM/skip-mode selection.

3 COMP Compensation (Output): Internal error amplifier output. Connect to capacitor

or series RC network to compensate the regulator control loop.

4 SGND Small Signal Ground (Return): Route separately from other ground traces to

the (–) terminal of COUT.

5 SYNC Frequency Synchronization (Input): Optional. Connect to external clock

signal to synchronize the oscillator. Leading edge of signal above the

threshold terminates the switching cycle. Connect to SGND if unused.

6 EN/UVLO Enable/Undervoltage Lockout (Input): Low-level signal powers down the

controller. Input below the 2.5V threshold disables switching and functions

as an accurate undervoltage lockout (UVLO). Input below the threshold

forces complete micropower (< 0.1µA) shutdown.

7 (fixed) VREF Reference Voltage (Output): 1.245V output. Requires 0.1µf capacitor to

ground.

7 (adj) FB Feedback (Input): Regulates FB pin to 1.245V. See “Application Information”

for resistor divider calculations.

8 CSH Current-Sense High (Input): Current-limit comparator noninverting input. A

built-in offset of 100mV between CSH and VOUT pins in conjunction with the

current-sense resistor set the current-limit threshold level. This is also the

positive input to the current sense amplifier.

9 VOUT Current-Sense Low (Input): Output voltage feedback input and inverting

input for the current limit comparator and the current sense amplifier.

10 VIN [Battery] Unregulated Input (Input): +4.5V to +32V supply input.

11 VDD 5V Internal Linear-Regulator (Output): VDD is the external MOSFET gate

drive supply voltage and an internal supply bus for the IC. Bypass to SGND

with 4.7µF. VDD can supply up to 5mA for external loads.

12 PGND MOSFET Driver Power Ground (Return): Connects to source of synchro-

nous MOSFET and the (–) terminal of CIN

13 LSD Low-Side Drive (Output): High-current driver output for external synchronous

MOSFET. Voltage swing is between ground and VDD.

14 BST Boost (Input): Provides drive voltage for the high-side MOSFET driver. The

drive voltage is higher than the input voltage by VDD minus a diode drop.

15 VSW Switch (Return): High-side MOSFET driver return.

16 HSD High-Side Drive (Output): High-current driver output for high-side MOSFET.

This node voltage swing is between ground and VIN + 5V – Vdiode drop.

MIC2182 Micrel

M9999-042204 4 April 22, 2004

Electrical Characteristics

VIN = 15V; SS = open; VPWM = 0V; VSHDN = 5V; ILOAD = 0.1A; TA = 25°C, bold values indicate –40°C ≤ TA ≤ +85°C; Note 4; unless

noted

Parameter Condition Min Typ Max Units

MIC2182 [Adjustable], (Note 5)

Feedback Voltage Reference 1.233 1.245 1.257 V

Feedback Voltage Reference 1.220 1.245 1.270 V

Feedback Voltage Reference 4.5V < VIN < 32V, 0 < VCSH – VOUT < 75mV 1.208 1.245 1.282 V

Feedback Bias Current 10 nA

Output Voltage Range 1.25 6 V

Output Voltage Line Regulation VIN = 4.5V to 32V, VCSH – VOUT = 50mV 0.03 %/V

Output Voltage Load Regulation 25mV < (VCSH – VOUT) < 75mV (PWM mode only) 0.5 %

Output Voltage Total Regulation

0mV < (V

CSH

– V

OUT

) < 75mV (full load range) 4.5V < V

< 32V

0.6 %

MIC2182-3.3

Output Voltage 3.267 3.3 3.333 V

Output Voltage 3.234 3.3 3.366 V

Output Voltage 4.5V < VIN < 32V, 0 < VCSH – VOUT < 75mV 3.201 3.3 3.399 V

Output Voltage Line Regulation VIN = 4.5V to 32V, VCSH – VOUT = 50mV 0.03 %/V

Output Voltage Load Regulation 25mV < (VCSH – VOUT) < 75mV (PWM mode only) 0.5 %

Output Voltage Total Regulation

0mV < (V

CSH

– V

OUT

) < 75mV (full load range) 4.5V < V

< 32V

0.8 %

MIC2182-5.0

Output Voltage 4.95 5.0 5.05 V

Output Voltage 4.90 5.0 5.10 V

Output Voltage 6.5V < VIN < 32V, 0 < VCSH – VOUT < 75mV 4.85 5.0 5.150 V

Output Voltage Line Regulation VIN = 6.5V to 32V, VCSH – VOUT = 50mV 0.03 %/V

Output Voltage Load Regulation 25mV < (VCSH – VOUT) < 75mV (PWM mode only) 0.5 %

Output Voltage Total Regulation 0mV < (VCSH – VOUT) < 75mV (full load range)

6.5V < V

< 32V

0.8 %

Input and VDD Supply

PWM Mode VPWM = 0V, excluding external MOSFET gate drive current 1.6 2.5 mA

Skip Mode IL = 0mA, VPWM floating (1nF capacitor to ground) 600 1500 µA

Shutdown Quiescent Current VEN/UVLO = 0V 0.1 5µA

Digital Supply Voltage (VDD)I

L = 0mA to 5mA 4.7 5.3 V

Undervoltage Lockout VDD upper threshold (turn on threshold) 4.2 V

VDD lower threshold (turn off threshold) 4.1 V

Absolute Maximum Ratings (Note 1)

Analog Supply Voltage (VIN) .......................................+34V

Digital Supply Voltage (VDD) .........................................+7V

Driver Supply Voltage (BST)....................................VIN +7V

Sense Voltage (VOUT, CSH) ............................. 7V to –0.3V

Sync Pin Voltage (VSYNC) ................................ 7V to –0.3V

Enable Pin Voltage (VEN/UVLO) ......................................VIN

Power Dissipation (PD)

SOP................................................400mW @ TA= 85°C

SSOP .............................................270mW @ TA= 85°C

Ambient Storage Temperature (TS) ......... –65°C to +150°C

ESD, Note 3

Operating Ratings (Note 2)

Analog Supply Voltage (VIN) ........................ +4.5V to +32V

Ambient Temperature (TA).........................–40°C to +85°C

Junction Temperature (TJ) ....................... –40°C to +125°C

Package Thermal Resistance

SOP (θJA) ..........................................................100°C/W

SSOP (θJA)........................................................150°C/W

April 22, 2004 5 M9999-042204

MIC2182 Micrel

Parameter Condition Min Typ Max Units

Reference Output (Fixed Versions Only)

Reference Voltage 1.220 1.245 1.270 V

Reference Line Regulation 6V < VIN < 32V 1 mV

Reference Load Regulation 0µA < IREF < 100µA2mV

Enable/UVLO

Enable Input Threshold 0.6 1.1 1.6 V

UVLO Threshold 2.2 2.5 2.8 V

Enable Input Current VEN/UVLO = 5V 0.1 5µA

Soft Start

Soft-Start Current VSS = 0V –3.5 –5–6.5 µA

Current Limit

Current-Limit Threshold Voltage VCSH = VOUT 75 100 135 mV

Error Amplifier

Error Sense Amplifier Gain 20

Current Amp

Current Sense Amplifier Gain 2.0

Oscillator Section

Oscillator Frequency 270 300 330 kHz

Maximum Duty Cycle 86 %

Minimum On-Time VOUT = VOUT(nominal) + 200mV 140 250 ns

SYNC Threshold Level 0.7 1.3 1.9 V

SYNC Input Current VSYNC = 5V 0.1 5µA

SYNC Minimum Pulse Width 200 ns

SYNC Capture Range Note 6 330 kHz

Frequency Foldback Threshold measured at VOUT pin 0.75 0.95 1.15 V

Foldback Frequency 60 kHz

Gate Drivers

Rise/Fall Time CL = 3000pF 60 ns

Output Driver Impedance source 5 8.5 Ω

sink 3.5 6Ω

Driver Nonoverlap Time 80 ns

PWM Input

PWM Input Current VPWM = 0V –10 µA

Note 1. Exceeding the absolute maximum rating may damage the device.

Note 2. The device is not guaranteed to function outside its operating rating.

Note 3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.

Note 4. 25°C limits are 100% production tested. Limits over the operating temperature range are guaranteed by design and are not production tested.

Note 5. VIN > 1.3 × VOUT (for the feedback voltage reference and output voltage line and total regulation).

Note 6. See applications information for limitations on the maximum operating frequency.

MIC2182 Micrel

M9999-042204 6 April 22, 2004

0.2

0.4

0.6

0.8

1.0

1.2

1.4

1.6

1.8

2.0

-40 -20 0 20 40 60 80 100120140

CURRENT (mA)

TEMPERATURE (°C)

Quiescent Current

vs. Temperature

PWM

Skip

0.05

0.10

0.15

0.20

-40 -20 0 20 40 60 80 100120140

TEMPERATURE (°C)

-0.50

0.50

1.00

1.50

Quiescent Current

vs. Temperature

UVLO Mode

(mA)

SHUTDOWN

(

CURRENT

(

)

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

0 4 8 121620242832

CURRENT (mA)

INPUT VOLTAGE (V)

Quiescent Current

vs. Supply Voltage

PWM

Skip

-0.5

0.5

1.0

1.5

0.1

0.2

0.3

0.4

0.5

0 4 8 12 16 20 24 28 32

SUPPLY VOLTAGE (V)

Quiescent Current

vs. Supply Voltage

UVLO Mode

(mA)

SHUTDOWN

(µA)

CURRENT (µA)

1.236

1.238

1.240

1.242

1.244

1.246

1.248

1.250

1.252

1.254

1.256

0 4 8 121620242832

REFERENCE VOLTAGE (V)

SUPPLY VOLTAGE (V)

REF

(Fixed Versions)

Line Regulation

1.200

1.210

1.220

1.230

1.240

1.250

1.260

0 200 400 600 800 1000

REFERENCE VOLTAGE (V)

LOAD CURRENT (µA)

REF

(Fixed Versions)

Load Regulation

1.240

1.245

1.250

1.255

1.260

-40 -20 0 20 40 60 80 100120140

REFERENCE VOLTAGE (V)

TEMPERATURE (°C)

REF

(Fixed Versions)

vs. Temperature

4.0

4.2

4.4

4.6

4.8

5.0

0 4 8 121620242832

VDD REGULATOR VOLTAGE (V)

SUPPLY VOLTAGE (V)

VDD

Line Regulation

4.80

4.85

4.90

4.95

5.00

0 5 10 15 20 25

VDD REGULATOR VOLTAGE (V)

LOAD CURRENT (mA)

VDD

Load Regulation

4.82

4.84

4.86

4.88

4.90

4.92

4.94

4.96

4.98

-40 -20 0 20 40 60 80 100120140

REGULATOR VOLTAGE (V)

TEMPERATURE (°C)

vs. Temperature

-10

-8

-6

-4

-2

-40 -20 0 20 40 60 80 100120140

FREQUENCY VARIATION (%)

TEMPERATURE (°C)

Oscillator Frequency

vs. Temperature

Typical Characteristics

-1.0

-0.8

-0.6

-0.4

-0.2

0.2

0.4

0.6

0.8

1.0

0 4 8 121620242832

FREQUENCY VARIATION (%)

SUPPLY VOLTAGE (V)

Oscillator Frequency

vs. Supply Voltage

April 22, 2004 7 M9999-042204

MIC2182 Micrel

4.0

4.2

4.4

4.6

4.8

5.0

-40 -20 0 20 40 60 80 100120140

CURRENT (µA)

TEMPERATURE (°C)

Soft-Start Current

vs. Temperature

0.08

0.09

0.10

0.11

0.12

-40 -20 0 20 40 60 80 100120140

OVERCURRENT THRESHOLD (V)

TEMPERATURE (°C)

Overcurrent Threshold

vs. Temperature

012345678

OUTPUT VOLTAGE (V)

OUTPUT CURRENT (A)

Current-Limit

Foldback

= 5V

OUT

= 3.3V

= 15mΩ

MIC2182 Micrel

M9999-042204 8 April 22, 2004

Block Diagrams

SGND

Gm = 0.2×10-3

Low

Comp

Current

Limit

Skip-Mode

Current

Limit

PWM Mode

to Skip

Mode

Hysteresis

Comp

VBG

1.245V

VBG

0.07V

–2%V

Error

Amp

Control

Logic

VIN

PWM

PWM OUTPUT

CORRECTIVE

RAMP

RESET

Reference

Oscillator

0.024V

EN/UVLO

611

PWM

SYNC

COMP

100k

MIC2182 [adj.]

VOUT

CSH

VDD

VBST

CIN

VIN

HSD Q2

Q1 D1

CBST

COUT

VIN

VOUT

VSW

LSD

PGND

CCOMP

RCOMP

4.7µF

AV = 2

V 1.245V R1

OUT











V6V

OUT(max)

=.0

Current

Sense

Amp

RCS

Figure 2a. Adjustable Output Voltage Version

April 22, 2004 9 M9999-042204

MIC2182 Micrel

Low

Comp

Current

Limit

Skip-Mode

Current

Limit

Hysteresis

Comp

VBG

1.245V

VBG

0.07V

–2%V

Error

Amp

Control

Logic

VIN

PWM

PWM OUTPUT

CORRECTIVE

RAMP

RESET

Reference

Oscillator

0.024V

EN/UVLO

611

PWM

SYNC

COMP VREF

100k Gm = 0.2×10-3

MIC2182-x.x

SGND

50k

R1*

VOUT

CSH

VDD

VBST

CIN

VIN

HSD Q2

Q1 D1

CBST RCS

COUT

VIN

VOUT

VSW

LSD

PGND

CCOMP

RCOMP

4.7µF

*82.5k for 3.3V Output

150k for 5V Output

AV = 2

Current

Sense

Amp

PWM Mode

to Skip

Mode

Figure 2b. Fixed Output Voltage Versions

MIC2182 Micrel

M9999-042204 10 April 22, 2004

Functional Description

See “Applications Information” following this section for com-

ponent selection information and Figure 14 and Tables 1

through 5 for predesigned circuits.

The MIC2182 is a BiCMOS, switched-mode, synchronous

step-down (buck) converter controller. Current-mode control

is used to achieve superior transient line and load regulation.

An internal corrective ramp provides slope compensation for

stable operation above a 50% duty cycle. The controller is

optimized for high-efficiency, high-performance dc-dc con-

verter applications.

The MIC2182 block diagrams are shown in Figure 2a and

Figure 2b.

The MIC2182 controller is divided into 6 functions.

•Control loop

- PWM operation

- Skip-mode operation

•Current limit

•Reference, enable, and UVLO

•MOSFET gate drive

•Oscillator and sync

•Soft start

Control Loop

PWM and Skip Modes of Operation

The MIC2182 operates in PWM (pulse-width-modulation)

mode at heavier output load conditions. At lighter load condi-

tions, the controller can be configured to automatically switch

to a pulse-skipping mode to improve efficiency. The potential

disadvantage of skip mode is the variable switching fre-

quency that accompanies this mode of operation. The occur-

rence of switching pulses depends on component values as

well as line and load conditions. There is an external sync

function that is disabled in skip mode. In PWM mode, the

synchronous buck converter forces continuous current to

flow in the inductor. In skip mode, current through the inductor

can settle to zero, causing voltage ringing across the induc-

tor. Pulling the PWM pin (pin 2) low will force the controller to

operate in PWM mode for all load conditions, which will

improve cross regulation of transformer-coupled, multiple

output configurations.

PWM Control Loop

The MIC2182 uses current-mode control to regulate the

output voltage. This method senses the output voltage (outer

loop) and the inductor current (inner loop). It uses inductor

current and output voltage to determine the duty cycle of the

buck converter. Sampling the inductor current removes the

inductor from the control loop, which simplifies compensa-

tion.

Current

Sense

Amp

1.245V

Error

Amp

VDD

CORRECTIVE

RAMP

PWM

COMPARATOR

RESET

Reference

Oscillator

COMP

100k

MIC2182 [adj.] PWM Mode

VOUT

CSH

VDD

VBST

VIN

HSD Q2

Q1 D1

BST

OUT

VSW

LSD

PGND

COMP

4.7µF

V 1.245V R1

OUT =+











SR Q

CONTROL LOGIC AND

PULSE-WIDTH MODULATOR

= 0.2×10

-3

= 2

0.024V

PWM Mode

to Skip

Mode

LOW

FORCES

SKIP MODE

Figure 3. PWM Operation

April 22, 2004 11 M9999-042204

MIC2182 Micrel

A block diagram of the MIC2182 PWM current-mode control

loop is shown in Figure 3 and the PWM mode voltage and

current waveforms are shown in figure 5A. The inductor

current is sensed by measuring the voltage across the

resistor, RCS. A ramp is added to the amplified current-sense

signal to provide slope compensation, which is required to

prevent unstable operation at duty cycles greater than 50%.

A transconductance amplifier is used for the error amplifier,

which compares an attenuated sample of the output voltage

with a reference voltage. The output of the error amplifier is

the COMP (compensation) pin, which is compared to the

current-sense waveform in the PWM block. When the current

signal becomes greater than the error signal, the comparator

turns off the high-side drive. The COMP pin (pin 3) provides

access to the output of the error amplifier and allows the use

of external components to stabilize the voltage loop.

Skip-Mode Control Loop

This control method is used to improve efficiency at light

output loads. At light output currents, the power drawn by the

MIC2182 is equal to the input voltage times the IC supply

current (IQ). At light output currents, the power dissipated by

the IC can be a significant portion of the total output power,

which lowers the efficiency of the power supply. The MIC2182

draws less supply current in skip mode by disabling portions

of the control and drive circuitry when the IC is not switching.

The disadvantage of this method is greater output voltage

ripple and variable switching frequency.

A block diagram of the MIC2182 skip mode is shown in Figure

4. Skip mode voltage and current waveforms are shown in

figure 5B.

Low

Comp

Skip-Mode

Current

Limit

VBG

1.245V

VBG

0.07V

LOW

FORCES

PWM MODE

–2%V

VIN

Reference

MIC2182 [adj.] Skip Mode

VOUT

CSH

VDD

VBST

CIN

VIN

HSD Q2

CBST

COUT

VIN

VOUT

VSW

LSD

PGND

4.7µF

V 1.245V R1

OUT











Current

Sense

Amp

SR Q

CONTROL LOGIC AND

SKIP-MODE LOGIC

RCS

AV = 2

ONE SHOT

LOW-SIDE DRIVER

ONE SHOT

Hysteresis

Comp

±1%

Figure 4. Skip-Mode Operation

MIC2182 Micrel

M9999-042204 12 April 22, 2004

A hysteretic comparator is used in place of the PWM error

amplifier and a current-limit comparator senses the inductor

current. A one-shot starts the switching cycle by momentarily

turning on the low side MOSFET to insure the high-side drive

boost capacitor, Cbst, is fully charged. The high-side MOS-

FET is turned on and current ramps up in the inductor, L1.

The high-side drive is turned off when either the peak voltage

on the input of the current-sense comparator exceeds the

threshold, typically 35mV, or the output voltage rises above

the hysteretic threshold of the output voltage comparator.

Once the high-side MOSFET is turned off, the load current

discharges the output capacitor, causing VOUT to fall. The

cycle repeats when VOUT falls below the lower threshold, –

1%.

The maximum peak inductor current depends on the skip-

mode current-limit threshold and the value of the current-

sense resistor, RCS.

I35mV

inductor(peak) sense

Figure 6 shows the improvement in efficiency that skip mode

makes when at lower output currents.

100

0.01 0.1 1 10 100

EFFICIENCY (%)

OUTPUT CURRENT (A)

Skip

PWM

Figure 6. Efficiency

VDD

VIN + VDD

VIN

VSW

Reset

Pulse

VHSD

VLSD

ILOAD

IL1

Figure 5a. PWM-Mode Timing

+1%

–1%

NOMINAL

OUT

LSD

HSD

one-shot

LIM(skip)

OUT

Figure 5b. Skip-Mode Timing

April 22, 2004 13 M9999-042204

MIC2182 Micrel

Switching from PWM to Skip Mode

The current sense amplifier in Figure 3 monitors the average

voltage across the current-sense resistor. The controller will

switch from PWM to skip mode when the average voltage

across the current-sense resistor drops below approximately

12mV. This is shown in Figure 7b. The average output current

at this transition level for is calculated below.

I0.012

OUT(skipmode) CS

where:

0.012 = threshold voltage of the internal comparator

RCS = current-sense resistor value

Switching from Skip to PWM Mode

The frequency of occurrence of the skip-mode current pulses

increase as the output current increases until the hysteretic

duty cycle reaches 100% (continuous pulses). Increasing the

current past this point will cause the output voltage will drop.

The low limit comparator senses the output voltage when it

drops below 2% of the set output and automatically switches

the converter to PWM mode.

The inductor current in skip mode is a triangular wave shape

a minimum value of 0 and a maximum value of 35mV/RCS

(see Figure 7b). The maximum average output current in skip

mode is the average value of the inductor waveform:

I 0.5 35mV

OUT(maxskipmode) CS

=×

The capacitor on the PWM pin (pin 2) is discharged when the

IC transitions from skip to PWM mode. This forces the IC to

remain in PWM mode for a fixed period of time. The added

delay prevents unwanted switching between PWM and skip

mode. The capacitor is charged with a 10uA current source

on pin 2. The threshold on pin 2 is 2.5V. The delay for a typical

1nF capacitor is:

tCV

I1nF 2.5V

10 A 250 s

delay PWM threshold

source

=×=×=

µµ

where:

CPWM = capacitor connected to pin 2

Current Limit

The current-limit circuit operates during PWM mode. The

output current is detected by the voltage drop across the

external current-sense resistor (RCS in Figure 2.). The cur-

rent-limit threshold is 100mV+35mV –25mV. The current-

sense resistor must be sized using the minimum current-limit

threshold. The external components must be designed to

withstand the maximum current limit. The current-sense

resistor value is calculated by the equation below:

R75mV

CS OUT(max)

The maximum output current is:

I135mV

OUT(max) CS

The current-sense pins CSH (pin 8) and VOUT (pin 9) are

noise sensitive due to the low signal level and high input

impedance. The PCB traces should be short and routed close

to each other. A small (1nF to 0.1µF) capacitor across the pins

will attenuate high frequency switching noise.

When the peak inductor current exceeds the current-limit

threshold, the current-limit comparator, in Figure 2, turns off

the high-side MOSFET for the remainder of the cycle. The

output voltage drops as additional load current is pulled from

the converter. When the output voltage reaches approxi-

mately 0.95V, the circuit enters frequency-foldback mode

and the oscillator frequency will drop to 60kHz while maintain-

ing the peak inductor current equal to the nominal 100mV

across the external current-sense resistor. This limits the

maximum output power delivered to the load under a short

circuit condition.

Reference, Enable, and UVLO Circuits

The output drivers are enabled when the following conditions

are satisfied:

•The VDD voltage (pin 11) is greater than its

undervoltage threshold (typically 4.2V).

•The voltage on the enable pin is greater than the

enable UVLO threshold (typically 2.5V)

The internal bias circuit generates a 1.245V bandgap refer-

ence voltage for the voltage error amplifier and a 5V VDD

voltage for the gate drive circuit. The reference voltage in the

fixed-output-voltage versions of the MIC2182 is buffered and

brought to pin 7. The VREF pin should be bypassed to GND

(pin 4) with a 0.1µF capacitor. The adjustable version of the

MIC2182 uses pin 7 for output voltage sensing. A decoupling

capacitor on pin 7 is not used in the adjustable output voltage

version.

LIM(skip)

Inductor

Current

35mV THRESHOLD

ACROSS R

Figure 7a. Maximum Skip-Mode-Load Inductor Current

MIN(PWM)

Inductor

Current

12mV THRESHOLD

OF AVERAGE VOLTAGE

ACROSS R

Figure 7b. Minimum PWM-Mode-Load Inductor Current for PWM Operation

MIC2182 Micrel

M9999-042204 14 April 22, 2004

The enable pin (pin 6) has two threshold levels, allowing the

MIC2182 to shut down in a low current mode, or turn off output

switching in UVLO mode. An enable pin voltage lower than

the shutdown threshold turns off all the internal circuitry and

reduces the input current to typically 0.1µA.

If the enable pin voltage is between the shutdown and UVLO

thresholds, the internal bias, VDD, and reference voltages are

turned on. The soft-start pin is forced low by an internal

discharge MOSFET. The output drivers are inhibited from

switching and remain in a low state. Raising the enable

voltage above the UVLO threshold of 2.5V allows the soft-

start capacitor to charge and enables the output drivers.

Either of two UVLO conditions will pull the soft-start capacitor

low.•When the VDD drops below 4.1V

•When the enable pin drops below the 2.5V

threshold

MOSFET Gate Drive

The MIC2182 high-side drive circuit is designed to switch an

N-channel MOSFET. Referring to the block diagram in Figure

2, a bootstrap circuit, consisting of D2 and CBST, supplies

energy to the high-side drive circuit. Capacitor CBST is

charged while the low-side MOSFET is on and the voltage on

the VSW pin (pin 15) is approximately 0V. When the high-side

MOSFET driver is turned on, energy from CBST is used to turn

the MOSFET on. As the MOSFET turns on, the voltage on the

VSW pin increases to approximately VIN. Diode D2 is re-

versed biased and CBST floats high while continuing to keep

the high-side MOSFET on. When the low-side switch is

turned back on, CBST is recharged through D2.

The drive voltage is derived from the internal 5V VDD bias

supply. The nominal low-side gate drive voltage is 5V and the

nominal high-side gate drive voltage is approximately 4.5V

due the voltage drop across D2. A fixed 80ns delay between

the high- and low-side driver transitions is used to prevent

current from simultaneously flowing unimpeded through both

MOSFETs.

Oscillator and Sync

The internal oscillator is free running and requires no external

components. The nominal oscillator frequency is 300kHz. If

the output voltage is below approximately 0.95V, the oscilla-

tor operates in a frequency-foldback mode and the switching

frequency is reduced to 60kHz.

The SYNC input (pin 5) allows the MIC2182 to synchronize

with an external clock signal. The rising edge of the sync

signal generates a reset signal in the oscillator, which turns

off the low-side gate drive output. The high-side drive then

turns on, restarting the switching cycle. The sync signal is

inhibited when the controller operates in skip mode or during

frequency foldback. The sync signal frequency must be

greater than the maximum specified free running frequency

of the MIC2182. If the synchronizing frequency is lower,

double pulsing of the gate drive outputs will occur. When not

used, the sync pin must be connected to ground.

Figure 8 shows the timing between the external sync signal

(trace 2), the low-side drive (trace 1) and the high-side drive

(trace R1). There is a delay of approximately 250ns between

the rising edge of the external sync signal and turnoff of the

low-side MOSFET gate drive.

Some concerns of operating at higher frequencies are:

•Higher power dissipation in the internal VDD

regulator. This occurs because the MOSFET

gates require charge to turn on the device. The

average current required by the MOSFET gate

increases with switching frequency. This in-

creases the power dissipated by the internal

VDD regulator. Figure 10 shows the total gate

charge which can be driven by the MIC2182

over the input voltage range, for different values

of switching frequency. The total gate charge

includes both the high- and low-side MOSFETs.

The larger SOP package is capable of dissipat-

ing more power than the SSOP package and

can drive larger MOSFETs with higher gate

drive requirements.

TIME

SYNC

SIGNAL LOW-SIDE

DRIVE HIGH-SIDE

DRIVE

Figure 8. Sync Waveforms

TIME

VSS VOUT

Figure 9. Startup Waveforms

April 22, 2004 15 M9999-042204

MIC2182 Micrel

•Reduced maximum duty cycle due to switching

transition times and constant delay times in the

controller. As the switching frequency increased,

the switching period decreases. The switching

transition times and constant delays in the

MIC2182 start to become noticeable. The effect

is to reduce the maximum duty cycle of the

controller. This will cause the minimum input to

output differential voltage (dropout voltage) to

increase.

100

0 4 8 12 16 20 24 28 32

GATE CHARGE (nC)

SUPPLY VOLTAGE (V)

400kHz

300kHz

500kHz

SOP

Figure 10a. SOP Gate Charge vs. Input Voltage

100

0 4 8 121620242832

GATE CHARGE (nC)

SUPPLY VOLTAGE (V)

SSOP

400kHz

300kHz

500kHz

Figure 10b. SSOP Gate Charge vs. Input Voltage

It is recommended that the user limits the maximum synchro-

nized frequency to 600kHz. If a higher synchronized fre-

quency is required, it may be possible and will be design

dependent. Please consult Micrel applications for assis-

tance.

Soft Start

Soft start reduces the power supply input surge current at

startup by controlling the output voltage rise time. The input

surge appears while the output capacitance is charged up. A

slower output rise time will draw a lower input surge current.

Soft start may also be used for power supply sequencing.

The soft-start voltage is applied directly to the PWM compara-

tor. A 5uA internal current source is used to charge up the

soft-start capacitor. The capacitor is discharged when either

the enable voltage drops below the UVLO threshold (2.5V) or

the VDD voltage drops below the UVLO level (4.1V).

The part switches at a minimum duty cycle when the soft-start

pin voltage is less than 0.4V. This maintains a charge on the

bootstrap capacitor and insures high-side gate drive voltage.

As the soft-start voltage rises above 0.4V, the duty cycle

increases from the minimum duty cycle to the operating duty

cycle. The oscillator runs at the foldback frequency of 60kHz

until the output voltage rises above 0.95V. Above 0.95V, the

switching frequency increases to 300kHz (or the sync’d

frequency), causing the output voltage to rise a greater rate.

The rise time of the output is dependent on the soft-start

capacitor, output capacitance, output voltage, and load cur-

rent. The oscilloscope photo in Figure 9 show the output

voltage and the soft-start pin voltage at startup.

Minimum Pulse Width

The MIC2182 has a specified minimum pulse width. This

minimum pulse width places a lower limit on the minimum

duty cycle of the buck converter. When the MIC2182 is

operating in forced PWM mode (pin 2 low) and when the

output current is very low or zero, there is a limit on the ratio

of VOUT/VIN. If this limit is exceeded, the output voltage will

rise above the regulated voltage level. A minimum load is

required to prevent the output from rising up. This will not

occur for output voltages greater than 3V.

Figure 11 should be used as a guide when the MIC2182 is

forced into PWM-only mode. The actual maximum input

voltage will depend on the exact external components used

(MOSFETs, inductors, etc.).

0123456

INPUT VOLTAGE (V)

OUTPUT VOLTAGE (V)

Figure 11. Max. Input Voltage in Forced-PWM Mode

This restriction does not occur when the MIC2182 is set to

automatic mode (pin 2 connected to a capacitor) since the

converter operates in skip mode at low output current.

MIC2182 Micrel

M9999-042204 16 April 22, 2004

Applications Information

The following applications information includes component

selection and design guidelines. See Figure 14 and Tables 1

through 5 for predesigned circuits.

Inductor Selection

Values for inductance, peak, and RMS currents are required

to select the output inductor. The input and output voltages

and the inductance value determine the peak to peak inductor

ripple current. Generally, higher inductance values are used

with higher input voltages. Larger peak to peak ripple currents

will increase the power dissipation in the inductor and

MOSFETs. Larger output ripple currents will also require

more output capacitance to smooth out the larger ripple

current. Smaller peak to peak ripple currents require a larger

inductance value and therefore a larger and more expensive

inductor. A good compromise between size, loss and cost is

to set the inductor ripple current to be equal to 20% of the

maximum output current.

The inductance value is calculated by the equation below.

LV(V V)

V f 0.2 I

OUT IN(max) OUT

IN(max) S OUT(max)

=×−

×× ×

where:

fS = switching frequency

0.2 = ratio of ac ripple current to dc output current

VIN(max) = maximum input voltage

The peak-to-peak inductor current (ac ripple current) is:

IV(V V)

VfL

PP OUT IN(max) OUT

IN(max) S

=×−

××

The peak inductor current is equal to the average output

current plus one half of the peak to peak inductor ripple

current.

I I 0.5 I

PK OUT(max) PP

=+×

The RMS inductor current is used to calculate the I2·R losses

in the inductor.

I (rms) I 1 1

inductor OUT(max) PP

OUT(max)

=×+











Maximizing efficiency requires the proper selection of core

material and minimizing the winding resistance. The high

frequency operation of the MIC2182 requires the use of ferrite

materials for all but the most cost sensitive applications.

Lower cost iron powder cores may be used but the increase

in core loss will reduce the efficiency of the power supply. This

is especially noticeable at low output power. The winding

resistance decreases efficiency at the higher output current

levels. The winding resistance must be minimized although

this usually comes at the expense of a larger inductor.

The power dissipated in the inductor is equal to the sum of the

core and copper losses. At higher output loads, the core

losses are usually insignificant and can be ignored. At lower

output currents, the core losses can be a significant contribu-

tor. Core loss information is usually available from the mag-

netics vendor.

Copper loss in the inductor is calculated by the equation

below:

P I (rms) R

inductorCu inductor 2winding

=×

The resistance of the copper wire, Rwinding, increases with

temperature. The value of the winding resistance used should

be at the operating temperature.

R R 1 0.0042 (T T )

winding(hot) winding(20 C) hot 20 C

=×+×−

()

°°

where:

THOT = temperature of the wire

under operating load

T20°C = ambient temperature

Rwinding(20°C) is room temperature winding resistance

(usually specified by the manufacturer)

Current-Sense Resistor Selection

Low inductance power resistors, such as metal film resistors

should be used. Most resistor manufacturers make low

inductance resistors with low temperature coefficients, de-

signed specifically for current-sense applications. Both resis-

tance and power dissipation must be calculated before the

resistor is selected. The value of RSENSE is chosen based on

the maximum output current and the maximum threshold

level. The power dissipated is based on the maximum peak

output current at the minimum overcurrent threshold limit.

R75mV

SENSE OUT(max)

The maximum overcurrent threshold is:

I135mV

overcurrent(max) CS

The maximum power dissipated in the sense resistor is:

PI R

D(R ) overcurrent(max)2CS

SENSE =×

MOSFET Selection

External N-channel logic-level power MOSFETs must be

used for the high- and low-side switches. The MOSFET gate-

to-source drive voltage of the MIC2182 is regulated by an

internal 5V VDD regulator. Logic-level MOSFETs, whose

operation is specified at VGS = 4.5V must be used.

It is important to note the on-resistance of a MOSFET

increases with increasing temperature. A 75°C rise in junc-

tion temperature will increase the channel resistance of the

MOSFET by 50% to 75% of the resistance specified at 25°C.

This change in resistance must be accounted for when

calculating MOSFET power dissipation.

Total gate charge is the charge required to turn the MOSFET

on and off under specified operating conditions (VDS and

VGS). The gate charge is supplied by the MIC2182 gate drive

circuit. At 300kHz switching frequency and above, the gate

April 22, 2004 17 M9999-042204

MIC2182 Micrel

charge can be a significant source of power dissipation in the

MIC2182. At low output load this power dissipation is notice-

able as a reduction in efficiency. The average current re-

quired to drive the high-side MOSFET is:

IQf

G[high-side](avg) G S

=×

where:

IG[high-side](avg) =

average high-side MOSFET gate current

QG = total gate charge for the high-side MOSFET

taken from manufacturer’s data sheet

with VGS = 5V.

The low-side MOSFET is turned on and off at VDS = 0

because the freewheeling diode is conducting during this

time. The switching losses for the low-side MOSFET is

usually negligable. Also, the gate drive current for the low-

side MOSFET is more accurately calculated using CISS at

VDS = 0 instead of gate charge.

For the low-side MOSFET:

ICVf

G[low-side](avg) ISS GS S

=××

Since the current from the gate drive comes from the input

voltage, the power dissipated in the MIC2182 due to gate

drive is:

PVI I

gatedrive IN G[high-side](avg) G[low-side](avg)

()

A convenient figure of merit for switching MOSFETs is the on-

resistance times the total gate charge (RDS(on) × QG). Lower

numbers translate into higher efficiency. Low gate-charge

logic-level MOSFETs are a good choice for use with the

MIC2182. Power dissipation in the MIC2182 package limits

the maximum gate drive current. Refer to Figure 10 for the

MIC2182 gate drive limits.

Parameters that are important to MOSFET switch selection

are:•Voltage rating

•On-resistance

•Total gate charge

The voltage rating of the MOSFETs are essentially equal to

the input voltage. A safety factor of 20% should be added to

the VDS(max) of the MOSFETs to account for voltage spikes

due to circuit parasitics.

The power dissipated in the switching transistor is the sum of

the conduction losses during the on-time (Pconduction) and the

switching losses that occur during the period of time when the

MOSFETs turn on and off (PAC).

PP P

SW conduction AC

where:

P I (rms) R

conduction SW 2SW

=×

PP P

AC AC(off) AC(on)

RSW = on-resistance of the MOSFET switch.

Making the assumption the turn-on and turnoff transition

times are equal, the transition time can be approximated by:

tCVC V

TISS GS OSS IN

=×+ ×

where:

CISS and COSS are measured at VDS = 0.

IG = gate drive current (1A for the MIC2182)

The total high-side MOSFET switching loss is:

P(VV)Itf

AC IN D PK T S

=+×××

where:

tT = switching transition time

(typically 20ns to 50ns)

VD = freewheeling diode drop, typically 0.5V.

fS it the switching frequency, nominally 300kHz

The low-side MOSFET switching losses are negligible and

can be ignored for these calculations.

RMS Current and MOSFET Power Dissipation Calculation

Under normal operation, the high-side MOSFET’s RMS

current is greatest when VIN is low (maximum duty cycle). The

low-side MOSFET’s RMS current is greatest when VIN is high

(minimum duty cycle). However, the maximum stress the

MOSFETs see occurs during short circuit conditions, where

the output current is equal to Iovercurrent(max). (See the Sense

Resistor section). The calculations below are for normal

operation. To calculate the stress under short circuit condi-

tions, substitute Iovercurrent(max) for IOUT(max). Use the formula

below to calculate D under short circuit conditions.

D 0.063 1.8 10 V

shortcircuit 3IN

=−××

−

The RMS value of the high-side switch current is:

I (rms) D I I12

SW(highside) OUT(max)2PP2

=× +











I (rms) 1 D I I12

SW(lowside) OUT(max)2PP2

=−

()











where:

D = duty cycle of the converter

DVV

OUT

=×η

η = efficiency of the converter.

Converter efficiency depends on component parameters,

which have not yet been selected. For design purposes, an

efficiency of 90% can be used for VIN less than 10V and 85%

can be used for VIN greater than 10V. The efficiency can be

more accurately calculated once the design is complete. If the

assumed efficiency is grossly inaccurate, a second iteration

through the design procedure can be made.

For the high-side switch, the maximum dc power dissipation

is:

P R I (rms)

switch1(dc) DS(on)1 SW1 2

=×

MIC2182 Micrel

M9999-042204 18 April 22, 2004

For the low-side switch (N-channel MOSFET), the dc power

dissipation is:

P R I (rms)

switch2(dc) DS(on)2 SW2 2

=×

Since the ac switching losses for the low side MOSFET is

near zero, the total power dissipation is:

low-side MOSFET(max) switch2(dc)

The total power dissipation for the high-side MOSFET is:

PPP

highsideMOSFET(max) SWITCH1(dc) AC

External Schottky Diode

An external freewheeling diode is used to keep the inductor

current flow continuous while both MOSFETs are turned off.

This dead time prevents current from flowing unimpeded

through both MOSFETs and is typically 80ns The diode

conducts twice during each switching cycle. Although the

average current through this diode is small, the diode must be

able to handle the peak current.

I I 2 80ns f

D(avg) OUT S

=×××

The reverse voltage requirement of the diode is:

V (rrm) V

diode IN

The power dissipated by the Schottky diode is:

PI V

diode D(avg) F

=×

where:

VF = forward voltage at the peak diode current

The external Schottky diode, D2, is not necessary for circuit

operation since the low-side MOSFET contains a parasitic

body diode. The external diode will improve efficiency and

decrease high frequency noise. If the MOSFET body diode is

used, it must be rated to handle the peak and average current.

The body diode has a relatively slow reverse recovery time

and a relatively high forward voltage drop. The power lost in

the diode is proportional to the forward voltage drop of the

diode. As the high-side MOSFET starts to turn on, the body

diode becomes a short circuit for the reverse recovery period,

dissipating additional power. The diode recovery and the

circuit inductance will cause ringing during the high-side

MOSFET turn-on.

An external Schottky diode conducts at a lower forward

voltage preventing the body diode in the MOSFET from

turning on. The lower forward voltage drop dissipates less

power than the body diode. The lack of a reverse recovery

mechanism in a Schottky diode causes less ringing and less

power loss. Depending on the circuit components and oper-

ating conditions, an external Schottky diode will give a 1/2%

to 1% improvement in efficiency. Figure 12 illustrates the

difference in noise on the VSW pin with and without a

Schottky diode.

Output Capacitor Selection

The output capacitor values are usually determined by the

capacitors ESR (equivalent series resistance). Voltage rating

and RMS current capability are two other important factors in

selecting the output capacitor. Recommended capacitors are

tantalum, low-ESR aluminum electrolytics, and OS-CON.

The output capacitor’s ESR is usually the main cause of

output ripple. The maximum value of ESR is calculated by:

ESR OUT

≤∆

where:

VOUT = peak to peak output voltage ripple

IPP = peak to peak inductor ripple current

The total output ripple is a combination of the ESR and the

output capacitance. The total ripple is calculated below:

∆VI(1D)

CfIR

OUT PP

OUT S

PP ESR 2

=×−









+×

()

where:

D = duty cycle

COUT = output capacitance value

fS = switching frequency

The voltage rating of capacitor should be twice the output

voltage for a tantalum and 20% greater for an aluminum

electrolytic or OS-CON.

The output capacitor RMS current is calculated below:

I (rms) I12

CPP

OUT =

The power dissipated in the output capacitor is:

P I (rms) R

DISS(C C 2ESR(C )

OUT OUT OUT

)

=×

Input Capacitor Selection

The input capacitor should be selected for ripple current

rating and voltage rating. Tantalum input capacitors may fail

when subjected to high inrush currents, caused by turning the

input supply on. Tantalum input capacitor voltage rating

should be at least 2 times the maximum input voltage to

maximize reliability. Aluminum electrolytic, OS-CON, and

multilayer polymer film capacitors can handle the higher

inrush currents without voltage derating.

TIME

WITH

FREEWHEELING DIODE WITHOUT

FREEWHEELING DIODE

Figure 12. Switch Output Noise

With and Without Shottky Diode

April 22, 2004 19 M9999-042204

MIC2182 Micrel

The input voltage ripple will primarily depend on the input

capacitors ESR. The peak input current is equal to the peak

inductor current, so:

∆VI R

IN inductor(peak) ESR(C )

=×

The input capacitor must be rated for the input current ripple.

The RMS value of input capacitor current is determined at the

maximum output current. Assuming the peak to peak induc-

tor ripple current is low:

I (rms) I D (1 D)

C OUT(max)

≈××−

The power dissipated in the input capacitor is:

P I (rms) R

DISS(C ) C 2ESR(C )

IN IN IN

=×

Voltage Setting Components

The MIC2182-3.3 and MIC2182-5.0 ICs contain internal

voltage dividers that set the output voltage. The MIC2182

adjustable version requires two resistors to set the output

voltage as shown in Figure 13.

Error

Amp

MIC2182 [adj.]

REF

1.245V

Figure 13. Voltage-Divider Configuration

The output voltage is determined by the equation:

VV 1

OREF

=×+











Where: VREF for the MIC2182 is typically 1.245V.

A typical value of R1 can be between 3k and 10k. If R1 is too

large it may allow noise to be introduced into the voltage

feedback loop. If R1 is too small in value it will decrease the

efficiency of the power supply, especially at low output loads.

Once R1 is selected, R2 can be calculated using:

R2 VR1

REF

OREF

=×

−

Voltage Divider Power Dissipation

The reference voltage and R2 set the current through the

voltage divider.

IVR2

divider REF

The power dissipated by the divider resistors is:

P (R1 R2) I

divider divider2

=+×

Efficiency Calculation and Considerations

Efficiency is the ratio of output power to input power. The

difference is dissipated as heat in the buck converter. Under

light output load, the significant contributors are:

•Supply current to the MIC2182

•MOSFET gate-charge power (included in the IC

supply current)

•Core losses in the output inductor

To maximize efficiency at light loads:

•Use a low gate-charge MOSFET or use the

smallest MOSFET, which is still adequate for

maximum output current.

•Allow the MIC2182 to run in skip mode at lower

currents.

•Use a ferrite material for the inductor core, which

has less core loss than an MPP or iron power

core.

Under heavy output loads the significant contributors to

power loss are (in approximate order of magnitude):

•Resistive on-time losses in the MOSFETs

•Switching transition losses in the MOSFETs

•Inductor resistive losses

•Current-sense resistor losses

•Input capacitor resistive losses (due to the

capacitors ESR)

To minimize power loss under heavy loads:

•Use logic-level, low on-resistance MOSFETs.

Multiplying the gate charge by the on-resistance

gives a Figure of merit, providing a good bal-

ance between low and high load efficiency.

•Slow transition times and oscillations on the

voltage and current waveforms dissipate more

power during turn-on and turnoff of the

MOSFETs. A clean layout will minimize parasitic

inductance and capacitance in the gate drive

and high current paths. This will allow the fastest

transition times and waveforms without oscilla-

tions. Low gate-charge MOSFETs will transition

faster than those with higher gate-charge

requirements.

•For the same size inductor, a lower value will

have fewer turns and therefore, lower winding

resistance. However, using too small of a value

will require more output capacitors to filter the

output ripple, which will force a smaller band-

width, slower transient response and possible

instability under certain conditions.

•Lowering the current-sense resistor value will

decrease the power dissipated in the resistor.

However, it will also increase the overcurrent

limit and will require larger MOSFETs and

inductor components.

•Use low-ESR input capacitors to minimize the

power dissipated in the capacitors ESR.

Decoupling Capacitor Selection

The 4.7µF decoupling capacitor is used to minimize noise on

the VDD pin. The placement of this capacitor is critical to the

proper operation of the IC. It must be placed right next to the

MIC2182 Micrel

M9999-042204 20 April 22, 2004

pins and routed with a wide trace. The capacitor should be a

good quality tantalum. An additional 1µF ceramic capacitor

may be necessary when driving large MOSFETs with high

gate capacitance. Incorrect placement of the VDD decoupling

capacitor will cause jitter or oscillations in the switching

waveform and large variations in the overcurrent limit.

A 0.1µF ceramic capacitor is required to decouple the VIN.

The capacitor should be placed near the IC and connected

directly to between pin 10 (Vcc) and pin 12 (PGND).

PCB Layout and Checklist

PCB layout is critical to achieve reliable, stable and efficient

performance. A ground plane is required to control EMI and

minimize the inductance in power, signal and return paths.

The following guidelines should be followed to insure proper

operation of the circuit.

•Signal and power grounds should be kept

separate and connected at only one location.

Large currents or high di/dt signals that occur

when the MOSFETs turn on and off must be

kept away from the small signal connections.

•The connection between the current-sense

resistor and the MIC2182 current-sense inputs

(pin 8 and 9) should have separate traces,

routed from the terminals directly to the IC pins.

The traces should be routed as closely as

possible to each other and their length should be

minimized. Avoid running the traces under the

inductor and other switching components. A 1nF

to 0.1µF capacitor placed between pins 8 and 9

will help attenuate switching noise on the current

sense traces. This capacitor should be placed

close to pins 8 and 9.

•When the high-side MOSFET is switched on, the

critical flow of current is from the input capacitor

through the MOSFET, inductor, sense resistor,

output capacitor, and back to the input capacitor.

These paths must be made with short, wide

pieces of trace. It is good practice to locate the

ground terminals of the input and output capaci-

tors close to each.

•When the low-side MOSFET is switched on,

current flows through the inductor, sense

resistor, output capacitor, and MOSFET. The

source of the low-side MOSFET should be

located close to the output capacitor.

•The freewheeling diode, D1 in Figure 2, con-

ducts current during the dead time, when both

MOSFETs are off. The anode of the diode

should be located close to the output capacitor

ground terminal and the cathode should be

located close to the input side of the inductor.

•The 4.7µF capacitor, which connects to the VDD

terminal (pin 11) must be located right at the IC.

The VDD terminal is very noise sensitive and

placement of this capacitor is very critical.

Connections must be made with wide trace. The

capacitor may be located on the bottom layer of

the board and connected to the IC with multiple

vias.

•The VIN bypass capacitor should be located

close to the IC and connected between pins 10

and 12. Connections should be made with a

ground and power plane or with short, wide

trace.

April 22, 2004 21 M9999-042204

MIC2182 Micrel

VDD

BST C6

0.1µF

100k

OUT

GND

2.2nF

1nF

0.1µF

50V

4.7µF

16V

C11

(table)

SD103BWS

(table)

(table) C7

(table) C12

0.1µF

50V

(table) R2

(table)

HSD

VSW

LSD

PGND

CSH

VOUT

VREF

SGND

GND

MIC2182

VIN

EN/UVLO

PWM

COMP

SYNC C13, 1nF

Figure 14. Basic Circuit Diagram for Use with Tables 3 through 6

noitacificepStimiL

elppirycneuqerfgnihctiwSegatlovtuptuofo%1

erutarepmettneibmamumixaM58°C

ytilibapactiucric-trohSsuounitnoC

ycneuqerfgnihctiwSzHk003

Table 1. Specifications for Figure 14 and Tables 3 through 6

Manufacturer Telephone Number (USA) Web Address

AVX (803) 946-0690 www.avxcorp.com

Central Semiconductor (516) 435-1110 www.centralsemi.com

Coiltronics (561) 241-7876 www.coiltronics.com

IRC (704) 264-8861

IR (310) 322-3331 www.irf.com

Micrel (408) 944-0800 www.micrel.com

Vishay/Lite On (805) 446-4800 www.vishay-liteon.com

(diodes)

Vishay/Siliconix (800) 554-5665 www.siliconix.com

(MOSFETs)

Vishay/Dale (800) 487-9437 www.vishaytechno.com

(inductors and resistors)

Sumida (847) 956-0666 www.japanlink.com/sumida

Table 2. Component Suppliers

Predesigned Circuits

A single schematic diagram, shown in Figure 14, can be used

to build power supplies ranging from 3A to 10A at the common

output voltages of 1.8V, 2.5V, 3.3V, and 5V. Components that

vary, depending upon output current and voltage, are listed

in the accompanying Tables 3 through 6.

Power supplies larger than 10A can also be constructed

using the MIC2182 using larger power-handling compo-

nents.

The “Power Supply Operating Characteristics” graphs follow-

ing the component and vendor tables provide useful informa-

tion about the actual performance of some of these circuits.

MIC2182 Micrel

M9999-042204 22 April 22, 2004

3A (6.5V–30V) 4A (6.5V–30V) 5A (6.5V–30V) 10A (6.5V–10V)

Reference Part No. / Description Part No. / Description Part No. / Description Part No. / Description

C7 qty: 2 qty: 2 qty: 2 qty: 2

TPSE227M010R0100 TPSE227M010R0100 TPSV227M010R0060 TPSV337M010R0060

AVX, 220µF 10V, AVX, 220µF 10V, AVX, 220µF 10V, AVX, 330µF 10V,

0.1Ω ESR, 0.1Ω ESR, 0.06Ω ESR, 0.06Ω ESR,

output filter capacitor output filter capacitor output filter capacitor output filter capacitor

C11 qty: 2 qty: 3 qty: 4 qty: 4

TPSE226M035R0300 TPSE226M035R0300 TPSE226M035R0300 TPSV107M020R0085

AVX, 22µF 35V, AVX, 22µF 35V, AVX, 22µF 35V, AVX, 100µF 20V,

0.3Ω ESR, 0.3Ω ESR, 0.3Ω ESR, 0.06Ω ESR,

input filter capacitor input filter capacitor input filter capacitor input filter capacitor

D1 qty: 1 B140, Vishay, qty: 1 B140, Vishay, qty: 1 B140, Vishay, qty: 1 B330, Vishay,

freewheeling diode freewheeling diode freewheeling diode freewheeling diode

L1 qty: 1 CDRH125-100, qty: 1 CDRH127-100, qty: 1 CDRH127-100 qty: 1 UP4B-3R3,

Sumida Inductor, Sumida Inductor, Sumida, Coiltronics,

10µH 4A, 10µH 5A, 10µH 5A, 3.3µH 11A,

output inductor output inductor output inductor output inductor

Q1 qty: 1 Si4800, Siliconix, qty: 1 Si4800, Siliconix, qty: 1 Si4884, Siliconix, qty: 2 Si4884, Siliconix

low-side MOSFET low-side MOSFET low-side MOSFET low-side MOSFET

Q2 qty: 1 Si4800, Siliconix, qty: 1 Si4800, Siliconix, qty: 1 Si4884, Siliconix, qty: 2 Si4884, Siliconix,

high-side MOSFET high-side MOSFET high-side MOSFET high-side MOSFET

R2 qty: 1 qty: 1 qty: 1 qty: 2

WSL-2010 .025 1%, WSL-2010 .020 1%, WSL-2512 .015 1%, WSL-2512 .015 1% ,

Vishay, 0.025, 1%, 0.5W, Vishay, 0.02, 1%, 0.5W, Vishay, 0.015, 1%, 1W, Vishay, 0.015, 1%, 1W,

current sense resistor current sense resistor current sense resistor current sense resistor

U1 MIC2182-5.0BSM or MIC2182-5.0BSM or MIC2182-5.0BSM or MIC2182-5.0BM

MIC2182-5.0BM MIC2182-5.0BM MIC2182-5.0BM

Table 3. Components for 5V Output

3A (4.5V–30V) 4A (4.5V–30V) 5A (4.5V–30V) 10A (4.5V–5.5V)

Reference Part No. / Description Part No. / Description Part No. / Description Part No. / Description

C7 qty: 2 qty: 2 qty: 2 qty: 2

TPSE227M010R0100 TPSE227M010R0100 TPSV227M010R0060 TPSV477M006R0055

AVX, 220µF 10V, AVX, 220µF 10V, AVX, 220µF 10V, AVX, 470µF 6.3V,

0.1Ω ESR, 0.1Ω ESR, 0.06Ω ESR, 0.055Ω ESR,

output filter capacitor output filter capacitor output filter capacitor output filter capacitor

C11 qty: 2 qty: 2 qty: 3 qty: 3

TPSE226M035R0300 TPSE226M035R0300 TPSE226M035R0300 TPSV227M016R0075

AVX, 22µF 35V, AVX, 22µF 35V, AVX, 22µF 35V, AVX, 220µF 16V,

0.3Ω ESR, 0.3Ω ESR, 0.3Ω ESR, 0.075Ω ESR,

input filter capacitor input filter capacitor input filter capacitor filter capacitor

D1 qty: 1 B140, Vishay, qty: 1 B140, Vishay, qty: 1 B140, Vishay, qty: 1 B330, Vishay,

freewheeling diode freewheeling diode freewheeling diode freewheeling diode

L1 qty: 1 CDRH125-100, qty: 1 CDRH127-100, qty: 1 CDRH127-100 qty: 1 UP4B-3R3,

Sumida Inductor, Sumida Inductor, Sumida, Coiltronics,

10µH 4A, 10µH 5A, 10µH 5A, 3.3µH 11A,

output inductor output inductor output inductor output inductor

Q1 qty: 1 Si4800, Siliconix, qty: 1 Si4800, Siliconix, qty: 1 Si4800, Siliconix, qty: 2 Si4884, Siliconix,

low-side MOSFET low-side MOSFET low-side MOSFET low-side MOSFET

Q2 qty: 1 Si4800, Siliconix, qty: 1 Si4800, Siliconix, qty: 1 Si4884, Siliconix, qty: 2 Si4884, Siliconix,

high-side MOSFET high-side MOSFET high-side MOSFET high-side MOSFET

R2 qty: 1 qty: 1 qty: 1 qty: 2

WSL-2010 .025 1%, WSL-2010 .020 1%, WSL-2512 .015 1%, WSL-2512 .015 1% ,

Vishay, 0.025, 1%, 0.5W, Vishay, 0.02, 1%, 0.5W, Vishay, 0.015, 1%, 1W, Vishay, 0.015, 1%, 1W,

current sense resistor current sense resistor current sense resistor current sense resistor

U1 MIC2182-3.3BSM or MIC2182-3.3BM or MIC2182-3.3BM or MIC2182-3.3BM

MIC2182-3.3BM MIC2182-3.3BSM MIC2182-3.3BSM

Table 4. Components for 3.3V Output

April 22, 2004 23 M9999-042204

MIC2182 Micrel

3A (4.5V–30V) 4A (4.5V–30V) 5A (4.5V–30V) 10A (4.5V–5.5V)

Reference Part No. / Description Part No. / Description Part No. / Description Part No. / Description

C7 qty: 2 qty: 2 qty: 2 qty: 2

TPSE227M010R0100 TPSE227M010R0100 TPSV227M010R0060 TPSV447M006R0055

AVX, 220µF 10V, AVX, 220µF 10V, AVX, 220µF 10V, AVX, 470µF 6.3V,

0.1Ω ESR, 0.1Ω ESR, 0.06Ω ESR, 0.06Ω ESR,

output filter capacitor output filter capacitor output filter capacitor output filter capacitor

C11 qty: 2 qty: 2 qty: 2 qty: 3

TPSE226M035R0300 TPSE226M035R0300 TPSE226M035R0300 TPSV227M016R0075

AVX, 22µF 35V, AVX, 22µF 35V, AVX, 22µF 35V, AVX, 220µF 16V,

0.3Ω ESR, 0.3Ω ESR, 0.3Ω ESR, 0.06Ω ESR,

input filter capacitor input filter capacitor input filter capacitor input filter capacitor

D1 qty: 1 B140, Vishay, qty: 1 B140, Vishay, qty: 1 B140, Vishay, qty: 1 B330, Vishay,

freewheeling diode freewheeling diode freewheeling diode freewheeling diode

L1 qty: 1 CDRH125-100, qty: 1 CDRH127-100, qty: 1 CDRH127-100 qty: 1 UP4B-3R3,

Sumida Inductor, Sumida Inductor, Sumida, Coiltronics,

10µH 4A, 10µH 5A, 10µH 5A, 3.3µH 11A,

output inductor output inductor output inductor output inductor

Q1 qty: 1 Si4800, Siliconix, qty: 1 Si4884, Siliconix, qty: 1 Si4884, Siliconix, qty: 2 Si4884, Siliconix

low-side MOSFET low-side MOSFET low-side MOSFET low-side MOSFET

Q2 qty: 1 Si4800, Siliconix, qty: 1 Si4800, Siliconix, qty: 1 Si4800, Siliconix, qty: 2 Si4884, Siliconix,

high-side MOSFET high-side MOSFET high-side MOSFET high-side MOSFET

R2 qty: 1 qty: 1 qty: 1 qty: 1

WSL-2010 .025 1%, WSL-2010 .020 1%, WSL-2512 .015 1%, WSL-2512 .015 1% ,

Vishay, 0.025, 1%, 0.5W, Vishay, 0.02, 1%, 0.5W, Vishay, 0.015, 1%, 1W, Vishay, 0.015, 1%, 1W,

current sense resistor current sense resistor current sense resistor current sense resistor

U1 MIC2182BSM or MIC2182BSM or MIC2182BSM or MIC2182BM

MIC2182BM MIC2182BM MIC2182BM

Table 5. Components for 2.5V Output

3A (4.5V–30V) 4A (4.5V–30V) 5A (4.5V–8V) 10A (4.5V–5.5V)

Reference Part No. / Description Part No. / Description Part No. / Description Part No. / Description

C7 qty: 2 qty: 2 qty: 2 qty: 2

TPSE227M010R0100 TPSE227M010R0100 TPSV227M010R0060 TPSV447M006R0055

AVX, 220µF 10V, AVX, 220µF 10V, AVX, 220µF 10V, AVX, 470µF 6.3V,

0.1Ω ESR, 0.1Ω ESR, 0.06Ω ESR, 0.06Ω ESR,

output filter capacitor output filter capacitor output filter capacitor output filter capacitor

C11 qty: 2 qty: 2 qty: 2 qty: 2

TPSE226M035R0300 TPSE226M035R0300 TPSE226M035R0300 TPSV227M016R0075

AVX, 22µF 35V, AVX, 22µF 35V, AVX, 22µF 35V, AVX, 220µF 16V,

0.3Ω ESR, 0.3Ω ESR, 0.3Ω ESR, 0.06Ω ESR,

input filter capacitor input filter capacitor input filter capacitor input filter capacitor

D1 qty: 1 B140, Vishay, qty: 1 B140, Vishay, qty: 1 B140, Vishay, qty: 1 B330, Vishay,

freewheeling diode freewheeling diode freewheeling diode freewheeling diode

L1 qty: 1 CDRH125-100, qty: 1 CDRH127-100, qty: 1 CDRH127-100 qty: 1 UP4B-3R3,

Sumida Inductor, Sumida Inductor, Sumida, Coiltronics,

10µH 4A, 10µH 5A, 10µH 5A, 3.3µH 11A,

output inductor output inductor output inductor output inductor

Q1 qty: 1 Si4800, Siliconix, qty: 1 Si4884, Siliconix, qty: 1 Si4884, Siliconix, qty: 2 Si4884, Siliconix

low-side MOSFET low-side MOSFET low-side MOSFET low-side MOSFET

Q2 qty: 1 Si4800, Siliconix, qty: 1 Si4800, Siliconix, qty: 1 Si4800, Siliconix, qty: 2 Si4884, Siliconix,

high-side MOSFET high-side MOSFET high-side MOSFET high-side MOSFET

R2 qty: 1 qty: 1 qty: 1 qty: 2

WSL-2010 .025 1%, WSL-2010 .020 1%, WSL-2512 .015 1%, WSL-2512 .015 1% ,

Vishay, 0.025, 1%, 0.5W, Vishay, 0.02, 1%, 0.5W, Vishay, 0.015, 1%, 1W, Vishay, 0.015, 1%, 1W,

current sense resistor current sense resistor current sense resistor current sense resistor

U1 MIC2182BSM or MIC2182BSM or MIC2182BSM or MIC2182BM

MIC2182BM MIC2182BM MIC2182BM

Table 6. Components for 1.8V Output

MIC2182 Micrel

M9999-042204 24 April 22, 2004

Effect of Soft-Start Capacitor (CSS) V alue

On Output Voltage Waveforms

During T urn-On

(10A Power Supply Configuration)

Effect of Soft-Start Capacitor (CSS) V alue

On Output Voltage Waveforms

During T urn-On

(4A Power Supply Configuration)

Normal (300kHz Switching Frequency) and

Output Short-Circuit (60kHz) Conditions

Switch Node (Pin 15) Waveforms

Converter Waveforms

QTY: 2

Si4884

HIGH-SIDE

MOSFETS

SWITCH-NODE

VOLTAGE

INDUCTOR CURRENT

LOW-SIDE MOSFET

GATE-TO-SOURCE V OLTAGE

HIGH-SIDE MOSFET

GATE-TO-SOURCE V OLTAGE

QTY: 2

Si4884

LOW-SIDE

MOSFETS

VIN = 7V

L1 = 3.3µH

VOUT = 3.3V

IOUT = 10A

VSW

PIN 15

VSW+HSD

PIN 16

VGS

LOW-SIDE

MOSFET

VGS

HIGH-SIDE

MOSFET

IL1

(2A/div)

HIGH-SIDE

DRIVE VOLTAGE

REFERENCED TO GROUND

10Amps

Typical Skip-Mode Waveforms

VOUT

VSW

Pin 15

IL1

(0.5A/div)

Typical PWM-Mode Waveforms

VOUT

VSW

Pin 15

IL1

(0.5A/div)

Power Supply Operating Characteristics

April 22, 2004 25 M9999-042204

MIC2182 Micrel

-40

-20

100

120

150

180

210

10x10

100x10

1x10

10x10

100x10

300x10

GAIN (dB)

PHASE (°)

FREQUENCY (Hz)

Bode Plot

(4A Power Supply Configuration)

GAIN

PHASE

Load T ransient Response

and Bode Plot

(4A Power Supply Configuration)

VIN = 12V

VOUT = 3.3V

L1 = 10µH

R2 = 20mΩ

IOUT

2A/div VOUT

Load T ransient Response

and Bode Plot

(10A Power Supply Configuration)

VIN = 6V

VOUT = 3.3V

L1 = 3.3µH

R2 = 7.5mΩ

IOUT

5A/div VOUT

-40

-20

100

120

150

180

210

10x10

100x10

1x10

10x10

100x10

300x10

GAIN (dB)

PHASE (°)

FREQUENCY (Hz)

Bode Plot

(10A Power Supply Configuration)

GAIN

PHASE

100

0.01 0.1 1 10

EFFICIENCY (%)

OUTPUT CURRENT (A)

Skip

PWM

R2 = 7.5mΩ

L1 = 3.3

2 high-side MOSFETs: Si4884

2 low-side MOSFETs: Si4884

Efficiency

(10A Power Supply Configuration)

100

0.01 0.1 1 4

EFFICIENCY (%)

OUTPUT CURRENT (A)

Skip PWM

VIN = 5V

R2 = 15mΩ

L1 = 10

1 high-side MOSFET: Si4800

1 low-side MOSFET: Si4800

5V Efficiency

(4A Power Supply Configuration)

100

0.01 0.1 1 4

EFFICIENCY (%)

OUTPUT CURRENT (A)

12V Efficiency

(4A Power Supply Configuration)

Skip PWM

= 12V

R2 = 15mΩ

L1 = 10

1 high-side MOSFET: Si4800

1 low-side MOSFET: Si4800

100

0.01 0.1 1 4

EFFICIENCY (%)

OUTPUT CURRENT (A)

Skip PWM

VIN = 24V

R2 = 15mΩ

L1 = 10

1 high-side MOSFET: Si4800

1 low-side MOSFET: Si4800

24V Efficiency

(4A Power Supply Configuration)

MIC2182 Micrel

M9999-042204 26 April 22, 2004

Package Information

45°0°–8°

0.244 (6.20)

0.228 (5.79)

0.394 (10.00)

0.386 (9.80) SEATING

PLANE

0.020 (0.51)

REF 0.020 (0.51)

0.013 (0.33)

0.157 (3.99)

0.150 (3.81)

0.050 (1.27)

0.016 (0.40)

0.0648 (1.646)

0.0434 (1.102)

0.050 (1.27)

BSC

PIN 1

DIMENSIONS:

INCHES (MM)

0.0098 (0.249)

0.0040 (0.102)

16-pin SOP (M)

2.00 (0.079)

1.73 (0.068)

0.21 (0.008)

0.05 (0.002)

COPLANARITY:

0.10 (0.004) MAX

1.25 (0.049) REF

0.65 (0.0260)

BSC

0.875

(0.034) REF

10°

4°

0°

–8°

5.40 (0.213)

5.20 (0.205)

7.90 (0.311)

7.65 (0.301)

6.33 (0.239)

6.07 (0.249)

0.38 (0.015)

0.25 (0.010)

0.22 (0.009)

0.13 (0.005)

0.95 (0.037)

0.55 (0.022)

DIMENSIONS:

MM (INCH)

16-Pin SSOP (SM)

April 22, 2004 27 M9999-042204

MIC2182 Micrel

M9999-042204 28 April 22, 2004

MICREL, INC. 1849 FORTUNE DRIVE SAN JOSE, CA 95131 USA

TEL + 1 (408) 944-0800 FAX + 1 (408) 474-1000 WEB http://www.micrel.com

The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use.

Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.

Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can

reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into

the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s

use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser’s own risk and Purchaser agrees to fully indemnify

Micrel for any damages resulting from such use or sale.