LT1512
1
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TYPICAL APPLICATION
DESCRIPTION
SEPIC Constant-Current/
Constant-Voltage Battery Charger
The LT
®
1512 is a 500kHz current mode switching regulator
specially confi gured to create a constant-current/constant-
voltage battery charger. In addition to the usual voltage
feedback node, it has a current sense feedback circuit for
accurately controlling output current of a fl yback or SEPIC
(Single-Ended Primary Inductance Converter) topology
charger. These topologies allow the current sense circuit
to be ground referred and completely separated from the
battery itself, simplifying battery switching and system
grounding problems. In addition, these topologies allow
charging even when the input voltage is lower than the
battery voltage.
Maximum switch current on the LT1512 is 1.5A. This allows
battery charging currents up to 1A for a single lithium-ion
cell. Accuracy of 1% in constant-voltage mode is perfect
for lithium battery applications. Charging current can be
easily programmed for all battery types.
Maximum Charging Current
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
FEATURES
APPLICATIONS
n Battery Charging of NiCd, NiMH, Lead-Acid or
Lithium Rechargeable Cells
n Precision Current Limited Power Supply
n Constant-Voltage/Constant-Current Supply
n Transducer Excitation
*Maximum Input Voltage = 40V – VBAT
n Charger Input Voltage May Be Higher, Equal to or
Lower Than Battery Voltage
n Charges Any Number of Cells Up to 30V*
n 1% Voltage Accuracy for Rechargeable Lithium
Batteries
n 500kHz Switching Frequency Minimizes
Inductor Size
n 100mV Current Sense Voltage for High Effi ciency
n Battery Can Be Directly Grounded
n Charging Current Easily Programmable or Shut Down
Figure 1. SEPIC Charger with 0.5A Output Current
LT1512
IFB
VC
VIN
L1 A*
L1 B*
0.5A
•
GND GND S
FB
1512 F01
VSW
SYNC
AND/OR
SHUTDOWN
WALL
ADAPTER
INPUT
S/S
C3
22μF
25V
C2**
2.2μF
C5
0.1μF
*
**
L1 A, L1 B ARE TWO 33mH WINDINGS ON A
SINGLE INDUCTOR: COILTRONICS CTX33-3
TOKIN CERAMIC 1E225ZY5U-C203-F
C4
0.22μF
R4
24Ω
+
CHARGE
SHUTDOWN
•
R1
R2
R3
0.2Ω
R5
1k
C1
22μF
25V
+
D1
MBRS130LT3
ACTUAL PROGRAMMED CHARGING CURRENT WILL BE INDEPENDENT OF INPUT
VOLTAGE AND BATTERY VOLTAGE IF IT DOES NOT EXCEED THE VALUES SHOWN.
THESE ARE ELECTRICAL LIMITATIONS BASED ON MAXIMUM SWITCH CURRENT.
PACKAGE THERMAL LIMITATIONS MAY REDUCE MAXIMUM CHARGING CURRENT.
SEE APPLICATIONS INFORMATION.
INPUT VOLTAGE (V)
0
CURRENT (A)
0.6
0.8
1.0
20
1512 TA02
0.4
0.2
0510 15 25
INDUCTOR = 33μH
DOUBLE
LITHIUM
CELL (8.2V)
6V BATTERY
12V BATTERY
SINGLE
LITHIUM
CELL (4.1V)
LT1512
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PIN CONFIGURATION ABSOLUTE MAXIMUM RATINGS
Input Voltage ........................................................... 30V
Switch Voltage ......................................................... 40V
S/S Pin Voltage ........................................................ 30V
FB Pin Voltage (Transient, 10ms) ........................... ±10V
VFB Pin Current ..................................................... 10mA
IFB Pin Voltage (Transient, 10ms) .......................... ±10V
Storage Temperature Range .................. –65°C to 150°C
Ambient Temperature Range
LT1512C (Note 2) ..................................... 0°C to 70°C
LT1512I ................................................ –40°C to 85°C
Operating Junction Temperature Range
LT1512C (Note 2) .............................. –20°C to 125°C
LT1512I .............................................. –40°C to 125°C
Short Circuit.......................................... 0°C to 150°C
Lead Temperature (Soldering, 10 sec) ................. 300°C
1
2
3
4
8
7
6
5
TOP VIEW
VC
FB
IFB
S/S
VSW
GND
GND S
VIN
N8 PACKAGE
8-LEAD PDIP
S8 PACKAGE
8-LEAD PLASTIC SO
TJMAX = 125°C, θJA = 100°C/W (N)
TJMAX = 125°C, θJA = 130°C/W (S)
NOTE: CONTACT FACTORY CONCERNING 16-LEAD
FUSED-LEAD GN PACKAGE WITH LOWER THERMAL RESISTANCE
ELECTRICAL CHARACTERISTICS
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
VREF VFB Reference Voltage Measured at FB Pin
VC = 0.8V l
1.233
1.228
1.245
1.245
1.257
1.262
V
V
FB Input Current VFB = VREF
l
300 550
600
nA
nA
FB Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 25V, VC = 0.8V l0.01 0.03 %/V
The l denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at TA = 25°C. VIN = 5V, VC = 0.6V, VFB = VREF, IFB = 0V, VSW and S/S pins open,
unless otherwise noted.
ORDER INFORMATION
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT1512CN8#PBF LT1512CN8#TRPBF 1512 8-Lead PDIP 0°C to 70°C
LT1512CS8#PBF LT1512CS8#TRPBF 1512 8-Lead Plastic SO 0°C to 70°C
LT1512IN8#PBF LT1512IN8#TRPBF 1512I 8-Lead PDIP –40°C to 85°C
LT1512IS8#PBF LT1512IS8#TRPBF 1512I 8-Lead Plastic SO –40°C to 85°C
LEAD BASED FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT1512CN8 LT1512CN8#TR 1512 8-Lead PDIP 0°C to 70°C
LT1512CS8 LT1512CS8#TR 1512 8-Lead Plastic SO 0°C to 70°C
LT1512IN8 LT1512IN8#TR 1512I 8-Lead PDIP –40°C to 85°C
LT1512IS8 LT1512IS8#TR 1512I 8-Lead Plastic SO –40°C to 85°C
Consult LTC Marketing for parts specifi ed with wider operating temperature ranges. *The temperature grade is identifi ed by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifi
cations, go to: http://www.linear.com/tapeandreel/
(Note 1)
LT1512
3
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Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: Commercial devices are guaranteed over 0°C to 125°C junction
temperature range and 0°C to 70°C ambient temperature range. These
parts are also designed, characterized and expected to operate over the
–20°C to 85°C extended ambient temperature range, but are not tested at
–20°C or 85°C. Devices with full guaranteed electrical specifi cations over
the ambient temperature range –40°C to 85°C are available as industrial
parts with an “I” suffi x.
ELECTRICAL CHARACTERISTICS
The l denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at TA = 25°C. VIN = 5V, VC = 0.6V, VFB = VREF, IFB = 0V, VSW and S/S pins open,
unless otherwise noted.
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
VIREF IFB Reference Voltage Measured at IFB Pin
VFB = 0V, VC = 0.8V l
–107
–110
–100
–100
–93
–90
mV
mV
IFB Input Current VIFB = VIREF (Note 3) l10 25 35 μA
IFB Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 25V, VC = 0.8V l0.01 0.05 %/V
gmError Amplifi er Transconductance ΔIC = ±25μA
l
1100
700
1500 1900
2300
μmho
μmho
Error Amplifi er Source Current VFB = VREF – 150mV, VC = 1.5V l120 200 350 μA
Error Amplifi er Sink Current VFB = VREF + 150mV, VC = 1.5V l1400 2400 μA
Error Amplifi er Clamp Voltage High Clamp, VFB = 1V
Low Clamp, VFB = 1.5V
1.70
0.25
1.95
0.40
2.30
0.52
V
V
AVError Amplifi er Voltage Gain 500 V/V
VC Pin Threshold Duty Cycle = 0% 0.8 1 1.25 V
f Switching Frequency 2.7V ≤ VIN ≤ 25V
0°C ≤ TJ ≤ 125°C
–40°C ≤ TJ < 0°C (LT1512I)
l
450
430
400
500
500
550
580
580
kHz
kHz
kHz
Maximum Switch Duty Cycle l85 95 %
Switch Current Limit Blanking Time 130 260 ns
BV Output Switch Breakdown Voltage 0°C ≤ TJ ≤ 125°C
–40°C ≤ TJ < 20°C (LT1512I)
l40
35
47 V
V
VSAT Output Switch ON Resistance ISW = 2A l0.5 0.8 Ω
ILIM Switch Current Limit Duty Cycle = 50%
Duty Cycle = 80% (Note 4)
l
l
1.5
1.3
1.9
1.7
2.7
2.5
A
A
ΔIIN
ΔISW
Supply Current Increase During Switch ON Time 15 25 mA/A
Control Voltage to Switch Current Transconductance 2 A/V
Minimum Input Voltage l2.4 2.7 V
IQSupply Current 2.7V ≤ VIN ≤ 25V l4 5.5 mA
Shutdown Supply Current 2.7V ≤ VIN ≤ 25V, VS/S ≤ 0.6V
0°C ≤ TJ ≤ 125°C
–40°C ≤ TJ ≤ 0°C (LT1512I)
l12 30
50
μA
μA
Shutdown Threshold 2.7V ≤ VIN ≤ 25V l0.6 1.3 2 V
Shutdown Delay l51225 μs
S/S Pin Input Current 0V ≤ VS/S ≤ 5V l–10 15 μA
Synchronization Frequency Range l600 800 kHz
Maximum allowable ambient temperature may be limited by power
dissipation. Parts may not necessarily be operated simultaneously
at maximum power dissipation and maximum ambient temperature.
Temperature rise calculations must be done as shown in the Applications
Information section to ensure that maximum junction temperature does
not exceed 125°C limit. With high power dissipation, maximum ambient
temperature may be less than 70°C.
Note 3: The IFB pin is servoed to its regulating state with VC = 0.8V.
Note 4: For duty cycles (DC) between 50% and 85%, minimum guaranteed
switch current is given by ILIM = 0.667 (2.75 – DC).
LT1512
4
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TYPICAL PERFORMANCE CHARACTERISTICS
SWITCH CURRENT (A)
0
SWITCH SATURATION VOLTAGE (V)
0.6
0.8
1.0
1.6
1512 G01
0.4
0.2
0.5
0.7
0.9
0.3
0.1
00.4 0.8 1.2 2.0
1.4
0.2 0.6 1.0 1.8
100°C
150°C
25°C
–55°C
DUTY CYCLE (%)
SWITCH CURRENT LIMIT (A)
3.0
2.5
2.0
1.5
1.0
0.5
020 40 60 80
1512 G02
10010
030 50 70 90
25°C AND
125°C
–55°C
TEMPERATURE (°C)
–50
1.8
INPUT VOLTAGE (V)
2.0
2.2
2.4
2.6
050
100 150
1512 G03
2.8
3.0
–25 25 75 125
TEMPERATURE (°C)
–50
0
MINIMUM SYNCHRONIZATION VOLTAGE (VP-P)
0.5
1.0
1.5
2.0
050
100 150
1512 G04
2.5
3.0
–25 25 75 125
fSYNC = 700kHz
TEMPERATURE (°C)
–50
FEEDBACK INPUT CURRENT (nA)
400
500
600
150
1512 G05
300
200
0050 100
100
800
700
–25 25 75 125
VFB = VREF
TEMPERATURE (°C)
–50
–50
NEGATIVE FEEDBACK INPUT CURRENT (μA)
–30
0
050 75
1512 G06
–40
–10
–20
–25 25 100 125 150
Minimum Peak-to-Peak
Synchronization Voltage vs Temp
Feedback Input Current
vs Temperature
Negative Feedback Input Current
vs Temperature
Switch Saturation Voltage
vs Switch Current
Switch Current Limit
vs Duty Cycle
Minimum Input Voltage
vs Temperature
VC: The compensation pin is primarily used for frequency
compensation, but it can also be used for soft starting and
current limiting. It is the output of the error amplifi er and
the input of the current comparator. Peak switch current
increases from 0A to 1.8A as the VC voltage varies from
1V to 1.9V. Current out of the VC pin is about 200μA when
the pin is externally clamped below the internal 1.9V clamp
level. Loop frequency compensation is performed with a
capacitor or series RC network from the VC pin directly to
the ground pin (avoid ground loops).
FB: The feedback pin is used for positive output voltage
sensing. This pin is the inverting input to the voltage
error amplifi er. The R1/R2 voltage divider connected to
FB defi nes Li-Ion fl oat voltage at full charge, or acts as a
voltage limiter for NiCd or NiMH applications. Input bias
current is typically 300nA, so divider current is normally
set to 100μA to swamp out any output voltage errors due
to bias current. The noninverting input of this amplifi er is
tied internally to a 1.245V reference. The grounded end of
the output voltage divider should be connected directly to
the LT1512 ground pin (avoid ground loops).
PIN FUNCTIONS
LT1512
5
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PIN FUNCTIONS
IFB: The current feedback pin is used to sense charging
current. It is the input to a current sense amplifi er that
controls charging current when the battery voltage is below
the programmed voltage. During constant-current opera-
tion, the IFB pin regulates at –100mV. Input resistance of
this pin is 5kΩ, so fi lter resistance (R4, Figure 1) should be
less than 50Ω. The 24Ω, 0.22μF fi lter shown in Figure 1 is
used to convert the pulsating current in the sense resistor
to a smooth DC current feedback signal.
S/S: This pin can be used for shutdown and/or synchro-
nization. It is logic level compatible, but can be tied to VIN
if desired. It defaults to a high ON state when fl oated. A
logic low state will shut down the charger to a micropower
state. Driving the S/S pin with a continuous logic signal of
600kHz to 800kHz will synchronize switching frequency
to the external signal. Shutdown is avoided in this mode
with an internal timer.
VIN: The input supply pin should be bypassed with a
low ESR capacitor located right next to the IC chip. The
grounded end of the capacitor must be connected directly
to the ground plane to which the GND pin is connected.
GND S, GND: The LT1512 uses separate ground pins for
switch current (GND) and the control circuitry (GND S).
This isolates the control ground from any induced volt-
age created by fast switch currents. Both pins should be
tied directly to the ground plane, but the external control
circuit components such as the voltage divider, frequency
compensation network and IFB bypass capacitor should be
connected directly to the GND S pin or to the ground plane
close to the point where the GND S pin is connected.
VSW: The switch pin is the collector of the power switch,
carrying up to 1.5A of current with fast rise and fall times.
Keep the traces on this pin as short as possible to mini-
mize radiation and voltage spikes. In particular, the path
in Figure 1 which includes SW to C2, D1, C1 and around
to the LT1512 ground pin should be as short as possible
to minimize voltage spikes at switch turn-off.
BLOCK DIAGRAM
–
+
IFBA
IFB
S/S
FB
5k
62k
0.08Ω
–
–
+
EA
VC
VIN
GND 1512 F02
GND S
1.245V
REF
500kHz
OSC
SYNC
SHUTDOWN
DELAY AND RESET
LOW DROPOUT
2.3V REG ANTI-SAT
LOGIC DRIVER
SW
SWITCH
–
+
IA
AV ≈ 6
COMP
Figure 2
LT1512
6
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OPERATION
The LT1512 is a current mode switcher. This means that
switch duty cycle is directly controlled by switch current
rather than by output voltage or current. Referring to the
Block Diagram, the switch is turned “on” at the start of
each oscillator cycle. It is turned “off” when switch current
reaches a predetermined level. Control of output voltage and
current is obtained by using the output of a dual feedback
voltage sensing error amplifi er to set switch current trip
level. This technique has the advantage of simplifi ed loop
frequency compensation. A low dropout internal regula-
tor provides a 2.3V supply for all internal circuitry on the
LT1512. This low dropout design allows input voltage to
vary from 2.7V to 25V. A 500kHz oscillator is the basic
clock for all internal timing. It turns “on” the output switch
via the logic and driver circuitry. Special adaptive antisat
circuitry detects onset of saturation in the power switch
and adjusts driver current instantaneously to limit switch
saturation. This minimizes driver dissipation and provides
very rapid turn-off of the switch.
A unique error amplifi er design has two inverting inputs
which allow for sensing both output voltage and current.
A 1.245V bandgap reference biases the noninverting input.
The fi rst inverting input of the error amplifi er is brought out
for positive output voltage sensing. The second inverting
input is driven by a “current” amplifi er which is sensing
output current via an external current sense resistor. The
current amplifi er is set to a fi xed gain of –12.5 which
provides a –100mV current limit sense voltage.
The error signal developed at the amplifi er output is brought
out externally and is used for frequency compensation.
During normal regulator operation this pin sits at a voltage
between 1V (low output current) and 1.9V (high output
current). Switch duty cycle goes to zero if the VC pin is
pulled below the VC pin threshold, placing the LT1512 in
an idle mode.
The LT1512 is an IC battery charger chip specifi cally op-
timized to use the SEPIC converter topology. The SEPIC
topology has unique advantages for battery charging. It
will operate with input voltages above, equal to or below
the battery voltage, has no path for battery discharge when
turned off and eliminates the snubber losses of fl yback
designs. It also has a current sense point that is ground
referred and need not be connected directly to the battery.
The two inductors shown are actually just two identical
windings on one inductor core, although two separate
inductors can be used.
A current sense voltage is generated with respect to ground
across R3 in Figure 1. The average current through R3 is
always identical to the current delivered to the battery. The
LT1512 current limit loop will servo the voltage across R3
to –100mV when the battery voltage is below the voltage
limit set by the output divider R1/R2. Constant current
charging is therefore set at 100mV/R3. R4 and C4 fi lter
the current signal to deliver a smooth feedback voltage to
the IFB pin. R1 and R2 form a divider for battery voltage
sensing and set the battery fl oat voltage. The suggested
value for R2 is 12.4k. R1 is calculated from:
APPLICATIONS INFORMATION
RRV
RμA
BAT
12 1 245
1 245 2 0 3
=
+
(–.)
.(.)
V
BAT = battery fl oat voltage
0.3μA = typical FB pin bias current
A value of 12.4k for R2 sets divider current at 100μA.
This is a constant drain on the battery when power to the
charger is off. If this drain is too high, R2 can be increased
to 41.2k, reducing divider current to 30μA. This introduces
an additional uncorrectable error to the constant voltage
fl oat mode of about ±0.5% as calculated by:
V Error= 0.15 A(R1)(R2)
1.245(R1+R2)
BAT
±μ
±0.15μA = expected variation in FB bias current around
the nominal 0.3μA typical value.
With R2 = 41.2k and R1 = 228k, (VBAT = 8.2V), the error
due to variations in bias current would be ±0.42%.
A second option is to disconnect the voltage divider with
a small NMOS transistor as shown in Figure 3. To ensure
LT1512
7
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adequate drive to the transistor (even when the VIN voltage
is at its lowest operating point of 2.4V), the FET gate is
driven wth a peak detected voltage via D2. Note that there
are two connections for D2. The L1 A connection must be
used if the voltage divider is set for less than 3.5V (fully
charged battery). Gate drive is equal to battery voltage
plus input voltage. The disadvantage of this connection is
that Q1 will still be “on” if the VIN voltage is active and the
charger is shut down via the S/S pin. The L1 B connection
allows Q1 to turn off when VIN is off or when shutdown is
initiated, but the reduced gate drive (= VBAT) is not adequate
to ensure a Q1 on-state for fully charged battery voltages
less than 3.5V. Do not substitute for Q1 unless the new
device has adequate VGS maximum rating, especially if
D2 is connected to L1A. C6 fi lters the gate drive and R5
pulls the gate low when switching stops.
Disconnecting the divider leaves only D1 diode leakage
as a battery drain. See Diode Selection for a discussion
of diode leakage.
Maximum Input Voltage
Maximum input voltage for the circuit in Figure 1 is partly
determined by battery voltage. A SEPIC converter has a
maximum switch voltage equal to input voltage plus out-
put voltage. The LT1512 has a maximum input voltage of
30V and a maximum switch voltage of 40V, so this limits
maximum input voltage to 30V, or 40V – VBAT, whichever
is less. Maximum VBAT = 40V – VIN.
Shutdown and Synchronization
The dual function S/S pin provides easy shutdown and
synchronization. It is logic level compatible and can be
pulled high or left fl oating for normal operation. A logic
low on the S/S pin activates shutdown, reducing input
supply current to 12μA. To synchronize switching, drive
the S/S pin between 600kHz and 800kHz.
Inductor Selection
L1A and L1B are normally just two identical windings on
one core, although two separate inductors can be used.
A typical value is 33μH, which gives about 0.25A peak-to-
peak inductor current. Lower values will give higher ripple
current, which reduces maximum charging current. 15μH
can be used if charging currents are at least 20% lower than
the values shown in the maximum charging current graph.
Higher inductance values give slightly higher maximum
charging current, but are larger and more expensive. A
low loss toroid core such as KoolMμ, Molypermalloy or
Metglas is recommended. Series resistance should be
less than 0.1Ω for each winding. “Open core” inductors,
such as rods or barrels are not recommended because
they generate large magnetic fi elds which may interfere
with other electronics close to the charger.
Input Capacitor
The SEPIC topology has relatively low input ripple current
compared to other topologies and higher harmonics are
APPLICATIONS INFORMATION
Figure 3. Eliminating Divider Current
LT1512
VIN
L1 A
L1 B
GND FB
1512 F03
VSW
SHUTDOWN
D2
1N4148
CONNECT D2 ANODE HERE IF FULLY CHARGED
BATTERY VOLTAGE IS GREATER THAN 3.5V AND
Q1 MUST BE TURNED OFF IN SHUTDOWN WITH
VIN STILL ACTIVE
CONNECT D2 ANODE HERE FOR FULLY
CHARGED BATTERY VOLTAGE LESS
THAN 3.5V. Q1 WILL NOT BE TURNED OFF
IN SHUTDOWN IF VIN IS PRESENT
S/S
C2
C6
470pF
•
R1
R5
470k
R2
R3
Q1
2N7002
+
D1
LT1512
8
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APPLICATIONS INFORMATION
especially low. RMS ripple current in the input capacitor is
less than 0.1A with L = 33μH and less than 0.2A with L =
15μH. A low ESR 22μF, 25V solid tantalum capacitor (AVX
type TPS or Sprague type 593D) is adequate for most appli-
cations with the following caveat. Solid tantalum capacitors
can be destroyed with a very high turn-on surge current
such as would be generated if a low impedance input source
were “hot switched” to the charger input. If this condition
can occur, the input capacitor should have the highest pos-
sible voltage rating, at least twice the surge input voltage if
possible. Consult with the capacitor manufacturer before
a fi nal choice is made. A 2.2μF ceramic capacitor such as
the one used for the coupling capacitor can also be used.
These capacitors do not have a turn-on surge limitation.
The input capacitor must be connected directly to the VIN
pin and the ground plane close to the LT1512.
Output Capacitor
It is assumed as a worst case that all the switching out-
put ripple current from the battery charger could fl ow in
the output capacitor. This is a desirable situation if it is
necessary to have very low switching ripple current in
the battery itself. Ferrite beads or line chokes are often
inserted in series with the battery leads to eliminate high
frequency currents that could create EMI problems. This
forces all the ripple current into the output capacitor. Total
RMS current into the capacitor has a maximum value of
about 0.5A, and this is handled with a 22μF, 25V capacitor
shown in Figure 1. This is an AVX type TPS or Sprague
type 593D surface mount solid tantalum unit intended
for switching applications. Do not substitute other types
without ensuring that they have adequate ripple current
ratings. See Input Capacitor section for details of surge
limitation on solid tantalum capacitors if the battery may
be “hot switched” to the output of the charger.
Coupling Capacitor
C2 in Figure 1 is the coupling capacitor that allows a SEPIC
converter topology to work with input voltages either
higher or lower than the battery voltage. DC bias on the
capacitor is equal to input voltage. RMS ripple current
in the coupling capacitor has a maximum value of about
0.5A at full charging current. A conservative formula to
calculate this is:
IIVV
V
COUP RMS CHRG IN BAT
IN
()
()(.)
()
=
+11
2
(1.1 is a fudge factor to account for inductor ripple current
and other losses)
2 WINDING
INDUCTOR
L1A
1512 F04a
L1B
R4
R1
1 4
32
C4 R2
D1
VIN
GND GND
VBATT
R3
C2AC2B
C3
C5
C1
U1
R5
S/S
12
3
4
VC
FB
IFB
S/S
VSW
GND
GND S
VIN
R5 C4
R3 R4
S/S
1512 F04b
C5
R2
C1
D1 C2
C3
R1
VBATT
+ +
+VIN
GND
L1B
L1A
R4
a. Double-Sided (Vias Connect to the Backside of Ground Plane. Dash
Lines Indicate Interconnects on Backside. Demo Board Uses This
Layout, Except that R5 Has Been Added to Increase Phase Margin) b. Single-Sided Alternative Layout
Figure 4. LT1512 Suggested Layouts for Critical Thermal and Electrical Paths
LT1512
9
1512fa
APPLICATIONS INFORMATION
With ICHRG = 0.5A, VIN = 15V and VBAT = 8.2V, ICOUP = 0.43A
The recommended capacitor is a 2.2μF ceramic type from
Marcon or Tokin. These capacitors have extremely low ESR
and high ripple current ratings in a small package. Solid
tantalum units can be substituted if their ripple current
rating is adequate, but typical values will increase to 22μF
or more to meet the ripple current requirements.
Diode Selection
The switching diode should be a Schottky type to minimize
both forward and reverse recovery losses. Average diode
current is the same as output charging current , so this
will be under 1A. A 1A diode is recommended for most
applications, although smaller devices could be used at
reduced charging current. Maximum diode reverse voltage
will be equal to input voltage plus battery voltage.
Diode reverse leakage current will be of some concern
during charger shutdown. This leakage current is a direct
drain on the battery when the charger is not powered. High
current Schottky diodes have relatively high leakage currents
(2μA to 200μA) even at room temperature. The latest very-
low-forward devices have especially high leakage currents.
It has been noted that surface mount versions of some
Schottky diodes have as much as ten times the leakage of
their through-hole counterparts. This may be because a low
forward voltage process is used to reduce power dissipation
in the surface mount package. In any case, check leakage
specifi cations carefully before making a fi nal choice for the
switching diode. Be aware that diode manufacturers want to
specify a maximum leakage current that is ten times higher
than the typical leakage. It is very diffi cult to get them to
specify a low leakage current in high volume production.
This is an on going problem for all battery charger circuits
and most customers have to settle for a diode whose typi-
cal leakage is adequate, but theoretically has a worst-case
condition of higher than desired battery drain.
Thermal Considerations
Care should be taken to ensure that worst-case conditions
do not cause excessive die temperatures. Typical thermal
resistance is 130°C/W for the S8 package but this number
will vary depending on the mounting technique (copper
area, air fl ow, etc).
Average supply current (including driver current) is:
ImA
VI
V
IN BAT CHRG
IN
=+40 024()( )(.)
Switch power dissipation is given by:
PIRVVV
V
SW CHRG SW BAT IN BAT
IN
=
+()()( )()
()
2
2
R
SW = output switch ON resistance
Total power dissipation of the die is equal to supply current
times supply voltage, plus switch power:
P
D(TOTAL) = (IIN)(VIN) + PSW
For VIN = 10V, VBAT = 8.2V, ICHRG = 0.5A, RSW = 0.65Ω
I
IN = 4mA + 10mA = 14mA
P
SW = 0.24W
P
D = (0.014)(10) + 0.24 = 0.38W
The S8 package has a thermal resistance of 130°C/W.
(Contact factory concerning 16-lead fused-lead pack-
age with footprint approximately same as S8 package
and with lower thermal resistance.) Die temperature rise
will be (0.38W)(130°C/W) = 49°C. A maximum ambient
temperature of 60°C will give a die temperature of 60°C +
49°C = 109°C. This is only slightly less than the maximum
junction temperature of 125°C, illustrating the importance
of doing these calculations!
Programmed Charging Current
LT1512 charging current can be programmed with a PWM
signal from a processor as shown in Figure 5. C6 and D2
form a peak detector that converts a positive logic signal
to a negative signal. The average negative signal at the
+
C6
1μF
C7
10μF
C4
0.22μF R3
1512 F05
L1B
IFB
LT1512
R5
4.02k
PWM
INPUT
≥1kHz D2
R6
4.02k
R4
200Ω
+
Figure 5. Programming Charge Current
LT1512
10
1512fa
APPLICATIONS INFORMATION
input to R5 is equal to the processor VCC level multiplied
by the inverse PWM ratio. This assumes that the PWM
signal is a CMOS output that swings rail-to-rail with a
source resistance less than a few hundred ohms. The
negative voltage is converted to a current by R5 and R6
and fi ltered by C7. This current multiplied by R4 generates
a voltage that subtracts from the 100mV sense voltage
of the LT1512. This is not a high precision technique
because of the errors in VCC and the diode voltage, but
it can typically be used to adjust charging current over a
20% to 100% range with good repeatability (full charg-
ing current accuracy is not affected). To reduce the load
on the logic signal, R4 has been increased from 24Ω to
200Ω. This causes a known increase in full-scale charging
PACKAGE DESCRIPTION
N8 Package
8-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
Dimensions in inches (millimeters) unless otherwise noted.
N8 1002
.065
(1.651)
TYP
.045 – .065
(1.143 – 1.651)
.130 p .005
(3.302 p 0.127)
.020
(0.508)
MIN
.018 p .003
(0.457 p 0.076)
.120
(3.048)
MIN
12 34
87 65
.255 p .015*
(6.477 p 0.381)
.400*
(10.160)
MAX
.008 – .015
(0.203 – 0.381)
.300 – .325
(7.620 – 8.255)
.325 +.035
–.015
+0.889
–0.381
8.255
NOTE:
1. DIMENSIONS ARE INCHES
MILLIMETERS
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .010 INCH (0.254mm)
.100
(2.54)
BSC
current (PWM = 0) of 3% due to the 5k input resistance of
the IFB pin. Note that 100% duty cycle gives full charging
current and that very low duty cycles (especially zero!)
will not operate correctly. Very low duty cycle (<10%)
is a problem because the peak detector requires a fi nite
up-time to reset C6.
More Help
Linear Technology Field Application Engineers have a CAD
spreadsheet program for detailed calculations of circuit
operating conditions, and our Applications Department is
always ready to lend a helping hand. For additional informa-
tion refer to the LT1372 data sheet. This part is identical to
the LT1512 except for the current amplifi er circuitry.
LT1512
11
1512fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
.016 – .050
(0.406 – 1.270)
.010 – .020
(0.254 – 0.508)s 45o
0o– 8o TYP
.008 – .010
(0.203 – 0.254)
SO8 0303
.053 – .069
(1.346 – 1.752)
.014 – .019
(0.355 – 0.483)
TYP
.004 – .010
(0.101 – 0.254)
.050
(1.270)
BSC
1234
.150 – .157
(3.810 – 3.988)
NOTE 3
8765
.189 – .197
(4.801 – 5.004)
NOTE 3
.228 – .244
(5.791 – 6.197)
.245
MIN .160 p.005
RECOMMENDED SOLDER PAD LAYOUT
.045 p.005
.050 BSC
.030 p.005
TYP
INCHES
(MILLIMETERS)
NOTE:
1. DIMENSIONS IN
2. DRAWING NOT TO SCALE
3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .006" (0.15mm)
LT1512
12
1512fa
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2008
LT 0109 REV A • PRINTED IN USA
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TYPICAL APPLICATION
PART NUMBER DESCRIPTION COMMENTS
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Main Battery Removed
LT C
®
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with Software Charging Profi les
LT1510 1.5A Constant-Current/Constant-Voltage Battery Charger Step-Down Charger for Li-Ion, NiCd and NiMH
LT1511 3.0A Constant-Current/Constant-Voltage Battery Charger
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Step-Down Charger that Allows Charging During Computer Operation and
Prevents Wall-Adapter Overload
LT1513 SEPIC Constant-Current/Constant-Voltage Battery Charger Step-Up/Step-Down Charger for Up to 2A Current
The circuit in Figure 6 will provide adapter current limit-
ing to ensure that the battery charger never overloads
the adapter. In addition, it adjusts charging current to a
lower value if other system power increases to the point
where the adapter would be overloaded. This allows the
LT1512 to charge the battery at the maximum possible
rate without concern about varying system power levels.
The LM301 op amp used here is unusual in that it can
operate with its inputs at a voltage equal to the positive
supply voltage.
Figure 6. Adding Adapter Current Limiting
LT1512
IFB
VC
VIN
L1 A*
L1 B*
0.5A
1
1
3
3
2
28
8
5
•
•
4
4
6
7
7
6
TO FB PIN
D2
1N4148
GND GND S
FB
1512 F06
VSW
SYNC
AND/OR
SHUTDOWN
WALL
ADAPTER
INPUT
S/S
C3
22μF
25V
SYSTEM
POWER
C2**
2.2μF
C5
0.1μF
R5
1k
R7
12k
Q1
2N3904
*
**
L1 A, L1 B ARE TWO 33mH WINDINGS ON A
COMMON CORE: COILTRONICS CTX33-3
TOKIN CERAMIC 1E225ZY5U-C203-F
C4
0.22μF
R4
24Ω
+
R1
R2
R3
0.2Ω
C1
22μF
25V
+
D1
MBRS130LT3
R6
0.2Ω
–+
30pF
LM301
