VOUT
VIN
CS
SS
OUT
FEEDBACK
WITH
ISOLATION
COMP GND
RT
VCC
LM5021
90 ~ 264 Vac
+
+
AC
LM5021
www.ti.com
SNVS359D –MAY 2005–REVISED MARCH 2013
LM5021 AC-DC Current Mode PWM Controller
Check for Samples: LM5021
1FEATURES DESCRIPTION
The LM5021 off-line pulse width modulation (PWM)
2• Ultra Low Start-up Current (25 µA maximum) controller contains all of the features needed to
• Current Mode Control implement highly efficient off-line single-ended
• Skip Cycle Mode for Low Standby Power flyback and forward power converters using current-
mode control. The LM5021 features include an ultra-
• Single Resistor Programmable Oscillator low (25 µA) start-up current, which minimizes power
• Synchronizable Oscillator losses in the high voltage start-up network. A skip
• Adjustable Soft-Start cycle mode reduces power consumption with light
loads for energy conserving applications (ENERGY
• Integrated 0.7A Peak Gate Driver STAR®, CECP, etc.). Additional features include
• Direct Opto-Coupler Interface under-voltage lockout, cycle-by-cycle current limit,
• Maximum Duty Cycle Limiting (80% for hiccup mode overload protection, slope com-
LM5021-1 or 50% for LM5021-2) pensation, soft-start and oscillator synchronization
capability. This high performance 8-pin IC has total
• Slope Compensation for (LM5021-1 Only) propagation delays less than 100nS and a 1MHz
• Under Voltage Lockout (UVLO) with Hysteresis capable oscillator that is programmed with a single
• Cycle-by-Cycle Over-Current Protection resistor.
• Hiccup Mode for Continuous Overload WHITE SPACE
Protection
• Leading Edge Blanking of Current Sense
Signal
• Packages: VSSOP-8 or PDIP-8
Simplified Application Diagram
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 2005–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
VIN
VCC
OUT
RT
CS
GND
SS
COMP 1
2
3
4
8
7
6
5
LM5021
SNVS359D –MAY 2005–REVISED MARCH 2013
www.ti.com
Connection Diagram
Figure 1. Top View
VSSOP-8
(See Package Number DGK0008A)
PDIP-8
(See Package Number P0008E)
PIN DESCRIPTIONS
Pin Name Description Application Information
1 COMP Control input for the Pulse Width Modulator and COMP pull-up is provided by an internal 5K resistor which may
Hiccup comparators. be used to bias an opto-coupler transistor.
2 VIN Input voltage. Input to start-up regulator. The VIN pin is clamped at 36V by
an internal zener diode.
3 VCC Output only of a linear bias supply regulator. VCC provides bias to controller and gate drive sections of the
Nominally 8.5V. LM5021. An external capacitor must be connected from this pin
to ground.
4 OUT MOSFET gate driver output. High current output to the external MOSFET gate input with
source/sink current capability of 0.3A and 0.7A respectively.
5 GND Ground return.
6 CS Current Sense input. Current sense input for current mode control and over-current
protection. Current limiting is accomplished using a dedicated
current sense comparator. If the CS comparator input exceeds
0.5 Volts the OUT pin switches low for cycle-by-cycle current
limit. CS is held low for 90ns after OUT switches high to blank
the leading edge current spike.
7 RT / SYNC Oscillator timing resistor pin and synchronization An external resistor connected from RT to GND sets the
input. oscillator frequency. This pin will also accept synchronization
pulses from an external clock.
8 SS Soft-start / Hiccup time An external capacitor and an internal 22 µA current source set
the soft-start ramp. The soft -start capacitor controls both the
soft-start rate and the hiccup mode period.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
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SNVS359D –MAY 2005–REVISED MARCH 2013
Absolute Maximum Ratings (1)(2)
VIN to GND -0.3V to 30V
VIN Clamp Continuous Current 5mA
CS to GND -0.3V to 1.25V
RT to GND -0.3V to 5.5V
All other pins to GND -0.3V to 7.0V
ESD Rating (3)
Human Body Model 2kV
Storage Temperature -65°C to +150°C
Operating Junction Temperature +150°C
(1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which
operation of the device is intended to be functional. For specifications and test conditions, see the Electrical Characteristics.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
(3) The human body model is a 100 pF capacitor discharged through a 1.5kΩresistor into each pin.
Operating Ratings (1)
VIN Voltage (2) 8V to 30V
Junction Temperature -40°C to +125°C
(1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which
operation of the device is intended to be functional. For specifications and test conditions, see the Electrical Characteristics.
(2) After initial turn-on at VIN = 20V.
Electrical Characteristics(1)
Specifications in standard type face are for TJ= +25°C and those in boldface type apply over the full Operating Junction
Temperature Range. Unless otherwise specified: VIN = 15V, RT= 44.2KΩ.
Symbol Parameter Conditions Min Typ Max Unit
STARTUP CIRCUIT
Start Up Current Before VCC Enable 18 25 µA
VCC Regulator enable threshold 17 20 23 V
VCC Regulator disable threshold 7.25 V
VIN ESD Clamp voltage I = 5mA 30 36 40 V
IVIN Operating supply current COMP = 0VDC 2.5 3.75 mA
VCC SUPPLY
Controller enable threshold 6.5 77.5 V
Controller disable threshold 5.3 5.8 6.3 V
VCC regulated output No External Load 88.5 9V
VCC dropout voltage (VIN - VCC) I = 5 mA 1.7 V
VCC regulator current limit VCC = 7.5V (2) 15 22 mA
SKIP CYCLE MODE COMPARATOR
Skip Cycle mode enable threshold ⅓[COMP - 1.25V] 75 125 175 mV
Skip Cycle mode hysteresis 5 mV
CURRENT LIMIT
CS limit to OUT delay CS stepped from 0 to 35 ns
0.6V, time to OUT
transition low, Cload =
0.
CS limit threshold 0.45 0.5 0.55 V
Leading Edge Blanking time 90 ns
CS blanking sinking impedance 35 55 Ω
(1) Min and Max limits are 100% production tested at 25°C. Limits over the operating temperature range are specified through correlation
using Statistical Quality Control (SQC) methods. Limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Device thermal limitations may limit usable range.
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Product Folder Links: LM5021
LM5021
SNVS359D –MAY 2005–REVISED MARCH 2013
www.ti.com
Electrical Characteristics(1) (continued)
Specifications in standard type face are for TJ= +25°C and those in boldface type apply over the full Operating Junction
Temperature Range. Unless otherwise specified: VIN = 15V, RT= 44.2KΩ.
Symbol Parameter Conditions Min Typ Max Unit
SOFT-START
VSS-OCV SS pin open-circuit voltage 4.3 5.2 6.1 V
Soft-start Current Source 15 22 30 µA
Soft-start to COMP Offset 0.35 0.55 0.75 V
COMP sinking impedance During SS ramp 60 Ω
OSCILLATOR
Frequency1 (RT = 44.2K) 135 150 165 kHz
Frequency2 (RT = 13.3K) 440 500 560 kHz
Sync threshold 2.4 3.2 3.8 V
PWM COMPARATOR
COMP to OUT delay COMP set to 2V 20 ns
CS stepped 0 to 0.4V,
time to OUT transition
low, Cload = 0.
Min Duty Cycle COMP = 0V 0%
Max Duty Cycle (-1 Device) 75 80 85 %
Max Duty Cycle (-2 Device) 50 %
COMP to PWM comparator gain 0.33
COMP Open Circuit Voltage 4.2 5.1 6V
COMP at Max Duty Cycle 2.75 V
COMP Short Circuit Current COMP = 0V 0.6 1.1 1.5 mA
SLOPE COMPENSATION
Slope Comp Amplitude (LM5021-1 CS pin to PWM 70 90 110 mV
only) Comparator offset at
maximum duty cycle
OUTPUT SECTION
OUT High Saturation IOUT = 50mA, VCC - 0.6 1.1 V
OUT
OUT Low Saturation IOUT = 100mA 0.3 1V
Peak Source Current OUT = VCC/2. 0.3 A
Peak Sink Current OUT = VCC/2. 0.7 A
Rise time Cload = 1nF 25 ns
Fall time Cload = 1nF 10 ns
HICCUP MODE
VOVLD Over load detection threshold COMP pin VSS-OCV – 0.8 VSS-OCV – 0.6 VSS-OCV– 0.4 V
VHIC Hiccup mode threshold SS pin VSS-OCV – 0.8 VSS-OCV – 0.6 VSS-OCV– 0.4 V
VRST Hiccup mode Restart threshold SS pin 0.1 0.3 0.5 V
IDTCS Dead-time current source 0.1 0.25 0.4 µA
IOVCS Overload detection timer current 610 14 µA
source
THERMAL RESISTANCE
θJA VSSOP-8 Junction to Ambient 0 LFM 200 °C/W
θJA PDIP-8 Junction to Ambient 0 LFM 107 °C/W
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Product Folder Links: LM5021
CS
SS
VIN
GND
VCC
8.5V LINEAR
REGULATOR
OSC
OUT
SLOPE COMPENSATION RAMP GENERATOR
DRIVER
PWM
COMPARATOR
5.2V
CLK
22 PA
10 PA
0.25 PA
COMP
SKIP CYCLE
COMPARATOR
MAX DUTY LIMIT
LM5021 - 1 (80%)
LM5021 - 2 (50%)
(LM5021 - 1 ONLY)
+
-
+
-
1.8k
5k
2R
R
125 mV
CURRENT LIMIT
COMPARATOR
36V
CLAMP
RT/ SYNC
50 PA
0 PA
VIN UVLO
20V RISING
7.25V FALLING EN
VCC_UVLO
QS
R
+
-PWM
LOGIC
+
-SS
5.2V
1.25V
550 mV
VCC_UVLO
VCC_UVLO
DIS
SOFTSTART
AND
HICCUP
MODE
LOGIC
COMP
VCC_UVLO
VCC UVLO
7V RISING
5.8V FALLING
1
500 mV
CLK
Leading Edge Blanking
7
23
4
8
5
6
LM5021
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SNVS359D –MAY 2005–REVISED MARCH 2013
Simplified Block Diagram
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1 10 100 1000
SOFTSTART CAPACITANCE (nF)
0.01
0.1
1
10
100
OFF TIME (s)
TEMPERATURE (oC)
OUT PEAK CURRENT (A)
-40 0 40 80 120
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
Sinking
Sourcing
0 5 10 15 20 25
0
5
10
15
20
25
VCC (V)
VIN (V)
0 500 1000 1500 2000
OUT DRIVER LOAD (pF)
VIN CURRENT (mA)
2
3
4
5
6
FS = 160 kHz
FS = 80 kHz
FS = 40 kHz
15 16 17 18 19 20 21
VIN VOLTAGE (V)
0
2
4
6
8
10
12
14
16
VIN CURRENT (PA)
LM5021
SNVS359D –MAY 2005–REVISED MARCH 2013
www.ti.com
Typical Performance Characteristics
Unless otherwise specified: TJ= 25°C.
VIN Start-Up Current VIN UVLO
Figure 2. Figure 3.
VIN Current vs OUT Load VIN Voltage Falling vs VCC Voltage
Figure 4. Figure 5.
OUT Driver Current vs Temperature Hiccup Mode Deadtime vs Softstart Capacitance
Figure 6. Figure 7.
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Product Folder Links: LM5021
RT (kQ)
OUT SWITCHING FREQUENCY (kHz)
1 10 100 1000
10
100
1000
LM5021-1
LM5021-2
LM5021
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SNVS359D –MAY 2005–REVISED MARCH 2013
Typical Performance Characteristics (continued)
Unless otherwise specified: TJ= 25°C. Output Switching Frequency vs RT
Figure 8.
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 7
Product Folder Links: LM5021
UPPER
LOWER
INTERNAL
BIAS
GENERATOR
VCC
UVLO
S
R
Q
+
HV
VIN VCC
REGULATOR VCC
Rstart
CVIN VIN
UVLO
CVCC
Enable Driver
TRANSFORMER
BIAS
WINDING
LM5021
SNVS359D –MAY 2005–REVISED MARCH 2013
www.ti.com
DETAILED OPERATING DESCRIPTION
START UP CIRCUIT
Referring to Figure 9, the input capacitor CVIN is trickle charged through the start-up resistor Rstart, when the
rectified ac input voltage HV is applied. The VIN current consumed by the LM5021 is only 18 µA (nominal) while
the capacitor CVIN is initially charged to the start-up threshold. When the input voltage, VIN reaches the upper
VIN UVLO threshold of 20V, the internal VCC linear regulator is enabled. The VCC regulator will remain on until
VIN falls to the lower UVLO threshold of 7.25V (12.5V hysteresis). When the VCC regulator is turned on, the
external capacitor at the VCC pin begins to charge. The PWM controller, soft-start circuit and gate driver are
enabled when the VCC voltage reaches the VCC UVLO upper threshold of 7V. The VCC UVLO has 1.2V
hysteresis between the upper and lower thresholds to avoid chattering during transients on the VCC pin. When
the VCC UVLO enables the switching power supply, energy is transferred from the primary to the secondary
transformer winding(s). A bias winding, shown in Figure 9, delivers power to the VIN pin to sustain the VCC
regulator. The voltage supplied should be from 11V (VCC regulated voltage maximum plus VCC regulator
dropout voltage) to 30V (maximum operating VIN voltage). The bias winding should always be connected to the
VIN pin as shown in Figure 9. Do not connect the bias winding to the VCC pin. The start-up sequence is
completed and normal operation begins when the voltage from the bias winding is sufficient to maintain VCC
level greater than the VCC UVLO threshold (5.8V typical).
The LM5021 is designed for ultra-low start-up current into the VIN pin. To accomplish this very low start-up
current, the VCC regulator of the LM5021 is unique as compared to the VCC regulator used in other controllers
of the LM5xxx family. The LM5021 is designed specifically for applications with the bias winding connected to the
VIN pin as shown in Figure 9.It is not recommended that the bias winding be connected to the VCC pin of
the LM5021.
The size of the start-up resistor Rstart not only affects power supply start-up time, but also power supply
efficiency since the resistor dissipates power in normal operation. The ultra low start-up current of the LM5021
allows a large value Rstart resistor (up to 3 MΩ) for improved efficiency with reasonable start-up time.
Figure 9. Start-Up Circuit Block Diagram
RELATIONSHIP BETWEEN INPUT CAPACITOR CIN & VCC CAPACITOR CVCC
The internal VCC linear regulator is enabled when VIN reaches 20V. The drop in VIN due to charge transfer from
CVIN to CVCC after the regulator is enabled can be calculated from the following equations where VIN' is the
voltage on CVIN immediately after the VCC regulator charges CVCC.
ΔVIN x CVIN =ΔVCC x CVCC (1)
(20V – VIN') CVIN = 8.5V CVCC (2)
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TVIN_THRESHOLD = RSTART x CVIN x ln 1 - HV
20V -1
Tmax = CVIN x (19.15V - 8.5V)
6.25 mA
= 17 ms
8.5V x CVCC
VIN = 20V - CVIN
LM5021
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SNVS359D –MAY 2005–REVISED MARCH 2013
(3)
Assuming CVIN value as 10 µF, and CVCC of 1µF, then the drop in VIN will be 0.85V, or the VIN value drops to
19.15V. The value of the VCC capacitor can be small (less than 1uF) as it supplies only transient gate drive
current of a short duration. The CVIN capacitor must be sized to supply the gate drive current and the quiescent
current of LM5021until the transformer bias winding delivers sufficient voltage to VIN to sustain the VCC voltage.
The CVIN capacitor value can be calculated from the operating VCC load current after it's output voltage reaches
the VCC UVLO threshold. For example, if the LM5021 is driving an external MOSFET with total gate charge (Qg)
of 25nC, the average gate drive current is Qg x Fsw, where Fsw is the switching frequency. Assuming a
switching frequency of 150KHz, the average gate drive current is 3.75mA. Since the IC consumes approximately
2.5mA operating current in addition to the gate current, the total current drawn from CVIN capacitor is the
operating current plus the gate charge current, or 6.25mA. The CVIN capacitor must supply this current for a brief
time until the transformer bias winding takes over. The CVIN voltage must not fall below 8.5V during the start-up
sequence or the cycle will be restarted. The maximum allowable start-up time can be calculated using the value
of CVIN, the change in voltage allow at VIN (19.15V – 8.5V) and the VCC regulator current (6.25mA). Tmax, the
maximum time allowed to energize the bias winding is:
(4)
If the calculated value of Tmax is too small, the value of Cin should be increased further to allow more time
before the transformer bias winding takes over and delivers the operating current to the VCC regulator.
Increasing CVIN will increase the time from the application of the rectified ac (HV in the Figure 9) to the time when
VIN reaches the 20V start threshold. The initial charging time of CVIN is:
(5)
PWM COMPARATOR/SLOPE COMPENSATION
The PWM comparator compares the current sense signal with the loop error voltage from the COMP pin. The
COMP pin voltage is reduced by 1.25V then attenuated by a 3:1 resistor divider. The PWM comparator input
offset voltage is designed such that less than 1.25V at the COMP pin will result in a zero duty cycle at the
controller output.
For duty cycles greater than 50 percent, current mode control circuits are subject to sub-harmonic oscillation. By
adding an additional fixed slope voltage ramp signal (slope compensation) to the current sense signal, this
oscillation can be avoided. The LM5021-1 integrates this slope compensation by summing a ramp signal
generated by the oscillator with the current sense signal. The slope compensation is generated by a current ramp
driven through an internal 1.8 kΩresistor connected to the CS pin. Additional slope compensation may be added
by increasing the resistance between the current sense filter capacitor and the CS pin, thereby increasing the
voltage ramp created by the oscillator current ramp. Since the LM5021-2 is not capable of duty cycles greater
than 50%, there is no slope compensation feature in this device.
CURRENT LIMIT/CURRENT SENSE
The LM5021 provides a cycle-by-cycle over current protection feature. Current limit is triggered by an internal
current sense comparator threshold which is set at 500mV. If the CS pin voltage plus the slope compensation
voltage exceeds 500mV, the OUT pin output pulse will be immediately terminated.
An RC filter, located near the LM5021, is recommended for the CS pin to attenuate the noise coupled from the
power FET's gate to source. The CS pin capacitance is discharged at the end of each PWM clock cycle by an
internal switch. The discharge switch remains on for an additional 90ns leading edge blanking interval to
attenuate the current sense transient that occurs when the external power FET is turned on. In addition to
providing leading edge blanking, this circuit also improves dynamic performance by discharging the current
sense filter capacitor at the conclusion of every cycle.
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Product Folder Links: LM5021
RT = 6.63 x 109
2 x FSW
RT = 6.63 x 109
FSW
LM5021
SNVS359D –MAY 2005–REVISED MARCH 2013
www.ti.com
The LM5021 CS comparator is very fast, and may respond to short duration noise pulses. Layout considerations
are critical for the current sense filter and sense resistor. The capacitor associated with the CS filter must be
placed very close to the device and connected directly to the pins of the IC (CS and GND). If a current sense
transformer is used, both leads of the transformer secondary should be routed to the sense resistor, which
should also be located close to the IC. If a current sense resistor located in the power FET's source is used for
current sense, a low inductance resistor is required. In this case, all of the noise sensitive low current grounds
should be connected in common near the IC and then a single connection should be made to the power ground
(sense resistor ground point).
OSCILLATOR, SHUTDOWN and SYNC CAPABILITY
A single external resistor connected between RT and GND pins sets the LM5021 oscillator frequency. The
LM5021-2 device, with 50% maximum duty cycle, includes an internal flip-flop that divides the oscillator
frequency by two. This method produces a precise 50% maximum duty cycle limit. Because of this frequency
divider, the oscillator frequency of the LM5021-2 is actually twice the frequency of the gate drive output (OUT).
For the LM5021-1 device, the oscillator frequency and the operational output frequency are the same. To set a
desired output switching frequency (Fsw), the RT resistor can be calculated from:
LM5021-1:
(6)
LM5021-2:
(7)
The LM5021 can also be synchronized to an external clock. The external clock must have a higher frequency
than the free running oscillator frequency set by the RT resistor. The clock signal should be capacitively coupled
into the RT pin with a 100pF capacitor. A peak voltage level greater than 3.8 Volts at the RT pin is required for
detection of the sync pulse. The dc voltage across the RT resistor is internally regulated at 2 volts. Therefore, the
ac pulse superimposed on the RT resistor must have 1.8V or greater amplitude to successfully synchronize the
oscillator. The sync pulse width should be set between 15ns to 150ns by the external components. The RT
resistor is always required, whether the oscillator is free running or externally synchronized. The RT resistor
should be located very close to the device and connected directly to the pins of the LM5021 (RT and GND).
GATE DRIVER and MAX DUTY CYCLE LIMIT
The LM5021 provides a gate driver (OUT), which can source peak current of 0.3A and sink 0.7A. The LM5021 is
available in two duty-cycle limit options. The maximum output duty-cycle is typically 80% for the LM5021-1
option, and precisely equal to 50% for the LM5021-2 option. The maximum duty cycle function for the LM5021-2
is accomplished with an internal toggle flip-flop to ensure an accurate duty cycle limit. The internal oscillator
frequency of the LM5021-2 is therefore twice the switching frequency of the PWM controller (OUT pin).
The 80% maximum duty-cycle function for the LM5021-1 is determined by the internal oscillator. For the
LM5021-1 the internal oscillator frequency and the switching frequency of the PWM controller are the same.
SOFT-START
The soft-start feature allows the power converter to gradually reach the initial steady state operating point, thus
reducing start-up stresses and current surges. An internal 22 µA current source charges an external capacitor
connected to the SS pin. The capacitor voltage will ramp up slowly, limiting the COMP pin voltage and the duty
cycle of the output pulses. The soft-start capacitor is also used to generate the hiccup mode delay time when the
output of the switching power supply is continuously overloaded.
HICCUP MODE OVERLOAD CURRENT LIMITING
Hiccup mode is a method of protecting the power supply from over-heating and damage during an extended
overload condition. When the output fault is removed the power supply will automatically restart.
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Toverload = CSS x (VSS_OCV - VHC)
IOVCS = CSS x 0.6V
10 PA
Toff = CSS x (VHC - VRST)
IDTCS = CSS x (4.6V - 0.3V)
0.25 PA
LM5021
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SNVS359D –MAY 2005–REVISED MARCH 2013
Figure 10,Figure 11, and Figure 12 illustrate the equivalent circuit of the hiccup mode for LM5021 and the
relevant waveforms. During start-up and in normal operation, the external soft-start capacitor Css is pulled up by
a current source that delivers 22 µA to the SS pin capacitor. In normal operation, the soft-start capacitor
continues to charge and eventually reaches the saturation voltage of the current source (VSS_OCV, nominally
5.2V). During start-up the COMP pin voltage follows the SS capacitor voltage and gradually increases the peak
current delivered by the power supply. When the output of the switching power supply reaches the desired
voltage, the voltage feedback amplifier takes control of the COMP signal (via the opto-coupler). In normal
operation the COMP level is held at an intermediate voltage between 1.25V and 2.75V controlled by the voltage
regulation loop. When the COMP pin voltage is below 1.25V, the duty-cycle is zero. When the COMP level is
above 2.75V, the duty cycle will be limited by the 0.5V threshold of cycle-by-cycle current limit comparator.
If the output of the power supply is overloaded, the voltage regulation loop demands more current by increasing
the COMP pin control voltage. When the COMP pin exceeds the over voltage detection threshold (VOVLD,
nominally 4.6V), the SS capacitor Css will be discharged by a 10 µA overload detection timer current source,
IOVCS. If COMP remains above VOVLD long enough for the SS capacitor to discharge to the Hiccup mode
threshold (VHIC, nominally 4.6V), the controller enters the hiccup mode. The OUT pin is then latched low and the
SS capacitor discharge current source is reduced from 10 µA to 0.25 µA, the dead-time current source, IDTCS.
The SS pin voltage is slowly reduced until it reaches the Restart threshold (VRST, nominally 0.3V). Then a new
start-up sequence commences with 22 µA current source charging the capacitor CSS. The slow discharge of the
SS capacitor from the Hiccup threshold to the Restart threshold provides an extended off time that reduces the
overheating of components including diodes and MOSFETs due to the continuous overload. The off time during
the hiccup mode can be calculated from the following equation:
(8)
Example:
Toff = 808 ms, assuming the CSS capacitor value is 0.047 µF
Short duration intermittent overloads will not trigger the hiccup mode. The overload duration required to trigger
the hiccup response is set by the capacitor CSS, the 10 µA discharge current source and voltage difference
between the saturation level of the SS pin and the Hiccup mode threshold. Figure 12 shows the waveform of SS
pin with a short duration overload condition. The overload time required to enter the hiccup mode can be
calculated from the following equation:
(9)
Example:
Toverload = 2.82 ms, assuming the CSS capacitor value is 0.047 µF
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SS
COMP
Soft-Start Normal
operation Overload
Detection SMPS latched
OFF Soft-Start
+22 PA
-10 PA
-0.25 PA+22 PA
5.2V
4.6V
0.3V
SS
OUT
5.2V
QS
+
-
550 mV
0.25 PA
EN
DRIVER
PWM
22 PA
EN
5.2V
RESTART
COMPARATOR
-
+
0.3V
10 PA
EN
4.6V
OVERLOAD
DETECTION
-
+
R
HICCUP MODE
COMPARATOR
+
-
4.6V
COMP
LM5021
SNVS359D –MAY 2005–REVISED MARCH 2013
www.ti.com
Figure 10. Hiccup Mode Control
WHITE SPACE
WHITE SPACE
Figure 11. Waveform at SS and COMP Pin due to Continuous Overload
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Product Folder Links: LM5021
SS
COMP
+22 PA
5.2V
4.6V -10 PA
during over
load
+22 PA
after releasing the over load
LM5021
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SNVS359D –MAY 2005–REVISED MARCH 2013
Figure 12. Waveform at SS and COMP Pin due to Brief Overload
SKIP CYCLE OPERATION
During light load conditions, the efficiency of the switching power supply typically drops as the losses associated
with switching and operating bias currents of the converter become a significant percentage of the power
delivered to the load. The largest component of the power loss is the switching loss associated with the gate
driver and external MOSFET gate charge. Each PWM cycle consumes a finite amout of energy as the MOSFET
is turned on and then turned off. These switching losses are proportional to the frequency of operation. The Skip
Cycle function integrated within the LM5021 controller reduces the average switching frequency to reduce
switching losses and improve efficiency during light load conditions.
When a light load condition occurs, the COMP pin voltage is reduced by the voltage feedback loop to reduce the
peak current delivered by the controller. Referring to Figure 13, the PWM comparator input tracks the COMP pin
voltage through a 1.25V level shift circuit and a 3:1 resistor divider. As the COMP pin voltage falls, the input to
the PWM comparator falls proportionately. When the PWM comparator input falls to 125mV, the Skip Cycle
comparator detects the light load condition and disables output pulses from the controller. The controller
continues to skip switching cycles until the power supply output falls and the COMP pin voltage increases to
demand more output current. The number of cycles skipped will depend on the load and the response time of the
frequency compensation network. Eventually the COMP voltage will increase when the voltage loop requires
more current to sustain the regulated output voltage. When the PWM comparator input exceeds 130mV (5mV
hysteresis), normal fixed frequency switching resumes. Typical power supply designs will produce a short burst
of output pulses followed by a long skip cycle interval. The average switching frequency in the Skip Cycle mode
can be a small fraction of the normal operating frequency of the power supply.
The skip cycle mode of operation can be disabled by adding an offset voltage to the CS pin (refer to Figure 14).
A resistive divider connected to a regulated source, injecting a 125mV offset (minimum) on the CS pin, will force
the voltage at the PWM Comparator to be greater than 125 mV, disabling the Skip Cycle Comparator.
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 13
Product Folder Links: LM5021
LM5021
VIN
CS
OUT
Voffset > 125 mV
RSense
VCC
PWM
LOGIC
PWM
COMPARATOR
5.2V
COMP
SKIP CYCLE
COMPARATOR
+
-
+
-
1.25V
5k 2R
R
125 mV
+
-
1.8k
CLK
500 mV
CURRENT LIMIT
COMPARATOR
CS
LEADING EDGE
BLANKING
LM5021
SNVS359D –MAY 2005–REVISED MARCH 2013
www.ti.com
Figure 13. Skip Cycle Control
Figure 14. Disabling the Skip Cycle Mode
14 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated
Product Folder Links: LM5021
LM5021
www.ti.com
SNVS359D –MAY 2005–REVISED MARCH 2013
Typical Application Circuit
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 15
Product Folder Links: LM5021
LM5021
SNVS359D –MAY 2005–REVISED MARCH 2013
www.ti.com
REVISION HISTORY
Changes from Revision C (March 2013) to Revision D Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 15
16 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated
Product Folder Links: LM5021
PACKAGE OPTION ADDENDUM
www.ti.com 1-Nov-2013
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM5021MM-1/NOPB ACTIVE VSSOP DGK 8 1000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 21-1
LM5021MM-2 NRND VSSOP DGK 8 1000 TBD Call TI Call TI -40 to 125 21-2
LM5021MM-2/NOPB ACTIVE VSSOP DGK 8 1000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 21-2
LM5021MMX-1/NOPB ACTIVE VSSOP DGK 8 3500 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 21-1
LM5021MMX-2 NRND VSSOP DGK 8 3500 TBD Call TI Call TI -40 to 125 21-2
LM5021MMX-2/NOPB ACTIVE VSSOP DGK 8 3500 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 21-2
LM5021NA-1/NOPB ACTIVE PDIP P 8 40 Green (RoHS
& no Sb/Br) CU SN Level-1-NA-UNLIM -40 to 125 LM5021NA
-1
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
PACKAGE OPTION ADDENDUM
www.ti.com 1-Nov-2013
Addendum-Page 2
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM5021MM-1/NOPB VSSOP DGK 8 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1
LM5021MM-2 VSSOP DGK 8 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1
LM5021MM-2/NOPB VSSOP DGK 8 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1
LM5021MMX-1/NOPB VSSOP DGK 8 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1
LM5021MMX-2 VSSOP DGK 8 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1
LM5021MMX-2/NOPB VSSOP DGK 8 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 11-Oct-2013
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM5021MM-1/NOPB VSSOP DGK 8 1000 210.0 185.0 35.0
LM5021MM-2 VSSOP DGK 8 1000 210.0 185.0 35.0
LM5021MM-2/NOPB VSSOP DGK 8 1000 210.0 185.0 35.0
LM5021MMX-1/NOPB VSSOP DGK 8 3500 367.0 367.0 35.0
LM5021MMX-2 VSSOP DGK 8 3500 367.0 367.0 35.0
LM5021MMX-2/NOPB VSSOP DGK 8 3500 367.0 367.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 11-Oct-2013
Pack Materials-Page 2
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