MIC2101/02
38V, Synchronous Buck Controllers
Featuring Adaptive On-Time Control
Hyper Speed Control¥
¥
Family
Hyper Speed Control, Hyper Light Load, and Any Capacitor are trademarks of Micrel, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
November 13, 2013 Revision 2.0
General Description
The Micrel MIC2101/02 are constant-frequency,
synchronous buck controllers featuring a unique adaptive
ON-time control architecture. The MIC2101/02 operates
over an input supply range from 4.5V to 38V and can be
used to supply up to 15A of output current. The output
voltage is adjustable down to 0.8V with a guaranteed
accuracy of ±1%. The device operates with programmable
switching frequency from 200kHz to 600kHz.
Micrel’s Hyper Light Load™ architecture provides the same
high-efficiency and ultra-fast transient response as the Hyper
Speed Control architecture under the medium to heavy loads,
but also maintains high efficiency under light load conditions
by transitioning to variable frequency, discontinuous-mode
operation.
The MIC2101/02 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include under-voltage lockout to ensure proper operation
under power-sag conditions, internal soft-start to reduce
inrush current, fold-back current limit, “hiccup” mode short-
circuit protection and thermal shutdown.
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
Features
xHyper Speed Control architecture enables:
High Delta V operation (VIN = 38V and VOUT = 1.2V)
Any Capacitor¥stable
x4.5V to 38V input voltage
xAdjustable output voltage from 0.8 V to 24V (also limited
by duty cycle)
x200kHz to 600kHz, programmable switching frequency
xHyper Light Load Control (MIC2101)
xHyper Speed Control (MIC2102)
xEnable input and Power Good output
xBuilt-in 5V regulator for single-supply operation
xProgrammable current limit and fold-back “hiccup” mode
short-circuit protection
x5ms internal soft-start, internal compensation, and
thermal shutdown
xSupports safe start-up into a pre-biased output
x–40qC to +125qC junction temperature range
xAvailable in 16-pin 3mm x 3mm QFN package
Applications
xDistributed power systems
xNetworking/telecom Infrastructure
xPrinters, scanners, graphic cards, and video cards
Typical Application
MIC2101/02 Wide Input, Hyper Light Load Buck Converter
MIC2101 Efficiency (VIN = 12V)
vs. Output Current (MIC2101)
10
20
30
40
50
60
70
80
90
100
0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16
OUTPUT CURRENT (A)
EFFICIENCY (%)
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
f
SW
= 600kHz (CCM)
Micrel, Inc. MIC2101/02
November 13, 2013 2 Revision 2.0
Ordering Information
Part Number Switching
Frequency Features Package
Junction
Temperature
Range
Lead Finish
MIC2101YML 200kHz to 600kHz Hyper Light Load 16-Pin 3mm x 3mm QFN –40°C to +125°C Pb-Free
MIC2102YML 200kHz to 600kHz Hyper Speed Control 16-Pin 3mm x 3mm QFN –40°C to +125°C Pb-Free
Pin Configuration
16-Pin 3mm x 3mm QFN (ML)
(Top View)
Pin Description
Pin Number Pin Name Pin Function
1VDD
Internal +5V Linear Regulator 2XWSXW9''LVWKHLQWHUQDOVXSSO\EXVIRUWKHGHYLFH$ȝ)
ceramic capacitor from VDD to AGND is required for decoupling. In the applications with VIN <
+5.5V, VDD should be tied to VIN to by-pass the linear regulator.
2 PVDD 5V supply input for the low-side N-channel MOSFET driver, which can be tied to VDD externally.
$ȝ)FHUDPLFFDSDFLWRUIURP39''WR3*1'LVUHFRPPHQGHGIRUGHFRXSOLQJ
3ILIM
Current Limit Setting. Connect a resistor from SW to ILIM to set the over-current threshold for the
converter.
4DL
Low-Side Drive output. High-current driver output for external low-side MOSFET of a buck
converter. The DL driving voltage swings from ground to VDD. Adding a small resistor between
DL pin and the gate of the low-side N-channel MOSFET can slow down the turn-on and turn-off
speed of the MOSFET.
5PGND
Power Ground. PGND is the return path for the buck converter power stage. The PGND pin
connects to the sources of low-side N-Channel external MOSFET, the negative terminals of input
capacitors, and the negative terminals of output capacitors. The return path for the power ground
should be as small as possible and separate from the signal ground (AGND) return path.
6FREQ
Switching Frequency Adjust input. Tie this pin to VIN to operate at 600kHz and place a resistor
divider to reduce the frequency.
Micrel, Inc. MIC2101/02
November 13, 2013 3 Revision 2.0
Pin Configuration (Continued)
Pin Number Pin Name Pin Function
7DH
High-Side Drive Output. High-current driver output for external high-side MOSFET of a buck
converter. The DH driving voltage is floating on the switch node voltage (VSW). Adding a small
resistor between DH pin and the gate of the high-side N-channel MOSFET can slow down the
turn-on and turn-off speed of the MOSFET.
8SW
Switch Node and Current-Sense input. High current output driver return. The SW pin connects
directly to the switch node. Due to the high-speed switching on this pin, the SW pin should be
routed away from sensitive nodes. The SW pin also senses the current by monitoring the voltage
across the low-side MOSFET during OFF time. In order to sense the current accurately, connect
the low-side MOSFET drain to the SW pin using a Kelvin connection.
9, 11 NC No Connection.
10 BST
Voltage supply input for the high-side N-channel MOSFET driver, which can be powered by a
ERRWVWUDSSHGFLUFXLWFRQQHFWHGEHWZHHQ9''DQG6:XVLQJD6FKRWWN\GLRGHDQGDȝ)
ceramic capacitor. Adding a small resistor at BST pin can slow down the turn-on speed of the
high-side MOSFET.
12 AGND Signal ground for VDD and the control circuitry, which is connected to thermal pad electronically.
The signal ground return path should be separate from the power ground (PGND) return path.
13 FB
Feedback Input. Input to the transconductance amplifier of the control loop. The FB pin is
regulated to 0.8V. A resistor divider connecting the feedback to the output is used to set the
desired output voltage.
14 PG Power Good Output. Open drain output, an external pull-up resistor to VDD or external power
rails is required.
15 EN
Enable Input. A logic signal to enable or disable the buck converter operation. The EN pin is
CMOS compatible. Logic high enables the device, logic low shutdowns the regulator. In the
disable mode, the VDD supply current for the device is minimized to 0.7mA typically. Don not pull
EN pin to VDD/PVDD.
16 VIN 6XSSO\9ROWDJH7KH9,1RSHUDWLQJYROWDJHUDQJHLVIURP9WR9$ȝ)FHUDPLFFDSDFLWRU
from VIN to AGND is required for decoupling.
EP ePad Exposed Pad. Connect the EPAD to PGND plain on the PCB to improve the thermal
performance.
Micrel, Inc. MIC2101/02
November 13, 2013 4 Revision 2.0
Absolute Maximum Ratings(1)
VIN ................................................................ 0.3V to +40V
VDD, VPVDD........................................................ 0.3V to +6V
VSW, VFREQ, VILIM, VEN............................ 0.3V to (VIN +0.3V)
VBST to VSW ........................................................ 0.3V to 6V
VBST ................................................................ 0.3V to 46V
VPG .....................................................0.3V to (VDD + 0.3V)
VFB .....................................................0.3V to (VDD + 0.3V)
PGND to AGND............................................ 0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS).........................65qC to +150qC
Lead Temperature (soldering, 10s)............................ 260°C
ESD Rating(2)................................................ ESD Sensitive
Operating Ratings(3)
Supply Voltage (VIN).......................................... 4.5V to 38V
Enable Input (VEN) .................................................. 0V to VIN
VSW, VFREQ, VILIM, VEN ............................................. 0V to VIN
Junction Temperature (TJ) ........................ 40qC to +125qC
Junction Thermal Resistance
3mm x 3mm QFN-16 (TJA) ....................................50.8°C/W
3mm x 3mm QFN-16 (TJC) ......................25.3°C/W
Electrical Characteristics(4)
VIN = 12V, VOUT =1.2V; VBST –V
SW = 5V; TA= 25°C, unless noted. Bold values indicate &7J&
Parameter Condition Min. Typ. Max. Units
Power Supply Input
Input Voltage Range (VIN)(5)4.5 38 V
Quiescent Supply Current (MIC2101) VFB = 1.5V 400 750 μA
Quiescent Supply Current (MIC2102) VFB = 1.5V 2.1 3mA
Shutdown Supply Current SW unconnected, VEN = 0V 0.1 10 μA
VDD Supply
VDD Output Voltage VIN = 7V to 38V, IDD = 10mA 4.8 5.2 5.4 V
VDD UVLO Threshold VDD rising 3.8 4.2 4.6 V
VDD UVLO Hysteresis 400 mV
Load Regulation IDD = 0 to 40mA 0.6 2 3.6 %
Reference
Feedback Reference Voltage TJ= 25°C (±1.0%) 0.792 0.8 0.808 V
40°C TJC (±2%) 0.784 0.8 0.816
FB Bias Current VFB = 0.8V 5 500 nA
Enable Control
EN Logic Level High 1.8 V
EN Logic Level Low 0.6 V
EN Hysteresis 200 mV
EN Bias Current VEN = 12V 6 30 μA
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kin series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. Specification for packaged product only.
5. The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have low voltage VTH.
Micrel, Inc. MIC2101/02
November 13, 2013 5 Revision 2.0
Electrical Characteristics(4)(Continued)
VIN = 12V, VOUT = 1.2V; VBST –V
SW = 5V; TA= 25°C, unless noted. Bold values indicate &7J&
Parameter Condition Min. Typ. Max. Units
Oscillator
Switching Frequency VFREQ = VIN 400 600 750 kHz
VFREQ = 50%VIN 300
Maximum Duty Cycle 85 %
Minimum Duty Cycle VFB > 0.8V 0 %
Minimum Off-Time 140 200 260 ns
Soft-Start
Soft-Start time 5ms
Short-Circuit Protection
Current-Limit Threshold VFB = 0.79V 30 14 0mV
Short-Circuit Threshold VFB = 0V 23 79mV
Current-Limit Source Current VFB = 0.79V 60 80 100 μA
Short-Circuit Source Current VFB = 0V 27 37 47 μA
FET Drivers
DH, DL Output Low Voltage ISINK = 10mA 0.1 V
DH, DL Output High Voltage ISOURCE = 10mA
VPVDD 0.1V
or
VBST 0.1V
V
DH On-Resistance, High State 2.1 3.3
DH On-Resistance, Low State 1.8 3.3
DL On-Resistance, High State 1.8 3.3
DL On-Resistance, Low State 1.2 2.3
SW, BST Leakage Current 50 μA
Power Good (PG)
PG Threshold Voltage Sweep VFB from Low to High 85 90 95 %VOUT
PG Hysteresis Sweep VFB from High to Low 6 %VOUT
PG Delay Time Sweep VFB from Low to High 100 μs
PG Low Voltage VFB < 90% x VNOM, IPG = 1mA 70 200 mV
Thermal Protection
Over-Temperature Shutdown TJRising 160 °C
Over-Temperature Shutdown Hysteresis 7 °C
Micrel, Inc. MIC2101/02
November 13, 2013 6 Revision 2.0
Typical Characteristics
V
IN
Operating Supply Current
vs. Input Voltage (MIC2101)
0.00
0.10
0.20
0.30
0.40
0.50
0.60
0.70
0.80
0.90
1.00
4 9 14 19 24 29 34
INPUT VOLTAGE (V)
SUPPLY CURRENT (mA)
V
OUT
= 3.3V
I
OUT
= 0A
Output Regulation
vs. Input Voltage (MIC2101)
-1.0%
-0.8%
-0.6%
-0.4%
-0.2%
0.0%
0.2%
0.4%
0.6%
0.8%
1.0%
5 101520253035
INPUT VOLTAGE (V)
OUTPUT REGULATION (%)
V
OUT
= 3.3V
I
OUT
= 0A to 12A
V
IN
Operating Supply Current
vs. Input Voltage (MIC2101)
0.00
0.10
0.20
0.30
0.40
0.50
0.60
0.70
0.80
0.90
1.00
5 101520253035
INPUT VOLTAGE (V)
SUPPLY CURRENT (mA)
V
OUT
= 1.2V
I
OUT
= 0A
Output Regulation
vs. Input Voltage (MIC2101)
-1.0%
-0.8%
-0.6%
-0.4%
-0.2%
0.0%
0.2%
0.4%
0.6%
0.8%
1.0%
510
15 20 25 30 35
INPUT VOLTAGE (V)
OUTPUT REGULATION (%)
V
OUT
= 1.2V
I
OUT
= 0A to 12A
Feedback Voltage
vs. Input Voltage (MIC2101)
0.776
0.784
0.792
0.800
0.808
0.816
0.824
5 101520253035
INPUT VOLTAGE (V)
FEEDBACK VOLTAGE (V)
V
OUT
= 3.3V
I
OUT
= 0A
Output Voltage
vs. Input Voltage (MIC2101)
3.217
3.234
3.250
3.267
3.283
3.300
3.316
3.333
5 101520253035
INPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
V
OUT
= 3.3V
I
OUT
= 0A
Output Voltage
vs. Input Voltage (MIC2101)
1.196
1.198
1.200
1.202
1.204
1.206
1.208
1.210
1.212
5 101520253035
INPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
V
OUT
= 1.2V
I
OUT
= 0A
VIN Operating Supply Current
vs. Temperature (MIC2101)
0.00
0.10
0.20
0.30
0.40
0.50
0.60
0.70
0.80
0.90
1.00
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
SUPPLY CURRENT (mA)
V
IN
= 12V
V
OUT
= 3.3V
I
OUT
= 0A
Feedback Voltage
vs. Temperature (MIC2101)
0.792
0.796
0.800
0.804
0.808
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
FEEBACK VOLTAGE (V)
V
IN
= 12V
V
OUT
= 3.3V
I
OUT
= 0A
Micrel, Inc. MIC2101/02
November 13, 2013 7 Revision 2.0
Typical Characteristics (Continued)
Load Regulation
vs. Temperature (MIC2101)
-0.3%
-0.2%
-0.1%
0.0%
0.1%
0.2%
0.3%
0.4%
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
LOAD REGULATION (%)
V
IN
= 12V
V
OUT
= 3.3V
I
OUT
= 0 to 12A
Line Regulation
vs. Temperature (MIC2101)
-1.8%
-1.6%
-1.4%
-1.2%
-1.0%
-0.8%
-0.6%
-0.4%
-0.2%
0.0%
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
LINE REGULATION (%)
V
IN
= 5V to 38V
V
OUT
= 3.3V
I
OUT
= 0A
Feedback Voltage
vs. Output Current (MIC2101)
0.792
0.796
0.800
0.804
0.808
0123456789101112
OUTPUT CURRENT (A)
FEEDBACK VOLTAGE (V)
V
IN
= 12V
V
OUT
= 3.3V
Line Regulation
vs. Output Current (MIC2101)
-3.0%
-2.0%
-1.0%
0.0%
1.0%
2.0%
3.0%
0 1 2 3 4 5 6 7 8 9 10 11 12
OUTPUT CURRENT (A)
LINE REGULATION (%)
VIN = 5V to 38V
VOUT = 3.3V
Efficiency (VIN = 5V)
vs. Output Current (MIC2101)
0
10
20
30
40
50
60
70
80
90
100
0481216
OUTPUT CURRENT (A)
EFFICIENCY (%)
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
f
SW
= 600kHz (CCM)
Efficiency (V
IN
=12V)
vs. Output Current (MIC2101)
0
10
20
30
40
50
60
70
80
90
100
0246810121416
OUTPUT CURRENT (A)
EFFICIENCY (%)
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
fSW = 600kHz (CCM)
Efficiency (VIN = 18V)
vs. Output Current (MIC2101)
0
10
20
30
40
50
60
70
80
90
100
0 2 4 6 8 10 12 14 16
OUTPUT CURRENT (A)
EFFICIENCY (%)
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
f
SW
= 600kHz (CCM)
Efficiency (V
IN
= 24V)
vs. Output Current (MIC2101)
0
10
20
30
40
50
60
70
80
90
100
0246810121416
OUTPUT CURRENT (A)
EFFICIENCY (%)
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
f
SW
= 600kHz (CCM)
Efficiency (V
IN
= 38V)
vs. Output Current (MIC2101)
0
10
20
30
40
50
60
70
80
90
100
0 2 4 6 8 10121416
OUTPUT CURRENT (A)
EFFICIENCY (%)
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
f
SW
= 600kHz (CCM)
Micrel, Inc. MIC2101/02
November 13, 2013 8 Revision 2.0
Typical Characteristics (Continued)
V
IN
Operating Supply Current
vs. Input Voltage (MIC2102)
0
12
24
36
48
60
4 9 14 19 24 29 34 39
INPUT VOLTAGE (V)
SUPPLY CURRENT (mA)
V
OUT
= 3.3V
I
OUT
= 0A
f
SW
= 600kHz
Feedback Voltage
vs. Input Voltage (MIC2102)
0.792
0.796
0.800
0.804
0.808
4 9 14 19 24 29 34 39
INPUT VOLTAGE (V)
FEEDBACK VOLTAGE (V)
V
OUT
= 3.3V
I
OUT
= 0A
f
SW
= 600kHz
Output Regulation
vs. Input Voltage (MIC2102)
-1.0%
-0.8%
-0.6%
-0.4%
-0.2%
0.0%
0.2%
0.4%
0.6%
0.8%
1.0%
5 101520253035
INPUT VOLTAGE (V)
OUTPUT REGULATION (%)
V
OUT
= 3.3V
I
OUT
= 0A to 12A
f
SW
= 600kHz
Output Regulation
vs. Input Voltage (MIC2102)
-1.0%
-0.8%
-0.6%
-0.4%
-0.2%
0.0%
0.2%
0.4%
0.6%
0.8%
1.0%
5 101520253035
INPUT VOLTAGE (V)
OUTPUT REGULATION (%)
V
OUT
= 1.2V
I
OUT
= 0A to 12A
f
SW
= 600kHz
`
V
IN
Operating Supply Current
vs. Input Voltage (MIC2102)
0
12
24
36
48
60
5 101520253035
INPUT VOLTAGE (V)
SUPPLY CURRENT (mA)
V
OUT
= 1.2V
I
OUT
= 0A
f
SW
= 600kHz
VIN Operating Supply Current
vs. Temperature (MIC2102)
0
10
20
30
40
50
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
SUPPLY CURRENT (mA)
V
IN
= 12V
V
OUT
= 3.3V
I
OUT
= 0A
f
SW
= 600kHz
Feedback Voltage
vs. Temperature (MIC2102)
0.792
0.796
0.800
0.804
0.808
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
FEEBACK VOLTAGE (V)
V
IN
= 12V
V
OUT
= 3.3V
I
OUT
= 0A
Load Regulation
vs. Temperature (MIC2102)
-0.3%
-0.2%
-0.1%
0.0%
0.1%
0.2%
0.3%
0.4%
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
LOAD REGULATION (%)
V
IN
= 12V
V
OUT
= 3.3V
I
OUT
= 0A to 12A
f
SW
= 600kHz
Line Regulation
vs. Temperature (MIC2102)
-0.3%
-0.2%
-0.1%
0.0%
0.1%
0.2%
0.3%
0.4%
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
LINE REGULATION (%)
V
IN
= 5V to 38V
V
OUT
= 3.3V
I
OUT
= 0A
Micrel, Inc. MIC2101/02
November 13, 2013 9 Revision 2.0
Typical Characteristics (Continued)
Switching Frequency
vs. Output Current (MIC2102)
100
150
200
250
300
350
400
450
500
550
600
650
700
024681012
OUTPUT CURRENT (A)
SWITCHING FREQUENCY
(kHz)
V
IN
= 12V
V
OUT
= 3.3V
25°C
-40°C
125°C
Feedback Voltage
vs. Output Current (MIC2102)
0.792
0.796
0.800
0.804
0.808
0123456789101112
OUTPUT CURRENT (A)
FEEDBACK VOLTAGE (V)
V
IN
= 12V
V
OUT
= 3.3V
f
SW
= 600kHz
Line Regulation
vs. Output Current (MIC2102)
-0.3%
-0.2%
-0.1%
0.0%
0.1%
0.2%
0.3%
0123456789101112
OUTPUT CURRENT (A)
LINE REGULATION (%)
V
IN
= 5V to 38V
V
OUT
= 3.3V
V
DD
= 5V
f
SW
= 600kHz
Efficiency (VIN = 5V)
vs. Output Current (MIC2102)
0
10
20
30
40
50
60
70
80
90
100
0481216
OUTPUT CURRENT (A)
EFFICIENCY (%)
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
fSW = 600kHz
Efficiency (VIN = 12V)
vs. Output Current (MIC2102)
30
40
50
60
70
80
90
100
0481216
OUTPUT CURRENT (A)
EFFICIENCY (%)
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
f
SW
= 600kHz
Efficiency (V
IN
= 18V)
vs. Output Current (MIC2102)
0
10
20
30
40
50
60
70
80
90
100
0481216
OUTPUT CURRENT (A)
EFFICIENCY (%)
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
V
SW
= 600kHz
Efficiency (V
IN
= 24V)
vs. Output Current (MIC2102)
0
10
20
30
40
50
60
70
80
90
100
0481216
OUTPUT CURRENT (A)
EFFICIENCY (%)
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
f
SW
= 600kHz
Efficiency (VIN = 38V)
vs. Output Current (MIC2102)
0
10
20
30
40
50
60
70
80
90
100
0481216
OUTPUT CURRENT (A)
EFFICIENCY (%)
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
f
SW
= 600kHz
Micrel, Inc. MIC2101/02
November 13, 2013 10 Revision 2.0
Typical Characteristics (Continued)
Case Temperature*: The temperature measurement was taken at the hottest point on the MIC2101/02 case mounted on a 5 square inch PCB, see
Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting
components.
Die Temperature* (V
IN
= 5.0V)
vs. Output Current
0
20
40
60
80
0123456789101112
OUTPUT CURRENT (A)
DIE TEMPERATURE (°C)
V
IN
= 5.0V
V
OUT
= 3.3V
f
SW
= 600kHz
Die Temperature* (VIN = 12V)
vs. Output Current
0
20
40
60
80
100
0123456789101112
OUTPUT CURRENT (A)
DIE TEMPERATURE (°C)
V
IN
= 12V
V
OUT
= 3.3V
f
SW
= 600kHz
Die Temperature* (V
IN
= 24V)
vs. Output Current
0
20
40
60
80
100
120
0123456789101112
OUTPUT CURRENT (A)
DIE TEMPERATURE (°C)
V
IN
= 24V
V
OUT
= 3.3V
f
SW
= 600kHz
Die Temperature* (VIN = 38V)
vs. Output Current
0
20
40
60
80
100
120
140
160
0 1 2 3 4 5 6 7 8 9 10 11 12
OUTPUT CURRENT (A)
DIE TEMPERATURE (°C)
V
IN
= 38V
V
OUT
= 3.3V
f
SW
= 600kHz
Die Temperature* (V
IN
= 5.0V)
vs. Output Current
0
20
40
60
80
0123456789101112
OUTPUT CURRENT (A)
DIE TEMPERATURE (°C)
V
IN
= 5.0V
V
OUT
= 1.2V
f
SW
= 600kHz
Die Temperature* (V
IN
= 12V)
vs. Output Current
0
20
40
60
80
100
0123456789101112
OUTPUT CURRENT (A)
DIE TEMPERATURE (°C)
VIN = 12V
VOUT = 1.2V
fSW = 600kHz
`
Die Temperature* (V
IN
= 24V)
vs. Output Current
0
20
40
60
80
100
120
0 1 2 3 4 5 6 7 8 9 10 11 12
OUTPUT CURRENT (A)
DIE TEMPERATURE (°C)
V
IN
= 24V
V
OUT
= 1.2V
f
SW
= 600kHz
Die Temperature* (VIN = 38V)
vs. Output Current
0
20
40
60
80
100
120
140
160
0123456789101112
OUTPUT CURRENT (A)
DIE TEMPERATURE (°C)
V
IN
= 38V
V
OUT
= 1.2V
f
SW
= 600kHz
Micrel, Inc. MIC2101/02
November 13, 2013 11 Revision 2.0
Typical Characteristics (Continued)
V
IN
Shutdown Current
vs. Input Voltage
0
6
12
18
24
30
4 9 14 19 24 29 34 39
INPUT VOLTAGE (V)
SHUTDOWN CURRENT (uA)
V
EN
= 0V
V
DD
Voltage
vs. Input Voltage
0
2
4
6
8
10
4 9 14 19 24 29 34 39
INPUT VOLTAGE (V)
V
DD
VOLTAGE (V)
V
OUT
= 3.3V
f
SW
= 600kHz
I
DD
= 10mA
I
DD
= 40mA
Enable Threshold
vs. Input Voltage
0.00
0.10
0.20
0.30
0.40
0.50
0.60
0.70
0.80
0.90
1.00
1.10
1.20
1.30
1.40
1.50
4 9 14 19 24 29 34 39
INPUT VOLTAGE (V)
ENABLE THRESHOLD (V)
HYST
RISING
FALLING
Switching Frequency
vs. Input Voltage
200
250
300
350
400
450
500
550
600
650
700
750
800
5 101520253035
INPUT VOLTAGE (V)
SWITCHING FREQUENCY
(kHz)
V
OUT
= 3.3V
I
OUT
= 2A
Output Peak Current Limit
vs. Input Voltage
0
5
10
15
20
25
5 101520253035
INPUT VOLTAGE (V)
CURRENT LIMIT (A)
V
OUT
= 3.3V
f
SW
= 600kHz
Switching Frequency
vs. Input Voltage
200
250
300
350
400
450
500
550
600
650
700
750
800
5 101520253035
INPUT VOLTAGE (V)
SWITCHING FREQUENCY (kHz)
V
OUT
= 1.2V
I
OUT
= 2A
V
IN
Shutdown Current
vs. Temperature
0
3
6
9
12
15
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
SHUTDOWN CURRENT (μA)
VIN =12V
VEN = 0V
IOUT = 0A
V
DD
Voltage
vs. Temperature
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
V
DD
Voltage (V)
V
IN
= 12V
I
OUT
= 0A
I
DD
= 10mA
I
DD
= 40mA
VDD UVLO Threshold
vs. Temperature
2.0
2.5
3.0
3.5
4.0
4.5
5.0
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
VDDTHRESHOLD (V)
VIN =12V
IOUT = 0A
FALLING
RISING
Micrel, Inc. MIC2101/02
November 13, 2013 12 Revision 2.0
Typical Characteristics (Continued)
Output Peak Current Limit
vs. Temperature
0
5
10
15
20
25
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
CURRENT LIMIT (A)
V
IN
=12V
V
OUT
= 3.3V
f
SW
= 600kHz
EN Bias Current
vs. Temperature
0
1
2
3
4
5
6
7
8
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
EN BIAS CURRENT (μA)
V
IN
=12V
V
EN
= 0V
Enable Threshold
vs. Temperature
0.5
0.8
1.1
1.4
1.7
2.0
-50 -25 0 25 50 75 100 125
TEMPERATURE (°C)
ENABLE THRESHOLD (V)
V
IN
= 12V
RI
S
IN
G
FALLING
Micrel, Inc. MIC2101/02
November 13, 2013 13 Revision 2.0
Functional Characteristics
Micrel, Inc. MIC2101/02
November 13, 2013 14 Revision 2.0
Functional Characteristics (Continued)
Micrel, Inc. MIC2101/02
November 13, 2013 15 Revision 2.0
Functional Characteristics (Continued)
Micrel, Inc. MIC2101/02
November 13, 2013 16 Revision 2.0
Functional Characteristics (Continued)
Micrel, Inc. MIC2101/02
November 13, 2013 17 Revision 2.0
Functional Characteristics (Continued)
Micrel, Inc. MIC2101/02
November 13, 2013 18 Revision 2.0
Functional Characteristics (Continued)
Micrel, Inc. MIC2101/02
November 13, 2013 19 Revision 2.0
Functional Diagram
Note:
ZC Detection* MIC2101 Only.
Figure 1. MIC2101/02 Functional Diagram
Micrel, Inc. MIC2101/02
November 13, 2013 20 Revision 2.0
Functional Description
The MIC2101/02 are adaptive on-time synchronous buck
controllers built for high-input voltage to low output
voltage applications. It is designed to operate over a
wide input voltage range from, 4.5V to 38V and the
output is adjustable with an external resistive divider. An
adaptive on-time control scheme is employed to obtain a
constant switching frequency and to simplify the control
compensation. Over-current protection is implemented
by sensing low-side MOSFET’s RDS(ON). The device
features internal soft-start, enable, UVLO, and thermal
shutdown.
Theory of Operation
Figure 1 illustrates the block diagram of the MIC2101/02.
The output voltage is sensed by the MIC2101/02
feedback pin FB via the voltage divider R1 and R2, and
compared to a 0.8V reference voltage VREF at the error
comparator through a low-gain transconductance (gm)
amplifier. If the feedback voltage decreases and the
amplifier output is below 0.8V, then the error comparator
will trigger the control logic and generate an ON-time
period. The ON-time period length is predetermined by
the “Fixed tON Estimator” circuitry:
SWIN
OUT
ED)ON(ESTIMAT fV
V
u
t
Eq. 1
where VOUT is the output voltage, VIN is the power stage
input voltage, and fSW is the switching frequency.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gmamplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
200ns, the MIC2101/02 control logic will apply the
tOFF(min) instead. tOFF(min) is required to maintain enough
energy in the boost capacitor (CBST) to drive the high-
side MOSFET.
The maximum duty cycle is obtained from the 200ns
tOFF(min):
SS
OFF(MIN)S
MAX
t
200ns
1
t
tt
D
Eq. 2
where tS= 1/fSW. It is not recommended to use
MIC2101/02 with a OFF-time close to tOFF(min) during
steady-state operation.
The adaptive ON-time control scheme results in a
constant switching frequency in the MIC2101/02. The
actual ON-time and resulting switching frequency will
vary with the different rising and falling times of the
external MOSFETs. Also, the minimum tON results in a
lower switching frequency in high VIN to VOUT
applications. During load transients, the switching
frequency is changed due to the varying OFF-time.
To illustrate the control loop operation, we will analyze
both the steady-state and load transient scenarios. For
easy analysis, the gain of the gmamplifier is assumed to
be 1. With this assumption, the inverting input of the
error comparator is the same as the feedback voltage.
Figure 2 shows the MIC2101/02 control loop timing
during steady-state operation. During steady-state, the
gmamplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple plus injected
voltage ripple, to trigger the ON-time period. The ON-
time is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
Figure 2. MIC2101/02 Control Loop Timing
Micrel, Inc. MIC2101/02
November 13, 2013 21 Revision 2.0
Figure 3a shows the operation of the MIC2101/02 during
a load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC2101/02
converter.
Figure 3a. MIC2101/02 Load Transient Response
Unlike true current-mode control, the MIC2101/02 uses
the output voltage ripple to trigger an ON-time period.
The output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough.
In order to meet the stability requirements, the
MIC2101/02 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gmamplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV over full input voltage range. If a low-ESR
output capacitor is selected, then the feedback voltage
ripple may be too small to be sensed by the gmamplifier
and the error comparator. Also, the output voltage ripple
and the feedback voltage ripple are not necessarily in
phase with the inductor current ripple if the ESR of the
output capacitor is very low. In these cases, ripple
injection is required to ensure proper operation. Please
refer to “Ripple Injection” subsection in Application
Information for more details about the ripple injection
technique.
Discontinuous Mode (MIC2101 only)
In continuous mode, the inductor current is always
greater than zero; however, at light loads the MIC2101 is
able to force the inductor current to operate in
discontinuous mode. Discontinuous mode is where the
inductor current falls to zero, as indicated by trace (IL)
shown in Figure 3b. During this period, the efficiency is
optimized by shutting down all the non-essential circuits
and minimizing the supply current. The MIC2101 wakes
up and turns on the high-side MOSFET when the
feedback voltage VFB drops below 0.8V.
The MIC2101 has a zero crossing comparator (ZC
Detection) that monitors the inductor current by sensing
the voltage drop across the low-side MOSFET during its
ON-time. If the VFB > 0.8V and the inductor current goes
slightly negative, then the MIC2101 automatically
powers down most of the IC circuitry and goes into a
low-power mode.
Once the MIC2101 goes into discontinuous mode, both
LSD and HSD are low, which turns off the high-side and
low-side MOSFETs. The load current is supplied by the
output capacitors and VOUT drops. If the drop of VOUT
causes VFB to go below VREF, then all the circuits will
wake up into normal continuous mode. First, the bias
currents of most circuits reduced during the
discontinuous mode are restored, then a tON pulse is
triggered before the drivers are turned on to avoid any
possible glitches. Finally, the high-side driver is turned
on. Figure 3b shows the control loop timing in
discontinuous mode.
Figure 3b. MIC2101 Control Loop Timing
(Discontinuous Mode)
During discontinuous mode, the bias current of most
circuits are reduced. As a result, the total power supply
Micrel, Inc. MIC2101/02
November 13, 2013 22 Revision 2.0
current during discontinuous mode is only about 400ȝA,
allowing the MIC2101 to achieve high efficiency in light
load applications.
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC2101/02 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in about 6ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a stair-
case VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
Current Limit
The MIC2101/02 uses the RDS(ON) and external resistor
connected from ILIM pin to SW node to decides the
current limit.
Figure 4. MIC2101/02 Current Limiting Circuit
In each switching cycle of the MIC2101/02 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. The sensed voltage V(ILIM) is
compared with the power ground (PGND) after a
blanking time of 150nS. In this way the drop voltage over
the resistor RCL (VCL) is compared with the drop over the
bottom FET generating the short current limit.
The small capacitor (CCL) connected from ILIM pin to
PGND filters the switching node ringing during the off
time allowing a better short limit measurement. The time
constant created by RCL and CCL should be much less
than the minimum off time.
The VCL drop allows programming of short limit through
the value of the resistor (RCL), If the absolute value of the
voltage drop on the bottom FET is greater than VCL’ in
that case the V(ILIM) is lower than PGND and a short
circuit event is triggered. A hiccup cycle to treat the short
event is generated. The hiccup sequence including the
soft start reduces the stress on the switching FETs and
protects the load and supply for severe short conditions.
The short circuit current limit can be programmed by
using the formula illustrated in Equation 3:
CL
CLDS(ON)PPCLIM
CL I
VR0.5)ǻ(I
Ruu
Eq. 3
Where ICLIM = Desired current limit
ǻPP = Inductor current peak-to-peak
RDS (ON) = On-resistance of low-side power MOSFET
VCL = Current-limit threshold, the typical absolute value is
14mV in Electrical Characteristic table
ICL = Current-limit source current, the typical value is
80μA in Electrical Characteristic table.
In case of hard short, the short limit is folded down to
allow an indefinite hard short on the output without any
destructive effect. It is mandatory to make sure that the
inductor current used to charge the output capacitance
during soft start is under the folded short limit, otherwise
the supply will go in hiccup mode and may not be
finishing the soft start successfully.
The MOSFET RDS(ON) varies 30% to 40% with
temperature; therefore, it is recommended to add a 50%
margin to ICL in the above equation to avoid false current
limiting due to increased MOSFET junction temperature
rise. It is also recommended to connect SW pin directly
to the drain of the low-side MOSFET to accurately sense
the MOSFETs RDS(ON).
Micrel, Inc. MIC2101/02
November 13, 2013 23 Revision 2.0
MOSFET Gate Drive
The MIC2101/02 high-side drive circuit is designed to
switch an N-Channel MOSFET. Figure 1 shows a
bootstrap circuit, consisting of D1 (a Schottky diode is
recommended) and CBST. This circuit supplies energy to
the high-side drive circuit. Capacitor CBST is charged
while the low-side MOSFET is on and the voltage on the
SW pin is approximately 0V. When the high-side
MOSFET driver is turned on, energy from CBST is used to
turn the MOSFET on. As the high-side MOSFET turns
on, the voltage on the SW pin increases to
approximately VIN. Diode D1 is reverse biased and CBST
floats high while continuing to keep the high-side
MOSFET on. The bias current of the high-side driver is
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the gate voltage with minimal droop for the power stroke
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turned back on, CBST is recharged through D1. A small
resistor RG, which is in series with CBST, can be used to
slow down the turn-on time of the high-side N-channel
MOSFET.
The drive voltage is derived from the VDD supply voltage.
The nominal low-side gate drive voltage is VDD and the
nominal high-side gate drive voltage is approximately
VDD –V
DIODE, where VDIODE is the voltage drop across
D1. An approximate 30ns delay between the high-side
and low-side driver transitions is used to prevent current
from simultaneously flowing unimpeded through both
MOSFETs.
Micrel, Inc. MIC2101/02
November 13, 2013 24 Revision 2.0
Application Information
Setting the Switching Frequency
The MIC2101/02 are adjustable-frequency, synchronous
buck controllers featuring a unique adaptive on-time
control architecture. The switching frequency can be
adjusted between 200kHz and 600kHz by changing the
resistor divider network consisting of R19 and R20.
Figure 5. Switching Frequency Adjustment
The following formula gives the estimated switching
frequency:
R20R19
R20
ff OSW_ADJ
u
Eq. 4
Where fO= Switching Frequency when R19 is 100k and
R20 being open, fOis typically 600kHz. For more precise
setting, it is recommended to use the following graph:
Figure 6. Switching Frequency vs. R20
MOSFET Selection
The MIC2101/02 controllers work from input voltages of
4.5V to 38V and has internal 5V VDD LDO. This internal
VDD LDO provides power to turn the external N-Channel
power MOSFETs for the high-side and low-side
switches. For applications where VDD < 5V, it is
necessary that the power MOSFETs used are sub-logic
level and are in full conduction mode for VGS of 2.5V. For
applications when VDD > 5V; logic-level MOSFETs,
whose operation is specified at VGS = 4.5V must be
used.
There are different criteria for choosing the high-side and
low-side MOSFETs. These differences are more
significant at lower duty cycles. In such an application,
the high-side MOSFET is required to switch as quickly
as possible to minimize transition losses, whereas the
low-side MOSFET can switch slower, but must handle
larger RMS currents. When the duty cycle approaches
50%, the current carrying capability of the high-side
MOSFET starts to become critical.
It is important to note that the on-resistance of a
MOSFET increases with increasing temperature. A 75°C
rise in junction temperature will increase the channel
resistance of the MOSFET by 50% to 75% of the
resistance specified at 25°C. This change in resistance
must be accounted for when calculating MOSFET power
dissipation and in calculating the value of current limit.
Total gate charge is the charge required to turn the
MOSFET on and off under specified operating conditions
(VDS and VGS). The gate charge is supplied by the
MIC2101/02 gate-drive circuit. At 600kHz switching
frequency, the gate charge can be a significant source of
power dissipation in the MIC2101/02. At low output load,
this power dissipation is noticeable as a reduction in
efficiency. The average current required to drive the
high-side MOSFET is:
SWGSIDE]-G[HIGH
fQ(AVG)Iu
Eq. 5
where:
IG[HIGHSIDE](avg) = Average high-side MOSFET gate
current
QG= Total gate charge for the high-side MOSFET taken
from the manufacturer’s data sheet for VGS = VDD.
fSW = Switching Frequency
The low-side MOSFET is turned on and off at VDS = 0
because an internal body diode or external freewheeling
diode is conducting during this time. The switching loss
for the low-side MOSFET is usually negligible. Also, the
gate-drive current for the low-side MOSFET is more
accurately calculated using CISS at VDS = 0 instead of
gate charge.
Switching Frequency
0.00
100.00
200.00
300.00
400.00
500.00
600.00
700.00
10.00 100.00 1000.00 10000.00
R20 (k Ohm)
SW FREQ (kHz)
R19 = 100k, I
OUT
=12A
VIN = 12V
VIN =38V
Micrel, Inc. MIC2101/02
November 13, 2013 25 Revision 2.0
For the low-side MOSFET:
SWGSISSSIDE]-G[LOW f VC(AVG)Iuu
Eq. 6
Since the current from the gate drive comes from the
VDD, the power dissipated in the MIC2101/02 due to gate
drive is:
(AVG))I
(AVG)(I VP
SIDE]-G[LOW
SIDE]-G[HIGHDDGATEDRIVE
u
Eq. 7
A convenient figure of merit for switching MOSFETs is
the on resistance times the total gate charge RDS(ON) ×
QG. Lower numbers translate into higher efficiency. Low
gate-charge logic-level MOSFETs are a good choice for
use with the MIC2101/02. Also, the RDS(ON) of the low-
side MOSFET will determine the current-limit value.
Please refer to “Current Limit” subsection is Functional
Description for more details.
Parameters that are important to MOSFET switch
selection are:
xVoltage rating
xOn-resistance
xTotal gate charge
The voltage ratings for the high-side and low-side
MOSFETs are essentially equal to the power stage input
voltage VHSD. A safety factor of 20% should be added to
the VDS(MAX) of the MOSFETs to account for voltage
spikes due to circuit parasitic elements.
The power dissipated in the MOSFETs is the sum of the
conduction losses during the on-time (PCONDUCTION) and
the switching losses during the period of time when the
MOSFETs turn on and off (PAC).
ACCONDUCTIONSW
PPP
Eq.8
DS(ON)
2
SW(RMS)CONDUCTION RIP u
Eq. 9
AC(on))AC(off
AC
P
PP
Eq. 10
where:
RDS(ON) = On-resistance of the MOSFET switch
D = Duty Cycle = VOUT / VHSD
Making the assumption that the turn-on and turn-off
transition times are equal; the transition times can be
approximated by:
G
HSDOSSINISS
TI
VCVC
tuu
Eq.11
where:
CISS and COSS are measured at VDS = 0
IG= Gate-drive current
The total high-side MOSFET switching loss is:
SWTPKDHSDAC ftI)V(VPuuu
Eq. 12
where:
tT= Switching transition time
VD= Body diode drop (0.5V)
fSW = Switching Frequency
The high-side MOSFET switching losses increase with
the switching frequency and the input voltage VHSD. The
low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor.
A good compromise between size, loss and cost is to set
the inductor ripple current to be equal to 20% of the
maximum output current.
Micrel, Inc. MIC2101/02
November 13, 2013 26 Revision 2.0
The inductance value is calculated by Equation 13:
OUT(MAX)swIN(MAX)
OUTIN(MAX)OUT
I20%fV
)V(VV
Luuu
u
Eq. 13
where:
fSW = Switching frequency
20% = Ratio of AC ripple current to DC output current
VIN(MAX) = Maximum power stage input voltage
The peak-to-peak inductor current ripple is:
LfV
)V(VV
ǻ,
swIN(MAX)
OUTIN(MAX)OUT
L(PP)
uu
u
Eq. 14
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
IL(pk) =IOUT(MAX) 0.5
u
ǻIL(PP) Eq. 15
The RMS inductor current is used to calculate the I2R
losses in the inductor.
12
ǻ,
II
2
L(PP)
2
OUT(MAX)L(RMS)
Eq. 16
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC2101/02 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor.
Copper loss in the inductor is calculated by Equation 17:
PINDUCTOR(Cu) = IL(RMS)
2uRWINDING Eq. 17
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
PWINDING(Ht) = RWINDING(20°C) u
(1 + 0.0042 × (TH–T
20°C)) Eq. 18
where:
TH= temperature of wire under full load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
Output Capacitor Selection
The type of the output capacitor is usually determined by
its equivalent series resistance (ESR). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are tantalum, low-ESR aluminum electrolytic, OS-
CON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view. The maximum value of ESR is
calculated:
L(PP)
OUT(pp)
Cǻ,
ǻ9
ESR OUT dEq. 19
where:
ǻVOUT(pp) = peak-to-peak output voltage ripple
ǻ,L(PP) = peak-to-peak inductor current ripple
Micrel, Inc. MIC2101/02
November 13, 2013 27 Revision 2.0
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 20:

2
CL(PP)
2
SWOUT
L(PP)
OUT(pp) OUT
ESRǻ,
8fC
ǻ,
ǻ9 u
¸
¸
¹
·
¨
¨
©
§
uu
Eq. 20
where:
D = duty cycle
COUT = output capacitance value
fsw = switching frequency
As described in the “Theory of Operation” subsection in
Functional Description, the MIC2101/02 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly.
Also, the output voltage ripple should be in phase with
the inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 21:
12
ǻ,
I
L(PP)
(RMS)C
OUT
Eq. 21
The power dissipated in the output capacitor is:
OUTOUTOUT C
2
(RMS)C)DISS(C ESRIP u
Eq. 22
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
ǻVIN = IL(pk) × ESRCIN Eq. 23
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
D)(1DII
OUT(max)CIN(RMS)
uu|
Eq. 24
The power dissipated in the input capacitor is:
PDISS(CIN) = ICIN(RMS)
2×ESR
CIN Eq. 25
Voltage Setting Components
The MIC2101/02 requires two resistors to set the output
voltage as shown in Figure 7:
Figure 7. Voltage-Divider Configuration
Micrel, Inc. MIC2101/02
November 13, 2013 28 Revision 2.0
The output voltage is determined by the equation:
)
R2
R1
(1VV FBOUT u
Eq. 26
where, VFB = 0.8V. A typical value of R1 can be between
Nȍ DQGNȍ,I5LVWRRODUJH, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small in value, it will decrease the efficiency of the power
supply, especially at light loads. Once R1 is selected, R2
can be calculated using:
FBOUT
FB
VV
R1V
R2
u
Eq. 27
Ripple Injection
The VFB ripple required for proper operation of the
MIC2101/02 gmamplifier and error comparator is 20mV
to 100mV. However, the output voltage ripple is
generally designed as 1% to 2% of the output voltage.
For a low output voltage, such as a 1V, the output
voltage ripple is only 10mV to 20mV, and the feedback
voltage ripple is less than 20mV. If the feedback voltage
ripple is so small that the gmamplifier and error
comparator can’t sense it, then the MIC2101/02 will lose
control and the output voltage is not regulated. In order
to have some amount of VFB ripple, a ripple injection
method is applied for low output voltage ripple
applications.
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1. Enough ripple at the feedback voltage due to the
large ESR of the output capacitors.
As shown in Figure 8a, the converter is stable
without any ripple injection. The feedback voltage
ripple is:
(pp)
LCFB(pp) ǻ,ESR
R2R1
R2
ǻ9
OUT
uu
Eq. 28
where ǻ,L(pp) is the peak-to-peak value of the
inductor current ripple.
2. Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin
through a feedforward capacitor Cff in this situation,
as shown in Figure 8b. The typical Cff value is
between 1nF and 100nF. With the feedforward
capacitor, the feedback voltage ripple is very close
to the output voltage ripple:
(pp)
LFB(pp)
ǻ,ESRǻ9 u|
Eq. 29
3. Virtually no ripple at the FB pin voltage due to the
very-low ESR of the output capacitors:
Figure 8a. Enough Ripple at FB
Figure 8b. Inadequate Ripple at FB
Figure 8c. Invisible Ripple at FB
Micrel, Inc. MIC2101/02
November 13, 2013 29 Revision 2.0
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor RINJ and a
capacitor Cinj, as shown in Figure 8c. The injected ripple
is:
W
u
uuuu
SW
divINFB(pp) f
1
D)-(1DKVǻ9 Eq.30
R1//R2R
R1//R2
K
INJ
div
Eq. 31
where:
VIN = Power stage input voltage
D = Duty cycle
fSW = Switching frequency
IJ= (R1//R2//Rinj)uCff
In Equations 30 and 32, it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
T
f
1
SW

u
WW
Eq. 32
,I WKH YROWDJH GLYLGHU UHVLVWRUV 5DQG 5 DUH LQ WKH Nȍ
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor CINJ is used in order to be considered as short
for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
Q)LI5DQG5DUHLQNȍUDQJH
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 35:
D)(1D
f
V
ǻ9
KSW
IN
FB(pp)
div u
u
u
W
Eq. 33
Then the value of RINJ is obtained as:
1)
K
1
((R1//R2)R
div
INJ
u
Eq. 34
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
Micrel, Inc. MIC2101/02
November 13, 2013 30 Revision 2.0
PCB Layout Guidelines
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC2101/02 converter.
IC
xThe 4.7μF ceramic capacitors, which are connected
to the VDD and PVDD pins, must be located right at
the IC. The VDD pin is very noise sensitive and
placement of the capacitor is very critical. Use wide
traces to connect to the VDD, PVDD and AGND,
PGND pins respectively.
xThe signal ground pin (AGND) must be connected
directly to the ground planes. Do not route the
AGND pin to the PGND pin on the top layer.
xPlace the IC close to the point of load (POL).
xUse fat traces to route the input and output power
lines.
xSignal and power grounds should be kept separate
and connected at only one location.
Input Capacitor
xPlace the input capacitor next.
xPlace the input capacitors on the same side of the
board and as close to the MOSFETs as possible.
xPlace several vias to the ground plane close to the
input capacitor ground terminal.
xUse either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
xDo not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
xIf a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
xIn “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over-
voltage spike seen on the input supply with power is
suddenly applied.
RC Snubber
xPlace the RC snubber on the same side of the board
and as close to the SW pin as possible.
Inductor
xKeep the inductor connection to the switch node
(SW) short.
xDo not route any digital lines underneath or close to
the inductor.
xKeep the switch node (SW) away from the feedback
(FB) pin.
xThe SW pin should be connected directly to the
drain of the low-side MOSFET to accurate sense the
voltage across the low-side MOSFET.
xTo minimize noise, place a ground plane underneath
the inductor.
Output Capacitor
xUse a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
xPhase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
xThe feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high-current
load trace can degrade the DC load regulation.
MOSFETs
xLow-side MOSFET gate drive trace (DL pin to
MOSFET gate pin) must be short and routed over a
ground plane. The ground plane should be the
connection between the MOSFET source and
PGND.
xChose a low-side MOSFET with a high CGS/CGD ratio
and a low internal gate resistance to minimize the
effect of dv/dt inducted turn-on.
xDo not put a resistor between the Low-side
MOSFET gate drive output and the gate.
xUse a 4.5V VGS rated MOSFET. Its higher gate
threshold voltage is more immune to glitches than a
2.5V or 3.3V rated MOSFET. MOSFETs that are
rated for operation at less than 4.5V VGS should not
be used.
Micrel, Inc. MIC2101/02
November 13, 2013 31 Revision 2.0
Evaluation Board Schematic
Figure 9. Schematic of MIC2101/02 Evaluation Board
(J1, J9, J12, R14, and R21 are for testing purposes)
Micrel, Inc. MIC2101/02
November 13, 2013 32 Revision 2.0
Bill of Materials
Item Part Number Manufacturer Description Qty
C1 EEU-FC1J221S Panasonic(6)220μF Aluminum Capacitor, 63V 1
C2, C3, C4 12105C225KAT2A AVX(7)
2.2μF/50V Ceramic Capacitor, X7R, Size 1210 3
C3225X7R1H225K TDK(8)
C14
GRM32ER60J107ME20L Murata(9)
100μF/6.3V Ceramic Capacitor, X7R, Size 1210 112106D107MAT2A AVX
C3225X5ROJ107M TDK
C6, C16, C10
GRM188R71H104KA93D Murata
0.1μF/50V Ceramic Capacitor, X7R, Size 0603 306035C104KAT2A AVX
C1608X7R1H104K TDK
C7, C17
GRM188R60J475KE19D Murata
4.7μF/6.3V Ceramic Capacitor, X7R, Size 0603 206036D475KAT2A AVX
C1608X5R0J475K TDK
C8
GRM188R70J105KA01D Murata
1μF/6.3V Ceramic Capacitor, X7R, Size 0603 1
06036C105KAT2A AVX
C1608X5R0J105K TDK
C9 GRM21BR72A474KA73 Murata 0.47μF/100V,X7R,0805 1
08051C474KAT2A AVX
C11
GRM188R71H102KA01D Murata
1nF/50V Cermiac Capacitor, X7R, Size 0603 106035C102KAT2A AVX
C1608X7R1H102K TDK
C12
GRM188R71H472MA01D Murata
4.7nF/50V Cermiac Capacitor, X7R, Size 0603 106035C472KAT2A AVX
C1608X7R1H472K TDK
C13 6SEPC470MX Sanyo(10)470μF/6.3V, 7m:, OSCON 1
6SEPC470M Sanyo 470μF/6.3V, 7m:, OSCON
C15 (OPEN) 6TPB470M Sanyo 470μF/6.3V, POSCAP
C5 (OPEN) GRM32ER60J107ME20L Murata 100μF/6.3V Ceramic Capacitor, X7R, Size 1210
C18 GRM1885C1H150JA01D Murata 15pF, 50V, 0603, NPO 1
06035A150JAT2A AVX
D1 BAT46W-TP MCC(11)100V Small Signal Schottky Diode, SOD123 1
L1 CDEP147NP- 1R5M-95 Sumida(12)1.5μH, 27/22Asat, 20Arms for 40C rise 1
Q1, Q3 BSC067N06LS3 Infineon(13)MOSFET, N-CH, Power SO-8 2
Notes:
6. Panasonic: www.panasonic.com.
7. AVX: www.avx.com
8. TDK: www.tdk.com.
9. Murata: www.murata.com.
10. Sanyo: www.sanyo.com.
11. MCC.: www.mccsemi.com.
12. Sumida: www.sumida.com
13. Infineon: www.infineon.com.
Micrel, Inc. MIC2101/02
November 13, 2013 33 Revision 2.0
Bill of Materials (Continued)
Item Part Number Manufacturer Description Qty.
Q2, Q4 (OPEN)
R1 CRCW060310K0FKEA Vishay Dale(14)10kȍResistor, Size 0603, 1% 1
R2, R23 CRCW08051R21FKEA Vishay Dale ȍ5HVLVWRU6L]H 2
R3 CRCW06035K23FKEA Vishay Dale 5.23K,1%,1/10W,0603. 1
R4 CRCW060380K6FKEA Vishay Dale 80.6kȍResistor, Size 0603, 1% 1
R5 CRCW060340K2FKEA Vishay Dale 40.2kȍResistor, Size 0603, 1% 1
R6 CRCW060320K0FKEA Vishay Dale 20kȍResistor, Size 0603, 1% 1
R7 CRCW060311K5FKEA Vishay Dale 11.5kȍResistor, Size 0603, 1% 1
R8 CRCW06038K06FKEA Vishay Dale 8.06kȍResistor, Size 0603, 1% 1
R9 CRCW06034K75FKEA Vishay Dale 4.75kȍResistor, Size 0603, 1% 1
R10 CRCW06033K24FKEA Vishay Dale 3.24kȍResistor, Size 0603, 1% 1
R11 CRCW06031K91FKEA Vishay Dale 1.91kȍResistor, Size 0603, 1% 1
R12 (OPEN) CRCW0603715R0FKEA Vishay Dale 715ȍResistor, Size 0603, 1%
R13 (OPEN) CRCW0603348R0FKEA Vishay Dale 348ȍ5HVLVWRU6L]H
R14, R15, R19 CRCW06030000FKEA Vishay Dale 0ȍ5HVLVWRU6L]H 5% 3
R16 CRCW08052R0FKEA Vishay Dale ȍ5HVLVWRU6L]H 1
R17 CRCW06031K65FKEA Vishay Dale 1.65Nȍ5HVLVWRU6L]H 1
R18 CRCW060349K9FKEA Vishay/Dale 49.9K,1%,1/10W,0603 1
R20 (OPEN) No Load
R21 CRCW060349R9FKEA Vishay Dale 49.9ȍResistor, Size 0603, 1% 1
R22 CRCW0603100KFKEA Vishay Dale 100kȍResistor, Size 0603, 1% 1
U1 MIC2101YML
MIC2102YML Micrel. Inc.(15)38V Synchronous Buck DC/DC Controller 1
Notes:
14. Vishay: www.vishay.com.
15. Micrel, Inc.: www.micrel.com.
Micrel, Inc. MIC2101/02
November 13, 2013 34 Revision 2.0
PCB Layout
Figure 10. MIC2101/02 Evaluation Board Top Layer
Figure 11. MIC2101/02 Evaluation Board Mid-Layer 1 (Ground Plane)
Micrel, Inc. MIC2101/02
November 13, 2013 35 Revision 2.0
PCB Layout (Continued)
Figure 12. MIC2101/02 Evaluation Board Mid-Layer 2
Figure 13. MIC2101/02 Evaluation Board Bottom Layer
Micrel, Inc. MIC2101/02
November 13, 2013 36 Revision 2.0
Package Information
16-Pin 3mm u
u
3mm QFN (ML)
N
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TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
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