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TOP264-271

TOPSwitch-JX Family

www.power.com May 2015

Integrated Off-Line Switcher with EcoSmart™ Technology

for Highly Efcient Power Supplies

This Product is Covered by Patents and/or Pending Patent Applications.

Product Highlights

EcoSmart – Energy Efﬁcient

• Ideal for applications from 10 W to 245 W

• Energy efﬁcient over entire load range

• No-load consumption below 100 mW at 265 VAC

• Up to 750 mW standby output power for 1 W input at 230 VAC

High Design Flexibility for Low System Cost

• Multi-mode PWM control maximizes efﬁciency at all loads

• 132 kHz operation reduces transformer and power supply size

• 66 kHz option for highest efﬁciency requirements

• Accurate programmable current limit

• Optimized line feed-forward for line ripple rejection

• Frequency jittering reduces EMI ﬁlter cost

• Fully integrated soft-start for minimum start-up stress

• 725 V rated MOSFET

• Simpliﬁes meeting design derating requirements

Extensive Protection Features

• Auto-restart limits power delivery to <3% during overload faults

• Output short-circuit protection (SCP)

• Output over-current protection (OCP)

• Output overload protection (OPP)

• Output overvoltage protection (OVP)

• User programmable for hysteretic/latching shutdown

• Simple fast AC reset

• Primary or secondary sensed

• Line undervoltage (UV) detection prevents turn-off glitches

• Line overvoltage (OV) shutdown extends line surge withstand

• Accurate thermal shutdown with large hysteresis (OTP)

Advanced Package Options

• eDIP™-12 package:

• 43 W / 117 W universal input power output capability with

PCB / metal heat sink

• Low proﬁle horizontal orientation for ultra-slim designs

• Heat transfer to both PCB and heat sink

• Optional external heat sink provides thermal impedance

equivalent to a TO-220

• eSIP™-7C package:

• 177 W universal input output power capability

• Vertical orientation for minimum PCB footprint

• Simple heat sink mounting using clip provides thermal

impedance equivalent to a TO-220

• eSOP™-12 package:

• 66 W universal input output power capability

• Low proﬁle surface mounted for ultra-slim designs

• Heat transfer to PCB via exposed pad and SOURCE pins

• Supports wave or reﬂow soldering

• Extended creepage to DRAIN pin

• Heat sink is connected to SOURCE for low EMI

Figure 1. Typical Flyback Application.

Figure 2. Package Options.

Description

TOPSwitch™-JX cost effectively incorporates a 725 V power

MOSFET, high-voltage switched current source, multi-mode

PWM control, oscillator, thermal shutdown circuit, fault

protection and other control circuitry onto a monolithic device.

Typical Applications

• Notebook or laptop adapter

• Generic adapter

• Printer

• LCD monitor

• Set-top box

• PC or LCD TV standby

• Audio ampliﬁer

Output Power Ratings

See next page.

eSOP-12B (K Package) eDIP-12B (V Package)

eSIP-7C (E Package)

Rev. F 05/15

TOP264-271

www.power.com

Table 1. Output Power Table.

Notes:

1. See Key Application Considerations section for more details.

2. Maximum continuous power in a typical non-ventilated enclosed adapter measured at +50 °C ambient temperature.

3. Maximum continuous power in an open frame design at +50 °C ambient temperature.

4. 230 VAC or 110/115 VAC with doubler.

5. Packages: E: eSIP-7C, V: eDIP-12, K: eSOP-12. See Part Ordering Information section.

Output Power Table

PCB Copper Area1Metal Heat Sink1

Product5

230 VAC ±15%485-265 VAC

Product5

230 VAC ±15% 485-265 VAC

Adapter2Open

Frame3Adapter2Open

Frame3

TOP264VG 21 W 34 W 12 W 22.5 W TOP264EG/VG 30 W 62 W 20 W 43 W

TOP264KG 30 W 49 W 16 W 30 W

TOP265VG 22.5 W 36 W 15 W 25 W TOP265EG/VG 40 W 81 W 26 W 57 W

TOP265KG 33 W 53 W 20 W 34 W

TOP266VG 24 W 39 W 17 W 28.5 W TOP266EG/VG 60 W 119 W 40 W 86 W

TOP266KG 36 W 58 W 23 W 39 W

TOP267VG 27.5 W 44 W 19 W 32 W TOP267EG/VG 85 W 137 W 55 W 103 W

TOP267KG 40 W 65 W 26 W 45 W

TOP268VG 30 W 48 W 21.5 W 36 W TOP268EG/VG 105 W 148 W 70 W 112 W

TOP268KG 46 W 73 W 30 W 50 W

TOP269VG 32 W 51 W 22.5 W 37.5 W TOP269EG/VG 128 W 162 W 80 W 120 W

TOP269KG 50 W 81 W 33 W 55 W

TOP270VG 34 W 55 W 24.5 W 41 W TOP270EG/VG 147 W 190 W 93 W 140 W

TOP270KG 56 W 91 W 36 W 60 W

TOP271VG 36 W 59 W 26 W 43 W TOP271EG/VG 177 W 244 W 118 W 177 W

TOP271KG 63 W 102 W 40 W 66 W

Rev. F 05/15

TOP264-271

www.power.com

Figure 3. Functional Block Diagram.

PI-4511-012810

SHUTDOWN/

AUTO-RESTART

CLOCK

CONTROLLED

TURN-ON

GATE DRIVER

CURRENT LIMIT

COMPARATOR

INTERNAL UV

COMPARATOR

INTERNAL

SUPPLY

5.8 V

4.8 V

SOURCE (S)

DMAX

STOP SOFT

START

CONTROL (C)

VOLTAGE

MONITOR (V)

FREQUENCY (F)

5.8 V

IFB

1 V

LEADING

EDGE

BLANKING

÷ 16

HYSTERETIC

THERMAL

SHUTDOWN

SHUNT REGULATOR/

ERROR AMPLIFIER +

DRAIN (D)

ON/OFF

DCMAX

66k/132k

OV/

OVPV

VI (LIMIT)

CURRENT

LIMIT

ADJUST

VBG + VT

LINE

SENSE

SOFT START

OFF

F REDUCTION

STOP LOGIC

EXTERNAL CURRENT

LIMIT (X)

OSCILLATOR

WITH JITTER

PWM

KPS(UPPER)

KPS(LOWER)

SOFT START

IFB

IPS(UPPER)

IPS(LOWER)

KPS(UPPER)

KPS(LOWER)

Pin Functional Description

DRAIN (D) Pin:

High-voltage power MOSFET DRAIN pin. The internal start-up

bias current is drawn from this pin through a switched high-

voltage current source. Internal current limit sense point for

drain current.

CONTROL (C) Pin:

Error ampliﬁer and feedback current input pin for duty cycle

control. Internal shunt regulator connection to provide internal

bias current during normal operation. It is also used as the

connection point for the supply bypass and auto-restart/

compensation capacitor.

EXTERNAL CURRENT LIMIT (X) Pin:

Input pin for external current limit adjustment remote-ON/OFF

and device reset. A connection to SOURCE pin disables all

functions on this pin. This pin should not be left ﬂoating.

VOLTAGE MONITOR (V) Pin:

Input for OV, UV, line feed-forward with DCMAX reduction, output

overvoltage protection (OVP), remote-ON/OFF. A connection to

the SOURCE pin disables all functions on this pin. This pin should

not be left ﬂoating.

FREQUENCY (F) Pin:

Input pin for selecting switching frequency 132 kHz if connected

to SOURCE pin and 66 kHz if connected to CONTROL pin. This

pin should not be left ﬂoating.

SOURCE (S) Pin:

Output MOSFET source connection for high-voltage power return.

Primary-side control circuit common and reference point.

NO CONNECTION (NC) Pin:

Internally not connected, ﬂoating potential pin.

Figure 4. Pin Conﬁguration (Top View).

12 S

11 S

10 S

9 S

8 S

7 S

V 1

X 2

C 3

F 4

D 6

1 V

2 X

3 C

4 F

6 D

S 12

S 11

S 10

S 9

S 8

S 7

PI-5568-061011

Exposed Pad

(Hidden)

Internally

Connected to

SOURCE Pin

E Package

(eSIP-7C)

V Package

(eDIP-12B)

K Package

(eSOP-12B)

Exposed Pad Internally

Connected to SOURCE Pin

Exposed Pad (On Bottom)

Internally Connected to

SOURCE Pin

Rev. F 05/15

TOP264-271

www.power.com

PI-5665-110609

Duty Cycle (%) Drain Peak Current

To Current Limit Ratio (%)

Frequency (kHz)

CONTROL

Current

CONTROL

Current

CONTROL

Current

ICOFF

IC03

IC02

IC01

ICD1

100

132

Slope = PWM Gain

Auto-Restart

Variable

Frequency

Mode

Low

Frequency

Mode

Multi-Cycle

Modulation

Jitter

Full Frequency Mode

TOP264-271 Functional Description

Like TOPSwitch-HX, TOP264-271 is an integrated switched

mode power supply chip that converts a current at the control

input to a duty cycle at the open drain output of a high-voltage

power MOSFET. During normal operation the duty cycle of the

power MOSFET decreases linearly with increasing CONTROL

pin current as shown in Figure 6.

In addition to the three terminal TOPSwitch features, such as

the high-voltage start-up, the cycle-by-cycle current limiting,

loop compensation circuitry, auto-restart and thermal shut-

down, the TOP264-271 incorporates many additional functions

that reduce system cost, increase power supply performance

and design ﬂexibility. A patented high-voltage CMOS technology

allows both the high-voltage power MOSFET and all the low

voltage control circuitry to be cost effectively integrated onto a

single monolithic chip.

Three terminals, FREQUENCY, VOLTAGE-MONITOR, and

EXTERNAL CURRENT LIMIT have been used to implement

some of the new functions. These terminals can be connected

to the SOURCE pin to operate the TOP264-271 in a TOPSwitch-

like three terminal mode. However, even in this three terminal

mode, the TOP264-271 offers many transparent features that do

not require any external components:

1. A fully integrated 17 ms soft-start signiﬁcantly reduces or

eliminates output overshoot in most applications by sweeping

both current limit and frequency from low to high to limit the

peak currents and voltages during start-up.

2. A maximum duty cycle (DCMAX) of 78% allows smaller input

storage capacitor, lower input voltage requirement and/or

higher power capability.

3. Multi-mode operation optimizes and improves the power

supply efﬁciency over the entire load range while maintaining

good cross regulation in multi-output supplies.

4. Switching frequency of 132 kHz reduces the transformer size

with no noticeable impact on EMI.

5. Frequency jittering reduces EMI in the full frequency mode at

high-load condition.

6. Hysteretic over-temperature shutdown ensures thermal fault

protection.

7. Packages with omitted pins and lead forming provide large

drain creepage distance.

8. Reduction of the auto-restart duty cycle and frequency to

improve the protection of the power supply and load during

open-loop fault, short-circuit, or loss of regulation.

9. Tighter tolerances on I2f power coefﬁcient, current limit

reduction, PWM gain and thermal shutdown threshold.

The VOLTAGE-MONITOR (V) pin is usually used for line sensing

by connecting a 4 MW resistor from this pin to the rectiﬁed DC

high-voltage bus to implement line overvoltage (OV), under-

voltage (UV) and dual-slope line feed-forward with DCMAX

reduction. In this mode, the value of the resistor determines the

OV/UV thresholds and the DCMAX is reduced linearly with a dual

slope to improve line ripple rejection. In addition, it also

provides another threshold to implement the latched and

Figure 6. Control Pin Characteristics (Multi-Mode Operation).

PI-5579-111210

Input

Voltage

CONTROL

RIL

RLS

12 kΩ

4 MΩ

VUV = IUV × RLS + VV (IV = IUV)

VOV = IOV × RLS + VV (IV = IOV)

For RLS = 4 MΩ

DCMAX@100 VDC = 76%

DCMAX@375 VDC = 41%

For RIL = 12 kΩ

ILIMIT = 61%

See Figure 37 for

other resistor values

(RIL) to select different

ILIMIT values.

VUV = 102.8 VDC

VOV = 451 VDC

Figure 5. Package Line-Sense and Externally Set Current Limit.

Rev. F 05/15

TOP264-271

www.power.com

hysteretic output overvoltage protection (OVP). The pin can

also be used as a remote-ON/OFF using the IUV threshold.

The EXTERNAL CURRENT LIMIT (X) pin can be used to reduce

the current limit externally to a value close to the operating peak

current, by connecting the pin to SOURCE through a resistor.

This pin can also be used as a remote-ON/OFF input.

The FREQUENCY (F) pin sets the switching frequency in the full

frequency PWM mode to the default value of 132 kHz when

connected to SOURCE pin. A half frequency option of 66 kHz

can be chosen by connecting this pin to the CONTROL pin

instead. Leaving this pin open is not recommended.

CONTROL (C) Pin Operation

The CONTROL pin is a low impedance node that is capable of

receiving a combined supply and feedback current. During

normal operation, a shunt regulator is used to separate the

feedback signal from the supply current. CONTROL pin voltage

VC is the supply voltage for the control circuitry including the

MOSFET gate driver. An external bypass capacitor closely

connected between the CONTROL and SOURCE pins is

required to supply the instantaneous gate drive current. The

total amount of capacitance connected to this pin also sets the

auto-restart timing as well as control loop compensation.

When rectiﬁed DC high-voltage is applied to the DRAIN pin

during start-up, the MOSFET is initially off, and the CONTROL

pin capacitor is charged through a switched high-voltage

current source connected internally between the DRAIN and

CONTROL pins. When the CONTROL pin voltage VC reaches

approximately 5.8 V, the control circuitry is activated and the

soft-start begins. The soft-start circuit gradually increases the

drain peak current and switching frequency from a low starting

value to the maximum drain peak current at the full frequency

over approximately 17 ms. If no external feedback/supply

current is fed into the CONTROL pin by the end of the soft-start,

the high-voltage current source is turned off and the CONTROL

pin will start discharging in response to the supply current

drawn by the control circuitry. If the power supply is designed

properly, and no fault condition such as open-loop or shorted

output exists, the feedback loop will close, providing external

CONTROL pin current, before the CONTROL pin voltage has

had a chance to discharge to the lower threshold voltage of

approximately 4.8 V (internal supply undervoltage lockout

threshold). When the externally fed current charges the CONTROL

pin to the shunt regulator voltage of 5.8 V, current in excess of

the consumption of the chip is shunted to SOURCE through an

NMOS current mirror as shown in Figure 3. The output current

of that NMOS current mirror controls the duty cycle of the

power MOSFET to provide closed loop regulation. The shunt

regulator has a ﬁnite low output impedance ZC that sets the gain

of the error ampliﬁer when used in a primary feedback

conﬁguration. The dynamic impedance ZC of the CONTROL pin

together with the external CONTROL pin capacitance sets the

dominant pole for the control loop.

When a fault condition such as an open-loop or shorted output

prevents the ﬂow of an external current into the CONTROL pin,

the capacitor on the CONTROL pin discharges towards 4.8 V.

At 4.8 V, auto-restart is activated, which turns the output

MOSFET off and puts the control circuitry in a low current

standby mode. The high-voltage current source turns on and

charges the external capacitance again. A hysteretic internal

supply undervoltage comparator keeps VC within a window of

typically 4.8 V to 5.8 V by turning the high-voltage current

source on and off as shown in Figure 8. The auto-restart circuit

has a divide-by-sixteen counter, which prevents the output

MOSFET from turning on again until sixteen discharge/charge

cycles have elapsed. This is accomplished by enabling the

output MOSFET only when the divide-by-sixteen counter

reaches the full count (S15). The counter effectively limits

TOP264-271 power dissipation by reducing the auto-restart

duty cycle to typically 2%. Auto-restart mode continues until

output voltage regulation is again achieved through closure of

the feedback loop.

Oscillator and Switching Frequency

The internal oscillator linearly charges and discharges an

internal capacitance between two voltage levels to create a

triangular waveform for the timing of the pulse width modulator.

This oscillator sets the pulse width modulator/current limit latch

at the beginning of each cycle.

The nominal full switching frequency of 132 kHz was chosen to

minimize transformer size while keeping the fundamental EMI

frequency below 150 kHz. The FREQUENCY pin, when shorted

to the CONTROL pin, lowers the full switching frequency to

66 kHz (half frequency), which may be preferable in some cases

such as noise sensitive video applications or a high efﬁciency

standby mode. Otherwise, the FREQUENCY pin should be

connected to the SOURCE pin for the default 132 kHz.

To further reduce the EMI level, the switching frequency in the

full frequency PWM mode is jittered (frequency modulated) by

approximately ±2.5 kHz for 66 kHz operation or ±5 kHz for

132 kHz operation at a 250 Hz (typical) rate as shown in Figure 7.

The jitter is turned off gradually as the system is entering the

variable frequency mode with a ﬁxed peak drain current.

Pulse Width Modulator

The pulse width modulator implements multi-mode control by

driving the output MOSFET with a duty cycle inversely

proportional to the current into the CONTROL pin that is in

excess of the internal supply current of the chip (see Figure 6).

The feedback error signal, in the form of the excess current, is

ﬁltered by an RC network with a typical corner frequency of

7 kHz to reduce the effect of switching noise in the chip supply

current generated by the MOSFET gate driver.

To optimize power supply efﬁciency, four different control

modes are implemented. At maximum load, the modulator

operates in full frequency PWM mode; as load decreases, the

modulator automatically transitions, ﬁrst to variable frequency

PWM mode, then to low frequency PWM mode. At light load,

the control operation switches from PWM control to multi-cycle-

modulation control, and the modulator operates in multi-cycle-

modulation mode. Although different modes operate differently

to make transitions between modes smooth, the simple

relationship between duty cycle and excess CONTROL pin

current shown in Figure 6 is maintained through all three PWM

Rev. F 05/15

TOP264-271

www.power.com

Figure 7. Switching Frequency Jitter (Idealized VDRAIN Waveforms).

PI-4530-041107

fOSC -

4 ms

Time

Switching

Frequency

VDRAIN

fOSC +

modes. Please see the following sections for the details of the

operation of each mode and the transitions between modes.

Full Frequency PWM mode: The PWM modulator enters full

frequency PWM mode when the CONTROL pin current (IC)

reaches IB. In this mode, the average switching frequency is

kept constant at fOSC (pin selectable 132 kHz or 66 kHz). Duty

cycle is reduced from DCMAX through the reduction of the on-time

when IC is increased beyond IB. This operation is identical to the

PWM control of all other TOPSwitch families. TOP264-271 only

operates in this mode if the cycle-by-cycle peak drain current

stays above kPS(UPPER) × ILIMIT(set), where kPS(UPPER) is 55% (typical)

and ILIMIT(set) is the current limit externally set via the EXTERNAL

CURRENT LIMIT (X) pin.

Variable Frequency PWM mode: When peak drain current is

lowered to kPS(UPPER) × ILIMIT(set) as a result of power supply load

reduction, the PWM modulator initiates the transition to variable

frequency PWM mode, and gradually turns off frequency jitter.

In this mode, peak drain current is held constant at kPS(UPPER) ×

ILIMIT(set) while switching frequency drops from the initial full

frequency of fOSC (132 kHz or 66 kHz) towards the minimum

frequency of fMCM(MIN) (30 kHz typical). Duty cycle reduction is

accomplished by extending the off-time.

Low Frequency PWM mode: When switching frequency

reaches fMCM(MIN) (30 kHz typical), the PWM modulator starts to

transition to low frequency mode. In this mode, switching

frequency is held constant at fMCM(MIN) and duty cycle is reduced,

similar to the full frequency PWM mode, through the reduction

of the on-time. Peak drain current decreases from the initial

value of kPS(UPPER) × ILIMIT(set) towards the minimum value of

kPS(LOWER) × ILIMIT(set), where kPS(LOWER) is 25% (typical) and ILIMIT(set)

is the current limit externally set via the X pin.

Multi-Cycle-Modulation mode: When peak drain current is

lowered to kPS(LOWER) × ILIMIT(set), the modulator transitions to

multi-cycle-modulation mode. In this mode, at each turn-on,

the modulator enables output switching for a period of TMCM(MIN)

at the switching frequency of fMCM(MIN) (4 or 5 consecutive pulses

at 30 kHz) with the peak drain current of kPS(LOWER) × ILIMIT(set),

and stays off until the CONTROL pin current falls below IC(OFF).

This mode of operation not only keeps peak drain current low

but also minimizes harmonic frequencies between 6 kHz and

30 kHz. By avoiding transformer resonant frequency this way,

all potential transformer audible noises are greatly suppressed.

Maximum Duty Cycle

The maximum duty cycle, DCMAX, is set at a default maximum

value of 78% (typical). However, by connecting the VOLTAGE-

MONITOR to the rectiﬁed DC high-voltage bus through a resistor

with appropriate value (4 MW typical), the maximum duty cycle

can be made to decrease from 78% to 40% (typical) when input

line voltage increases from 88 V to 380 V, with dual gain slopes.

Error Ampliﬁer

The shunt regulator can also perform the function of an error

ampliﬁer in primary-side feedback applications. The shunt

regulator voltage is accurately derived from a temperature-

compensated bandgap reference. The CONTROL pin dynamic

impedance ZC sets the gain of the error ampliﬁer. The CONTROL

pin clamps external circuit signals to the VC voltage level. The

CONTROL pin current in excess of the supply current is

separated by the shunt regulator and becomes the feedback

current IFB for the pulse width modulator.

On-Chip Current Limit with External Programmability

The cycle-by-cycle peak drain current limit circuit uses the

output MOSFET ON-resistance as a sense resistor. A current

limit comparator compares the output MOSFET ON-state drain

to source voltage VDS(ON) with a threshold voltage. High drain

current causes VDS(ON) to exceed the threshold voltage and turns

the output MOSFET off until the start of the next clock cycle.

The current limit comparator threshold voltage is temperature

compensated to minimize the variation of the current limit due

to temperature related changes in RDS(ON) of the output MOSFET.

The default current limit of TOP264-271 is preset internally.

However, with a resistor connected between EXTERNAL

CURRENT LIMIT (X) pin and SOURCE pin, current limit can be

programmed externally to a lower level between 30% and 100%

of the default current limit. By setting current limit low, a larger

TOP264-271 than necessary for the power required can be used

to take advantage of the lower RDS(ON) for higher efﬁciency/

smaller heat sinking requirements. With a second resistor

connected between the EXTERNAL CURRENT LIMIT (X) pin

and the rectiﬁed DC high-voltage bus, the current limit is

reduced with increasing line voltage, allowing a true power

limiting operation against line variation to be implemented. When

using an RCD clamp, this power limiting technique reduces

maximum clamp voltage at high-line. This allows for higher

reﬂected voltage designs as well as reducing clamp dissipation.

The leading edge blanking circuit inhibits the current limit

comparator for a short time after the output MOSFET is turned

on. The leading edge blanking time has been set so that, if a

power supply is designed properly, current spikes caused by

primary-side capacitances and secondary-side rectiﬁer reverse

recovery time should not cause premature termination of the

switching pulse. The current limit is lower for a short period

after the leading edge blanking time. This is due to dynamic

characteristics of the MOSFET. During start-up and fault

conditions the controller prevents excessive drain currents by

reducing the switching frequency.

Line Undervoltage Detection (UV)

At power-up, UV keeps TOP264-271 off until the input line

voltage reaches the undervoltage threshold. At power-down,

Rev. F 05/15

TOP264-271

www.power.com

UV prevents auto-restart attempts after the output goes out of

regulation. This eliminates power-down glitches caused by

slow discharge of the large input storage capacitor present in

applications such as standby supplies. A single resistor

connected from the VOLTAGE-MONITOR pin to the rectiﬁed DC

high-voltage bus sets UV threshold during power-up. Once the

power supply is successfully turned on, the UV threshold is

lowered to 44% of the initial UV threshold to allow extended

input voltage operating range (UV low threshold). If the UV low

threshold is reached during operation without the power supply

losing regulation, the device will turn off and stay off until UV

(high threshold) has been reached again. If the power supply

loses regulation before reaching the UV low threshold, the

device will enter auto-restart. At the end of each auto-restart

cycle (S15), the UV comparator is enabled. If the UV high

threshold is not exceeded, the MOSFET will be disabled during

the next cycle (see Figure 8). The UV feature can be disabled

independent of the OV feature.

Line Overvoltage Shutdown (OV)

The same resistor used for UV also sets an overvoltage

threshold, which, once exceeded, will force TOP264-271 to

stop switching instantaneously (after completion of the current

switching cycle). If this condition lasts for at least 100 ms, the

TOP264-271 output will be forced into off state. When the line

voltage is back to normal with a small amount of hysteresis

provided on the OV threshold to prevent noise triggering, the

state machine sets to S13 and forces TOP264-271 to go

through the entire auto-restart sequence before attempting to

switch again. The ratio of OV and UV thresholds is preset at

4.5, as can be seen in Figure 9. When the MOSFET is off, the

rectiﬁed DC high-voltage surge capability is increased to the

voltage rating of the MOSFET (725 V), due to the absence of the

reﬂected voltage and leakage spikes on the drain. The OV

feature can be disabled independent of the UV feature.

In order to reduce the no-load input power of TOP264-271

designs, the V pin operates at very low currents. This requires

careful layout considerations when designing the PCB to avoid

noise coupling. Traces and components connected to the V pin

should not be adjacent to any traces carrying switching currents.

These include the drain, clamp network, bias winding return or

power traces from other converters. If the line sensing features

are used, then the sense resistors must be placed within 10 mm

of the V pin to minimize the V pin node area. The DC bus

should then be routed to the line-sense resistors. Note that

external capacitance must not be connected to the V pin as this

may cause misoperation of the V pin related functions.

Hysteretic or Latching Output Overvoltage Protection (OVP)

The detection of the hysteretic or latching output overvoltage

protection (OVP) is through the trigger of the line overvoltage

threshold. The V pin voltage will drop by 0.5 V, and the

controller measures the external attached impedance immediately

after this voltage drops. If IV exceeds IOV(LS) (336 mA typical)

longer than 100 ms, TOP264-271 will latch into a permanent

off-state for the latching OVP. It only can be reset if IX exceeds

IX(TH) = -27 mA (typ) or VC goes below the power-up reset

threshold (VC(RESET)) and then back to normal. If IV does not

exceed IOV(LS) or exceeds no longer than 100 ms, TOP264-271

will initiate the line overvoltage and the hysteretic OVP. Their

behavior will be identical to the line overvoltage shutdown (OV)

that has been described in detail in the previous section.

During a fault condition resulting from loss of feedback, output

voltage will rapidly rise above the nominal voltage. The increase

in output voltage will also result in an increase in the voltage at

the output of the bias winding. A voltage at the output of the

bias winding that exceeds of the sum of the voltage rating of the

Zener diode connected from the bias winding output to the V

pin and V pin voltage, will cause a current in excess of IV to be

injected into the V pin, which will trigger the OVP feature.

PI-4531-121206

S13 S12 S0 S15 S13 S12 S0 S15S14 S13 S15

S14 S14 5.8 V

4.8 V

S15

0 V

VLINE

VDRAIN

VOUT

Note: S0 through S15 are the output states of the auto-restart counter

1234

0 V

S0 S15

VUV

S12

Figure 8. Typical Waveforms for (1) Power-Up (2) Normal Operation (3) Auto-Restart (4) Power-Down.

Rev. F 05/15

TOP264-271

www.power.com

If the power supply is operating under heavy load or low input

line conditions when an open-loop occurs, the output voltage

may not rise signiﬁcantly. Under these conditions, a latching

shutdown will not occur until load or line conditions change.

Nevertheless, the operation provides the desired protection by

preventing signiﬁcant rise in the output voltage when the line or

load conditions do change. Primary-side OVP protection with

the TOP264-271 in a typical application will prevent a nominal

12 V output from rising above approximately 20 V under open-

loop conditions. If greater accuracy is required, a secondary

sensed OVP circuit is recommended.

Line Feed-Forward with DCMAX Reduction

The same resistor used for UV and OV also implements line voltage

feed-forward, which minimizes output line ripple and reduces

power supply output sensitivity to line transients. Note that for the

same CONTROL pin current, higher line voltage results in smaller

operating duty cycle. As an added feature, the maximum duty

cycle DCMAX is also reduced from 78% (typical) at a voltage slightly

lower than the UV threshold to 36% (typical) at the OV threshold.

DCMAX of 36% at high-line was chosen to ensure that the power

capability of the TOP264-271 is not restricted by this feature under

normal operation. TOP264-271 provides a better ﬁt to the ideal

feed-forward by using two reduction slopes: -1% per mA for all bus

voltage less than 195 V (typical for 4 MW line impedance) and

-0.25% per mA for all bus voltage more than 195 V.

Remote-ON/OFF

TOP264-271 can be turned on or off by controlling the current into

the VOLTAGE-MONITOR pin or out from the EXTERNAL CURRENT

LIMIT pin. In addition, the VOLTAGE-MONITOR pin has a 1 V

threshold comparator connected at its input. This voltage

threshold can also be used to perform remote-ON/OFF control.

When a signal is received at the VOLTAGE-MONITOR pin or the

EXTERNAL CURRENT LIMIT pin to disable the output through

any of the pin functions such as OV, UV and remote-ON/OFF,

TOP264-271 always completes its current switching cycle before

the output is forced off.

As seen above, the remote-ON/OFF feature can also be used

as a standby or power switch to turn off the TOP264-271 and

keep it in a very low power consumption state for indeﬁnitely long

periods. If the TOP264-271 is held in remote-off state for long

enough time to allow the CONTROL pin to discharge to the

internal supply undervoltage threshold of 4.8 V (approximately

32 ms for a 47 mF CONTROL pin capacitance), the CONTROL

pin goes into the hysteretic mode of regulation. In this mode,

the CONTROL pin goes through alternate charge and discharge

cycles between 4.8 V and 5.8 V (see CONTROL pin operation

section above) and runs entirely off the high-voltage DC input,

but with very low power consumption (<100 mW typical at

230 VAC with X pin open). When the TOP264-271 is remotely

Voltage Monitor and External Current Limit Pin Table*

Figure Number 13 14 15 16 17 18 19 20 21 22 23 24

Three Terminal Operation 3

Line Undervoltage (UV) 3 3 3 3 3

Line Overvoltage (OV) 3 3 3 3 3

Line Feed-Forward (DCMAX)3 3 3 3

Output Overvoltage Protection (OVP) 3

Overload Power Limiting (OPP) 3

External Current Limit 3 3 3 3 3 3

Remote-ON/OFF 333

Device Reset 333

Fast AC Reset 3

AC Brown-Out 3

*This table is only a partial list of many VOLTAGE MONITOR and EXTERNAL CURRENT LIMIT Pin Conﬁgurations that are possible.

Table 2. VOLTAGE MONITOR (V) Pin and EXTERNAL CURRENT LIMIT (X) Pin Conﬁguration Options.

Rev. F 05/15

TOP264-271

www.power.com

Figure 9. VOLTAGE MONITOR and EXTERNAL CURRENT LIMIT Pin Characteristics.

-250 -200 -150 -100 -50 0 25 50 75 100 125 336

PI-5528-060409

Output

MOSFET

Switching

(Enabled)

(Disabled)

(Non-Latching) (Latching)

ILIMIT (Default)

DCMAX (78%)

Current

Limit

V PinX Pin

Maximum

Duty Cycle

VBG

IUV

IREM(N) IOV IOV(LS)

Pin Voltage

Note: This figure provides idealized functional characteristics with typical performance values. Please refer to the parametric

table and typical performance characteristics sections of the data sheet for measured data. For a detailed description of

each functional pin operation refer to the Functional Description section of the data sheet.

X and V Pins Current (µA)

Disabled when supply

output goes out of

regulation

turned on after entering this mode, it will initiate a normal

start-up sequence with soft-start the next time the CONTROL

pin reaches 5.8 V. In the worst-case, the delay from remote-on

to start-up can be equal to the full discharge/charge cycle time

of the CONTROL pin, which is approximately 125 ms for a

47 mF CONTROL pin capacitor. This reduced consumption

remote-off mode can eliminate expensive and unreliable in-line

mechanical switches. It also allows for microprocessor

controlled turn-on and turn-off sequences that may be required

in certain applications such as inkjet and laser printers.

Soft-Start

The 17 ms soft-start sweeps the peak drain current and switching

frequency linearly from minimum to maximum value by operating

through the low frequency PWM mode and the variable

frequency mode before entering the full frequency mode. In

addition to start-up, soft-start is also activated at each restart

attempt during auto-restart and when restarting after being in

hysteretic regulation of CONTROL pin voltage (VC), due to

remote-off or thermal shutdown conditions. This effectively

minimizes current and voltage stresses on the output MOSFET,

Rev. F 05/15

TOP264-271

www.power.com

the clamp circuit and the output rectiﬁer during start-up. This

feature also helps minimize output overshoot and prevents

saturation of the transformer during start-up.

Shutdown/Auto-Restart (for OCP, SCP, OPP)

To minimize TOP264-271 power dissipation under fault

conditions such as over current (OC), short-circuit (SC) or over

power (OP), the shutdown/auto-restart circuit turns the power

supply on and off at an auto-restart duty cycle of typically 2% if

an out of regulation condition persists. Loss of regulation

interrupts the external current into the CONTROL pin. VC

regulation changes from shunt mode to the hysteretic auto-

restart mode as described in CONTROL pin operation section.

When the fault condition is removed, the power supply output

becomes regulated, VC regulation returns to shunt mode, and

normal operation of the power supply resumes.

Hysteretic Over-Temperature Protection (OTP)

Temperature protection is provided by a precision analog circuit

that turns the output MOSFET off when the junction temperature

exceeds the thermal shutdown temperature (142 °C typical).

When the junction temperature cools to below the lower

hysteretic temperature point, normal operation resumes, thus

providing automatic recovery. A large hysteresis of 75 °C

(typical) is provided to prevent overheating of the PC board due

to a continuous fault condition. VC is regulated in hysteretic

mode, and a 4.8 V to 5.8 V (typical) triangular waveform is

present on the CONTROL pin while in thermal shutdown.

Bandgap Reference

All critical TOP264-271 internal voltages are derived from a

temperature-compensated bandgap reference. This voltage

reference is used to generate all other internal current references,

which are trimmed to accurately set the switching frequency,

MOSFET gate drive current, current limit, and the line OV/UV/

OVP thresholds. TOP264-271 has improved circuitry to

maintain all of the above critical parameters within very tight

absolute and temperature tolerances.

High-Voltage Bias Current Source

This high-voltage current source biases TOP264-271 from the

DRAIN pin and charges the CONTROL pin external capacitance

during start-up or hysteretic operation. Hysteretic operation

occurs during auto-restart, remote-off and over-temperature

shutdown. In this mode of operation, the current source is

switched on and off, with an effective duty cycle of approxi-

mately 35%. This duty cycle is determined by the ratio of

CONTROL pin charge (IC) and discharge currents (ICD1 and ICD2).

This current source is turned off during normal operation when

the output MOSFET is switching. The effect of the current

source switching will be seen on the DRAIN voltage waveform

as small disturbances and is normal.

Rev. F 05/15

TOP264-271

www.power.com

VBG + VT

1 V

VREF

200 µA

400 µA

CONTROL (C)

(Voltage Sense, ON/OFF)

(Positive Current Sense - Undervoltage,

Overvoltage, ON/OFF, Maximum Duty

Cycle Reduction, Output Over-

voltage Protection)

(Negative Current Sense - ON/OFF,

Current Limit Adjustment, OVP Latch Reset)

PI-5567-030910

VOLTAGE MONITOR (V)

EXTERNAL CURRENT LIMIT (X)

Figure 10. VOLTAGE MONITOR (V) and EXTERNAL CURRENT LIMIT (X) Pin Input Simpliﬁed Schematic.

Rev. F 05/15

TOP264-271

www.power.com

Typical Uses of FREQUENCY (F) Pin

PI-2654-071700

Input

Voltage

CONTROL

PI-2655-071700

Input

Voltage

CONTROL

Figure 11. Full Frequency Operation (132 kHz). Figure 12. Half Frequency Operation (66 kHz).

Rev. F 05/15

TOP264-271

www.power.com

Typical Uses of VOLTAGE MONITOR (V) and EXTERNAL CURRENT LIMIT (X) Pins

PI-4717-120307

Input

Voltage

CONTROL

4 MΩRLS

VUV = IUV × RLS + VV (IV = IUV)

VOV = IOV × RLS + VV (IV = IOV)

For RLS = 4 MΩ

VUV = 102.8 VDC

VOV = 451 VDC

DCMAX@100 VDC = 76%

DCMAX@375 VDC = 41%

PI-4719-120307

Input

Voltage

Sense Output Voltage

D V

VUV = IUV × RLS + VV (IV = IUV)

VOV = IOV × RLS + VV (IV = IOV)

For RLS = 4 MΩ

VUV = 102.8 VDC

VOV = 451 VDC

DCMAX @ 100 VDC = 76%

DCMAX @ 375 VDC = 41%

CONTROL

RLS 4 MΩ

ROVP >3kΩ

VROVP

ROVP

Figure 14. Line-Sensing for Undervoltage, Overvoltage and Line Feed-Forward.

Figure 15. Line-Sensing for Undervoltage, Overvoltage, Line Feed-Forward and

Hysteretic Output Overvoltage Protection.

Figure 13. Three Terminal Operation (VOLTAGE MONITOR and EXTERNAL

CURRENT LIMIT Features Disabled. FREQUENCY Pin Tied to

SOURCE or CONTROL Pin.)

X F

Input

Voltage

D C

S D

CONTROL

V X C SF D

PI-6119-061011

E Package

(eSIP-7C)

V Package (eDIP-12)

1 V

2 X

3 C

4 F

6 D

S 12

S 11

S 10

S 9

S 8

S 7

K Package (eSOP-12)

S 12

S 11

S 10

S 9

S 8

S 7

1 V

2 X

3 C

4 F

6 D

PI-4720-120307

Input

Voltage

D V

VUV = RLS × IUV + VV (IV = IUV)

For Values Shown

VUV = 103.8 VDC

RLS

6.2 V

4 MΩ

40 kΩ

CONTROL

PI-4721-120307

Input

Voltage

CONTROL

4 MΩ

55 kΩ

RLS

1N4148

VOV = IOV × RLS + VV (IV = IOV)

For Values Shown

VOV = 457.2 VDC

Figure 16. Line-Sensing for Undervoltage Only (Overvoltage Disabled).

Figure 17. Line-Sensing for Overvoltage Only (Undervoltage Disabled). Maximum

Duty Cycle Reduced at Low-Line and Further Reduction with

Increasing Line Voltage.

Figure 18. External Set Current Limit.

PI-5580-111210

Input

Voltage

RIL

For RIL = 12 kΩ

ILIMIT = 61%

See Figure 37 for other

resistor values (RIL).

For RIL = 19 kΩ

ILIMIT = 37%

CONTROL

Rev. F 05/15

TOP264-271

www.power.com

Typical Uses of VOLTAGE MONITOR (V) and EXTERNAL CURRENT LIMIT (X) Pins (cont.)

PI-5465-061009

Input

Voltage

2.5 MΩRLS

6 kΩ

RIL

100% @ 100 VDC

53% @ 300 VDC

ILIMIT =

CONTROL

PI-5466-061009

Input

Voltage

ON/OFF

47 KΩ

QR can be an optocoupler

output or can be replaced by

a manual switch.

CONTROL

ON/OFF

16 kΩ

PI-5531-072309

Input

Voltage

RIL QR

12 kΩ

For RIL =

ILIMIT = 61%

19 kΩ

For RIL =

ILIMIT = 37%

QR can be an optocoupler

output or can be replaced

by a manual switch.

CONTROL

Figure 19. Current Limit Reduction with Line Voltage. Figure 20. Active-On (Fail Safe) Remote-ON/OFF, and Latch Reset.

Figure 21. Active-On Remote-ON/OFF with Externally Set Current Limit,

and Latch Reset

ON/OFF

16 kΩ

PI-5467-061009

Input

Voltage

CONTROL

RIL

RLS

4 MΩ

VUV = IUV × RLS + VV (IV = IUV)

VOV = IOV × RLS + VV (IV = IoV)

DCMAX@100 VDC = 76%

DCMAX@375 VDC = 41%

12 kΩ

For RIL =

ILIMIT = 61%

QR can be an optocoupler

output or can be replaced

by a manual switch.

PI-5565-111210

Input

Voltage

CONTROL

RIL

RLS

12 kΩ

4 MΩ

VUV = IUV x RLS + VV (IV = IUV)

VOV = IOV x RLS + VV (IV = IoV)

For RLS = 4 MΩ

DCMAX @ 100 VDC = 76%

DCMAX @ 375 VDC = 41%

For RIL = 12 kΩ

ILIMIT = 61%

See Figure 37 for

other resistor values

(RIL) to select different

ILIMIT values.

VUV = 102.8 VDC

VOV = 451 VDC

Figure 22. Active-On Remote-ON/OFF with Line-Sense and External

Current Limit, and Latch Reset.

Figure 23. Line Sensing and Externally Set Current Limit. Figure 24. Externally Set Current Limit, Fast AC Latch Reset and Brown-Out.

Input

4 MΩ1N4007

47 nF

PI-5652-110609

RIL QR

Typ. 65 VAC brownout threshold.

<3 s AC latch reset time.

Higher gain QR allows increasing R1/

decreasing C1 for lower no-load input

power.

CONTROL

39 kΩ

Input

Voltage

Rev. F 05/15

TOP264-271

www.power.com

PI-5667-030810

R15

33 Ω

R19

20 kΩ

R18

10 kΩ

R17

147 kΩ

R16

20 kΩ

R27

10 kΩ

R22

1.6 kΩ

LMV431AIMF

RS1K

BAV19WS R12

4.7 kΩ

2.2 MΩ

R13

6.8 Ω

1/8 W

R20

191 kΩ

R25

20 Ω

1/8 W

150 Ω

300 Ω

R29

300 Ω

R11

300 Ω

R28

300 Ω

U3B

PS2501-

1-H-A

U3A

PS2501-

1-H-A

200 µH

12 mH

TOP269EG

47 µF

16 V

C16

22 nF

50 V

C22

100 nF

50 V

C19

6.8 nF

50 V

C10

56 µF

35 V

BAV21WS-

7-F

V30100C

100 nF

50 V

GBU8J

600 V

C13

470 µF

25 V

C14

470 µF

25 V

C21

10 nF

50 V

C12

1 nF

100 V

RM10 FL1

FL2

C11

1 nF

250 VAC

330 nF

275 VAC

4 A

TOPSwitch-JX

19 V, 3.42 A

RTN

120 µF

400 V C9

220 nF

25 V

1000 pF

630 V

2.2 nF

1 kV

VR2

SMAJ250A

90 - 265

VAC

CONTROL

5.1 MΩ

10 MΩ

5.1 MΩ

11 kΩ

10 MΩ

R24

2.2 Ω

R14

20 Ω

C15

470 pF

50 V

MMBT3904

VR1

ZMM5244B-7

MMBT4403

Input Voltage (VAC) 90 230

Full Power Efciency (%) 86.6 89.1

Average Efciency (%) 89.5

No-load Input Power (mW) 57.7

115

88.4

89.8

59.7 86.7

R10

100 Ω

Figure 25. Schematic of High Efﬁciency 19 V, 65 W, Universal Input Flyback Supply with Low No-load.

Application Example

Low No-Load, High Efﬁciency, 65 W, Universal Input

Adapter Power Supply

The circuit shown in Figure 25 shows a 90 VAC to 265 VAC

input, 19 V, 3.42 A output power supply, designed for operation

inside a sealed adapter case type. The goals of the design were

highest full load efﬁciency, highest average efﬁciency (average of

25%, 50%, 75% and 100% load points), and very low no-load

consumption. Additional requirements included latching output

overvoltage shutdown and compliance to safety agency limited

power source (LPS) limits. Measured efﬁciency and no-load

performance is summarized in the table shown in the schematic

which easily exceed current energy efﬁciency requirements.

In order to meet these design goals the following key design

decisions were made.

PI Part Selection

• One device size larger selected than required for power

delivery to increase efﬁciency

The current limit programming feature of TOPSwitch-JX allows

the selection of a larger device than needed for power delivery.

This gives higher full load, low-line efﬁciency by reducing the

MOSFET conduction losses (IRMS

2 × RDS(ON)) but maintains the

overload power, transformer and other components size as if a

smaller device had been used.

For this design one device size larger than required for power

delivery (as recommended by the power table) was selected.

This typically gives the highest efﬁciency. Further increases in

device size often results in the same or lower efﬁciency due to

the larger switching losses associated with a larger MOSFET.

Line-Sense Resistor Values

• Increasing line-sensing resistance from 4 MW to 10.2 MW to

reduce no-load input power dissipation by 16 mW

Line-sensing is provided by resistors R3 and R4 and sets the

line undervoltage and overvoltage thresholds. The combined

value of these resistors was increased from the standard 4 MW

to 10.2 MW. This reduced the resistor dissipation, and therefore

contribution to no-load input power, from ~26 mW to ~10 mW. To

compensate the resultant change in the UV (turn-on) threshold

resistor R20 was added between the CONTROL and VOLTAGE-

MONITOR pins. This adds a DC current equal to ~16 mA into the

VOLTAGE MONITOR pin, requiring only 9 mA to be provided via

R3 and R4 to reach the VOLTAGE MONITOR pin UV (turn-on)

threshold current of 25 mA and setting the UV threshold to 95 VDC.

This technique does effectively disable the line OV feature as

the resultant OV threshold is raised from ~450 VDC to ~980 VDC.

However in this design there was no impact as the value of

input capacitance (C2) was sufﬁcient to allow the design to

withstand differential line surges greater than 2 kV without the

peak drain voltage reaching the BVDSS rating of U1.

Speciﬁc guidelines and detailed calculations for the value of

R20 may be found in the TOPSwitch-JX Application Note (AN-47).

Clamp Conﬁguration – RZCD vs RCD

• An RZCD (Zener bleed) was selected over an RCD clamp to

give higher light load efﬁciency and lower no-load consumption

The clamp network is formed by VR2, C4, R5, R6, R11, R28,

R29 and D2. It limits the peak drain voltage spike caused by

leakage inductance to below the BVDSS rating of the internal

Rev. F 05/15

TOP264-271

www.power.com

TOPSwitch-JX MOSFET. This arrangement was selected over

a standard RCD clamp to improve light load efﬁciency and no-load

input power.

In a standard RCD clamp C4 would be discharged by a parallel

resistor rather than a resistor and series Zener. In an RCD clamp

the resistor value is selected to limit the peak drain voltage

under full load and overload conditions. However under light or

no-load conditions this resistor value now causes the capacitor

voltage to discharge signiﬁcantly as both the leakage inductance

energy and switching frequency are lower. As the capacitor has

to be recharged to above the reﬂected output voltage each

switching cycle the lower capacitor voltage represents wasted

energy. It has the effect of making the clamp dissipation

appear as a signiﬁcant load just as if it were connected to the

output of the power supply.

The RZCD arrangement solves this problem by preventing the

voltage across the capacitor discharging below a minimum

value (deﬁned by the voltage rating of VR2) and therefore

minimizing clamp dissipation under light and no-load conditions.

Resistors R6 and R28 provide damping of high frequency

ringing to reduce EMI. Due to the resistance in series with VR2,

limiting the peak current, standard power Zeners vs a TVS type

may be used for lower cost (although a TVS type was selected

due to availability of a SMD version). Diode D2 was selected to

have an 800 V vs the typical 600 V rating due to its longer

reverse recovery time of 500 ns. This allows some recovery of

the clamp energy during the reverse recovery time of the diode

improving efﬁciency. Multiple resistors were used in parallel to

share dissipation as SMD components were used.

Feedback Conﬁguration

• A Darlington connection formed together with optocoupler

transistor to reduce secondary-side feedback current and

therefore no-load input power.

• Low voltage, low current voltage reference IC used on

secondary-side to reduce secondary-side feedback current

and therefore no-load input power.

• Bias winding voltage tuned to ~9 V at no-load, high-line to

reduce no-load input power.

Typically the feedback current into the CONTROL pin at

high-line is ~3 mA. This current is both sourced from the bias

winding (voltage across C10) and directly from the output. Both

of these represent a load on the output of the power supply.

To minimize the dissipation from the bias winding under no-load

conditions the number of bias winding turns and value of C10

was adjusted to give a minimum voltage across C10 of ~9 V.

This is the minimum required to keep the optocoupler biased.

To minimize the dissipation of the secondary-side feedback

circuit Q2 was added to form a Darlington connection with U3B.

This reduced the feedback current on the secondary to ~1 mA.

The increased loop gain (due to the hFE of the transistor) was

compensated by increasing the value of R16 and the addition of

R25. A standard 2.5 V TL431 voltage reference was replaced

with the 1.24 V LMV431 to reduce the supply current requirement

from 1 mA to 100 mA.

Output Rectiﬁer Choice

• Higher current rating, low VF Schottky rectiﬁer diode selected

for output rectiﬁer.

A dual 15 A, 100 V Schottky rectiﬁer diode with a VF of 0.455 V

at 5 A was selected for D5. This is a higher current rating than

required to reduce resistive and forward voltage losses to improve

both full load and average efﬁciency. The use of a 100 V Schottky

was possible due to the high transformer primary to secondary

turns ratio (VOR = 110 V) which was in turn possible due to the

high-voltage rating of the TOPSwitch-JX internal MOSFET.

Increased Output Overvoltage Shutdown Sensitivity

• Transistor Q1 and VR1 added to improve the output over-

voltage shutdown sensitivity.

During an open-loop condition the output and therefore bias

winding voltage will rise. When this exceeds the voltage of VR1

plus a VBE voltage drop Q1 turns on and current is fed into the

VOLTAGE MONITOR pin. The addition of Q1 ensures that the

current into the VOLTAGE MONITOR pin is sufﬁcient to exceed

the latching shutdown threshold even when the output is fully

loaded while the supply is operating at low-line as under this

condition the output voltage overshoot is relatively small

Output overload power limitation is provided via the current limit

programming feature of the X pin and R7, R8 and R9. Resistors

R8 and R9 reduce the device current limit as a function of

increasing line voltage to provide a roughly ﬂat overload power

characteristic, below the 100 VA limited power source (LPS)

requirement. In order to still meet this under a single fault

condition (such as open circuit of R8) the rise in the bias voltage

that occurs during an overload condition is also used to trigger

a latching shutdown.

Rev. F 05/15

TOP264-271

www.power.com

Very Low No-Load, High Efﬁciency, 30 W, Universal

Input, Open Frame, Power Supply

The circuit shown in Figure 26 below shows an 85 VAC to

265 VAC input, 12 V, 2.5 A output power supply. The goals of

the design were highest full load efﬁciency, average efﬁciency

(average of 25%, 50%, 75% and 100% load points), very low no-

load consumption. Additional requirements included latching

output overvoltage shutdown and compliance to safety agency

limited power source (LPS) limits. Actual efﬁciency and no-load

performance is summarized in the table shown in the schematic

which easily exceed current energy efﬁciency requirements.

In order to meet these design goals the following key design

decisions were made.

PI Part Selection

• Ambient of 40 °C allowed one device size smaller than

indicated by the power table

The device selected for this design was based on the 85-265 VAC,

Open Frame, PCB heat sinking column of power table (Table 1).

One device size smaller was selected (TOP266V vs TOP267V)

due to the ambient speciﬁcation of 40 °C (vs the 50°C assumed

in the power table) and the optimum PCB area and layout for

the device heat sink. The subsequent thermal and efﬁciency

data conﬁrmed this choice. The maximum device temperature

was 107 °C at full load, 40 °C, 85 VAC, 47 Hz (worst-case

conditions) and average efﬁciency exceeded 83% ENERGY

STAR and EuP Tier 2 requirements.

Transformer Core Selection

• 132 kHz switching frequency allowed the selection of smaller

core for lower cost.

The size of the magnetic core is a function of the switching

frequency. The choice of the higher switching frequency of

132 kHz allowed for the use of a smaller core size. The higher

switching frequency does not negatively impact the efﬁciency in

TOPSwitch-JX designs due its small drain to source capacitance

(COSS) as compared to that of discrete MOSFETs.

Line-Sense Resistor Values

• Increasing line-sensing resistance from 4 MW to 10.2 MW to

reduce no-load input power dissipation by 16 mW.

Line-sensing is provided by resistors R1 and R2 and sets the

line undervoltage and overvoltage thresholds. The combined

value of these resistors was increased from the standard 4 MW

to 10.2 MW. This reduces the current into the VOLTAGE

MONITOR pin, and therefore contribution to no-load input power,

from ~26 mW to ~10 mW. To compensate the resultant change in

the UV threshold resistor R12 was added between the CONTROL

and VOLTAGE-MONITOR pins. This adds a DC current equal to

~16 mA into the VOLTAGE MONITOR pin, requiring only 9 mA to

be provided via R1 and R2 to reach the VOLTAGE MONITOR

pin UV threshold current of 25 mA and setting the UV threshold

to approximately 95 VDC.

This technique does effectively disable the line OV feature as

the resultant OV threshold is raised from ~450 VDC to ~980 VDC.

However in this design there was no impact as the value of

input capacitance (C3) was sufﬁcient to allow the design to

withstand differential line surges greater than 1 kV without the

peak drain voltage reaching the BVDSS rating of U1.

Speciﬁc guidelines and detailed calculations for the value of R12

may be found in the TOPSwitch-JX Application Note.

Figure 26. Schematic of High Efﬁciency 12 V, 30 W, Universal Input Flyback Supply with Very Low No-load.

PI-5775-030810

R17

22 Ω

R23

10 kΩ

R21

86.6 kΩ

D10

LL4148

R19

470 Ω

R18

110 Ω

LMV431A

FR107

BAV19WS

VR3

ZMM5245B-7

R16

6.8 Ω

1/8 W

10 Ω

R12

191 kΩ

10 kΩ

1/2 W

U2B

LTV817D

U2A

LTV817D

14 mH

TOP266VG

C10

47 µF

25 V

C18

47 nF

50 V

47 µF

25 V

BAV21WS-

7-F

D8,9

SB560

100 nF

50 V

1N4007

C14

680 µF

25 V

C15

680 µF

25 V

3.3 µH

C16

100 µF

25 V

C12

1 nF

200 V

7,8

11,12

EF25

C11

1 nF

250 VAC

100 nF

275 VAC

3.15 A

TOPSwitch-JX

12 V, 2.5 A

RTN

82 µF

400 V

4.7 nF

1 kV

VR1

P6KE180A

85 - 264

VAC

CONTROL

5.1 MΩ

10 MΩ

5.1 MΩ

R15

14.3 kΩ

10 MΩ

C20

33 nF

50 V

Input Voltage (VAC) 85 230

Full Load Efﬁciency (%) 81.25 86.21

Average Efﬁciency (%) 85.13

No-load Input Power (mW) 60.8

115

83.94

84.97

61.98 74.74

Rev. F 05/15

TOP264-271

www.power.com

Clamp Conﬁguration – RZCD vs RCD

• An RZCD (Zener bleed) was selected over RCD to give higher

light load efﬁciency and lower no-load consumption.

The clamp network is formed by VR1, C4, R5 and D5. It limits

the peak drain voltage spike caused by leakage inductance to

below the BVDSS rating of the internal TOPSwitch-JX MOSFET.

This arrangement was selected over a standard RCD clamp to

improve light load efﬁciency and no-load input power.

In a standard RCD clamp C4 would be discharged by a parallel

resistor rather than a resistor and series Zener. In an RCD

clamp the resistor value of R5 is selected to limit the peak drain

voltage under full load and overload conditions. However under

light or no-load conditions this resistor value now causes the

capacitor voltage to discharge signiﬁcantly as both the leakage

inductance energy and switching frequency are lower. As the

capacitor has to be recharged to above the reﬂected output

voltage each switching cycle the lower capacitor voltage

represents wasted energy. It has the effect of making the

clamp dissipation appear as a signiﬁcant load just as if it were

connected to the output of the power supply.

The RZCD arrangement solves this problem by preventing the

voltage across the capacitor discharging below a minimum

value (deﬁned by the voltage rating of VR1) and therefore

minimizing clamp dissipation under light and no-load conditions.

Zener VR1 is shown as a high peak dissipation capable TVS

however a standard lower cost Zener may also be used due to

the low peak current that component experiences.

In many designs a resistor value of less than 50 W may be used

in series with C4 to damp out high frequency ringing and

improve EMI but this was not necessary in this case.

Feedback Conﬁguration

• A high CTR optocoupler was used to reduce secondary bias

currents and no-load input power.

• Low voltage, low current voltage reference IC used on

secondary-side to reduce secondary-side feedback current

and no-load input power.

• Bias winding voltage tuned to ~9 V at no-load, high-line to

reduce no-load input power.

Typically the feedback current into the CONTROL pin at

high-line is ~3 mA. This current is both sourced from the bias

winding (voltage across C10) and directly from the output. Both

of these represent a load on the output of the power supply.

To minimize the dissipation from the bias winding under no-load

conditions the number of bias winding turns and value of C7

was adjusted to give a minimum voltage across C7 of ~9 V.

This is the minimum required to keep the optocoupler biased

and the output in regulation.

To minimize the dissipation of the secondary-side feedback

circuit a high CTR (CTR of 300 – 600%) optocoupler type was

used. This reduces the secondary-side opto-led current from

~3 mA to <~1 mA and therefore the effective load on the output.

A standard 2.5 V TL431 voltage reference was replaced with the

1.24 V LMV431 to reduce the supply current requirement of this

component from 1 mA to 100 mA.

Output Rectiﬁer Choice

• Use of high VOR allows the use of a 60 V Schottky diode for

high efﬁciency and lower cost.

The higher BVDSS rating of the TOPSwitch-JX of 725 V

(compared to 600 V or 650 V rating of typical power MOSFETs)

allowed a higher transformer primary to secondary turns ratio

(reﬂected output voltage or VOR). This reduced the output diode

voltage stress and allowed the use of cheaper and more efﬁcient

60 V (vs 80 V or 100 V) Schottky diodes. The efﬁciency

improvement occurs due the lower forward voltage drop of the

lower voltage diodes. Two parallel connected axial 5 A, 60 V

Schottky rectiﬁer diodes were selected for both low-cost and

high efﬁciency. This allowed PCB heat sinking of the diode for

low cost while maintaining efﬁciency compared to a single

higher current TO-220 packaged diode mounted on a heat sink.

For this conﬁguration the recommendation is that each diode is

rated at twice the output current and that the diodes share a

common cathode PCB area for heat sinking so that their

temperatures track. In practice the diodes current share quite

effectively as can be demonstrated by monitoring their

individual temperatures.

Output Inductor Post Filter Soft-Finish

• Inductor L2 used to provide an output soft-ﬁnish and eliminate

a capacitor.

To prevent output overshoot during start-up the voltage

appearing across L2 is used to provide a soft-ﬁnish function.

When the voltage across L2 exceeds the forward drop of U2A

and D10 current ﬂows though the optocoupler LED and

provides feedback to the primary. This arrangement acts to

limit the rate of rise of the output voltage until it reaches

regulation and eliminates the capacitor that is typically placed

across U3 to provide the same function.

Key Application Considerations

TOPSwitch-JX vs. TOPSwitch-HX

Table 3 compares the features and performance differences

between TOPSwitch-JX and TOPSwitch-HX. Many of the new

features eliminate the need for additional discrete components.

Other features increase the robustness of design, allowing cost

savings in the transformer and other power components.

TOP264-271 Design Considerations

Power Table

The data sheet power table (Table 1) represents the maximum

practical continuous output power based on the following

conditions:

1. 12 V output.

2. Schottky or high efﬁciency output diode.

3. 135 V reﬂected voltage (VOR) and efﬁciency estimates.

4. A 100 VDC minimum DC bus for 85-265 VAC and 250 VDC

minimum for 230 VAC.

5. Sufﬁcient heat sinking to keep device temperature ≤110 °C.

Rev. F 05/15

TOP264-271

www.power.com

6. Power levels shown in the power table for the V package

device assume 6.45 cm2 of 610 g/m2 copper heat sink area

in an enclosed adapter, or 19.4 cm2 in an open frame.

The provided peak power depends on the current limit for the

respective device.

TOP264-271 Selection

Selecting the optimum TOP264-271 depends upon required

maximum output power, efﬁciency, heat sinking constraints,

system requirements and cost goals. With the option to

externally reduce current limit, TOP264-271 may be used for

lower power applications where higher efﬁciency is needed or

minimal heat sinking is available.

Input Capacitor

The input capacitor must be chosen to provide the minimum

DC voltage required for the TOP264-271 converter to maintain

regulation at the lowest speciﬁed input voltage and maximum

output power. Since TOP264-271 has a high DCMAX limit and an

optimized dual slope line feed forward for ripple rejection, it is

possible to use a smaller input capacitor. For TOP264-271, a

capacitance of 2 mF per watt is possible for universal input with

an appropriately designed transformer.

Primary Clamp and Output Reﬂected Voltage VOR

A primary clamp is necessary to limit the peak TOP264-271 drain

to source voltage. A Zener clamp requires few parts and takes

up little board space. For good efﬁciency, the clamp Zener

should be selected to be at least 1.5 times the output reﬂected

voltage VOR, as this keeps the leakage spike conduction time

short. When using a Zener clamp in a universal input application,

a VOR of less than 135 V is recommended to allow for the absolute

tolerances and temperature variations of the Zener. This will

ensure efﬁcient operation of the clamp circuit and will also keep

the maximum drain voltage below the rated breakdown voltage

of the TOP264-271 MOSFET. A high VOR is required to take full

advantage of the wider DCMAX of TOP264-271. An RCD (or

RCDZ) clamp provides tighter clamp voltage tolerance than a

Zener clamp and allows a VOR as high as 150 V. RCD clamp

dissipation can be minimized by reducing the external current

limit as a function of input line voltage (see Figure 19). The RCD

clamp is more cost effective than the Zener clamp but requires

more careful design (see Quick Design Checklist).

Output Diode

The output diode is selected for peak inverse voltage, output

current, and thermal conditions in the application (including heat

sinking, air circulation, etc.). The higher DCMAX of TOP264-271,

along with an appropriate transformer turns ratio, can allow the

use of a 80 V Schottky diode for higher efﬁciency on output

voltages as high as 15 V.

Bias Winding Capacitor

Due to the low frequency operation at no-load, a bias winding

capacitance of 10 mF minimum is recommended. Ensure a

minimum bias winding voltage of >9 V at zero load for correct

operation and output voltage regulation.

Soft-Start

Generally, a power supply experiences maximum stress at

start-up before the feedback loop achieves regulation. For a

period of 17 ms, the on-chip soft-start linearly increases the

drain peak current and switching frequency from their low

starting values to their respective maximum values. This

causes the output voltage to rise in an orderly manner, allowing

time for the feedback loop to take control of the duty cycle.

This reduces the stress on the TOP264-271 MOSFET, clamp

circuit and output diode(s), and helps prevent transformer

saturation during start-up. Also, soft-start limits the amount of

output voltage overshoot and, in many applications, eliminates

the need for a soft-ﬁnish capacitor. Note that as soon as the

loop closes the soft-start function ceases even if this is prior to

the end of the 17 ms soft-start period.

EMI

The frequency jitter feature modulates the switching frequency

over a narrow band as a means to reduce conducted EMI peaks

associated with the harmonics of the fundamental switching

frequency. This is particularly beneﬁcial for average detection

TOPSwitch-HX vs. TOPSwitch-JX

Function TOPSwitch-HX TOPSwitch-JX TOPSwitch-JX Advantages

CONTROL current IC(OFF) at

0% duty cycle

IC(OFF) = IB + 3.4 mA

(TOP256-258)

IB = External bias current

IC(OFF) = IB + 1.6 mA

(TOP266-268)

• Reduced CONTROL current

• Better no-load performance (<0.1 W)

• Better standby performance

eDIP-12 / eSOP-12

packages

Not available Available • 66/132 kHz frequency option for DIP style heat sink

less designs

• Better thermal performance for increased power

capability over DIP-8 / SMD-8 packages

Breakdown voltage BVDSS Min. 700 V at TJ = 25 °C Min. 725 V at TJ = 25 °C • Simpliﬁes meeting customer derating requirements

(e.g. 80%)

• Extended line surge withstand

Fast AC reset 3 external transistor circuits

using the V pin

1 external transistor circuit

using the X pin

• Saves 5 components

Table 3. Comparison Between TOPSwitch-HX and TOPSwitch-JX.

Rev. F 05/15

TOP264-271

www.power.com

mode. As can be seen in Figures 27 and 28, the beneﬁts of jitter

increase with the order of the switching harmonic due to an

increase in frequency deviation. The FREQUENCY pin offers a

switching frequency option of 132 kHz or 66 kHz. In applications

that require heavy snubber on the drain node for reducing high

frequency radiated noise (for example, video noise sensitive

applications such as VCRs, DVDs, monitors, TVs, etc.), operating

at 66 kHz will reduce snubber loss, resulting in better efﬁciency.

Also, in applications where transformer size is not a concern, use

of the 66 kHz option will provide lower EMI and higher efﬁciency.

Note that the second harmonic of 66 kHz is still below 150 kHz,

above which the conducted EMI speciﬁcations get much tighter.

For 10 W or below, it is possible to use a simple inductor in place

of a more costly AC input common mode choke to meet

worldwide conducted EMI limits.

Transformer Design

It is recommended that the transformer be designed for

maximum operating ﬂux density of 3000 Gauss and a peak ﬂux

density of 4200 Gauss at maximum current limit. The turns ratio

should be chosen for a reﬂected voltage (VOR) no greater than

135 V when using a Zener clamp or 150 V (max) when using an

RCD clamp with current limit reduction with line voltage (overload

protection). For designs where operating current is signiﬁcantly

lower than the default current limit, it is recommended to use an

externally set current limit close to the operating peak current to

reduce peak ﬂux density and peak power (see Figure 18).

Standby Consumption

Frequency reduction can signiﬁcantly reduce power loss at light

or no-load, especially when a Zener clamp is used. For very

low secondary power consumption, use a TL431 regulator for

feedback control. A typical TOP264-271 circuit automatically

enters MCM mode at no-load and the low frequency mode at

light load, which results in extremely low losses under no-load

or standby conditions.

High Power Designs

The TOP264-271 family contains parts that can deliver up to

162 W. High power designs need special considerations.

Guidance for high power designs can be found in the Design

Guide for TOP264-271 (AN-47).

TOP264-271 Layout Considerations

The TOP264-271 has multiple pins and may operate at high

power levels. The following guidelines should be carefully

followed.

Primary Side Connections

Use a single point (Kelvin) connection at the negative terminal of

the input ﬁlter capacitor for the SOURCE pin and bias winding

return. This improves surge capabilities by returning surge

currents from the bias winding directly to the input ﬁlter capacitor.

The CONTROL pin bypass capacitor should be located as

close as possible to the SOURCE and CONTROL pins, and its

SOURCE connection trace should not be shared by the main

MOSFET switching currents. All SOURCE pin referenced

components connected to the VOLTAGE MONITOR (V) pin or

EXTERNAL CURRENT LIMIT (X) pin should also be located

closely between their respective pin and SOURCE. Once again,

the SOURCE connection trace of these components should not

be shared by the main MOSFET switching currents. It is very

critical that SOURCE pin switching currents are returned to the

input capacitor negative terminal through a separate trace that is

not shared by the components connected to CONTROL,

VOLTAGE MONITOR or EXTERNAL CURRENT LIMIT pins. This

is because the SOURCE pin is also the controller ground

reference pin. Any traces to the VOLTAGE MONITOR,

EXTERNAL CURRENT LIMIT or CONTROL pins should be kept

as short as possible and away from the DRAIN trace to prevent

noise coupling. Voltage monitor resistors (RLS in Figures 14, 15,

19, 22, 23, 26, 30) and primary-side OVP circuit components

VZOV/ROV in Figures (29, 30) should be located close to the

VOLTAGE MONITOR pin to minimize the trace length on the

VOLTAGE MONITOR pin side. Resistors connected to the

VOLTAGE MONITOR or EXTERNAL CURRENT LIMIT pin should

be connected as close to the bulk capacitor positive terminal as

possible while routing these connections away from the power

switching circuitry. In addition to the 47 mF CONTROL pin

Figure 28. TOPSwitch-JX Full Range EMI Scan (132 kHz with Jitter) with

Identical Circuitry and Conditions.

-20

-10

0.15 1 10 30

Frequency (MHz)

Amplitude (dBµV)

PI-2576-010600

EN55022B (QP)

EN55022B (AV)

EN55022B (QP)

EN55022B (AV)

-20

-10

0.15 1 10 30

Frequency (MHz)

Amplitude (dBµV)

PI-5583-090309

TOPSwitch-JX (with jitter)

Figure 27. Fixed Frequency Operation without Jitter.

Rev. F 05/15

TOP264-271

www.power.com

capacitor, a high frequency bypass capacitor (CBP) in parallel

should be used for better noise immunity. The feedback

optocoupler output should also be located close to the

CONTROL and SOURCE pins of TOP264-271 and away from the

drain and clamp component traces. The primary-side clamp

circuit should be positioned such that the loop area from the

transformer end (shared with DRAIN) and the clamp capacitor is

minimized. The bias winding return node should be connected

via a dedicated trace directly to the bulk capacitor and not to

the SOURCE pins. This ensures that surge currents are routed

away from the SOURCE pins of the TOPSwitch-JX.

Y Capacitor

The Y capacitor should be connected close to the secondary

output return pin(s) and the positive primary DC input pin of the

transformer. If the Y capacitor is returned to the negative end of

the input bulk capacitor (rather than the positive end) a dedicated

trace must be used to make this connection. This is to “steer”

leakage currents away from the SOURCE pins in case of a

common-mode surge event.

Heat Sinking

The exposed pad of the E package (eSIP-7C), K package

(eSOP-12) and the V package (eDIP-12) are internally electrically

tied to the SOURCE pin. To avoid circulating currents, a heat

sink attached to the exposed pad should not be electrically tied

to any primary ground/source nodes on the PC board. On

double sided boards, top side and bottom side areas connected

with vias can be used to increase the effective heat sinking

area. The K package exposed pad may be directly soldered to

a copper area for optimum thermal transfer. In addition,

sufﬁcient copper area should be provided at the anode and

cathode leads of the output diode(s) for heat sinking. In Figure

29, a narrow trace is shown between the output rectiﬁer and

output ﬁlter capacitor. This trace acts as a thermal relief

between the rectiﬁer and ﬁlter capacitor to prevent excessive

heating of the capacitor.

Quick Design Checklist

In order to reduce the no-load input power of TOP264-271

designs, the VOLTAGE MONITOR pin operates at very low

current. This requires careful layout considerations when

designing the PCB to avoid noise coupling. Traces and

components connected to the VOLTAGE MONITOR pin should

not be adjacent to any traces carrying switching currents.

These include the drain, clamp network, bias winding return or

power traces from other converters. If the line-sensing features

are used, then the sense resistors must be placed within 10 mm

of the VOLTAGE MONITOR pin to minimize the VOLTAGE

MONITOR pin node area. The DC bus should then be routed to

the line-sense resistors. Note that external capacitance must

not be connected to the VOLTAGE MONITOR pin as this may

cause misoperation of the VOLTAGE MONITOR pin related

functions. As with any power supply design, all TOP264-271

designs should be veriﬁed on the bench to make sure that

components speciﬁcations are not exceeded under worst-case

conditions. The following minimum set of tests is strongly

recommended:

1. Maximum drain voltage – Verify that peak VDS does not

exceed 675 V at highest input voltage and maximum

overload output power. Maximum overload output power

occurs when the output is overloaded to a level just before

the power supply goes into auto-restart (loss of regulation).

2. Maximum drain current – At maximum ambient temperature,

maximum input voltage and maximum output load, verify

drain current waveforms at start-up for any signs of trans-

former saturation and excessive leading edge current spikes.

TOP264-271 has a leading edge blanking time of 220 ns to

prevent premature termination of the ON-cycle. Verify that

the leading edge current spike is below the allowed current

limit envelope (see Figure 34) for the drain current waveform

at the end of the 220 ns blanking period.

3. Thermal check – At maximum output power, both minimum

and maximum voltage and ambient temperature; verify that

temperature speciﬁcations are not exceeded for TOP264-

271, transformer, output diodes and output capacitors.

Enough thermal margin should be allowed for the part-to-

part variation of the RDS(ON) of TOP264-271, as speciﬁed in

the data sheet. The margin required can either be calculated

from the values in the parameter table or it can be accounted

for by connecting an external resistance in series with the

DRAIN pin and attached to the same heat sink, having a

resistance value that is equal to the difference between the

measured RDS(ON) of the device under test and the worst-case

maximum speciﬁcation.

Design Tools

Up-to-date information on design tools can be found at the

Power Integrations website: www.power.com

Rev. F 05/15

TOP264-271

www.power.com

Figure 29. Layout Considerations for TOPSwitch-JX using V Package and Operating at 132 kHz.

Figure 30. Layout Considerations for TOPSwitch-JX using K Package and Operating at 132 kHz.

+–

PI-5752-061311

Output

Rectifiers

Transformer L2

C18

C16

C11

VR1

R5D5

C10

R16

ROV

RLS1

RPL1

RPL2

RLS2

R12

RIL

CBP

VZOV

Y-

Capacitor

C17

Output Filter

Capacitors

HF LC

Post-Filter

Input Filter

Capacitor

Clamp Circuit

Maximize Copper Area

for Optimum Heat Sinking

OUT

–

+–

PI-6173-081412

Output

Rectiers

Transformer L2

C18

C16

C11

VR1

C10

R16CBP

CB+

ROV

RLS1

RPL1

RIL

RPL2

RLS2

R12

RIL

CBP

VZOV

Y-

Capacitor

C17

Output Filter

Capacitors

HF LC

Post-Filter

Input Filter

Capacitor

Clamp Circuit

Maximize Copper Area

for Optimum Heat Sinking

OUT

–

Rev. F 05/15

TOP264-271

www.power.com

+- DC

OUT PI-5793-030910

Y-

Capacitor

Isolation Barrier

Output

Rectifier

RLS2

RLS1

RPL1

ROV

R6 D5

C6 R7

CBP

RIL

R10

VZOV

RPL2

VR1

HS1

R15

R13

R17

JP2

C21

R21

R20

C19

HS2

C16 R12

C18

C17

Output Filter

Capacitors

HF LC

Post-Filter

Input Filter

Capacitor

Transformer

Clamp

Circuit

Figure 31. Layout Considerations for TOPSwitch-JX using E Package and Operating at 132 kHz.

Rev. F 05/15

TOP264-271

www.power.com

Absolute Maximum Ratings(2)

DRAIN Pin Peak Voltage ..................................... -0.3 V to 725 V

DRAIN Pin Peak Current: TOP264 ................................... 2.08 A

DRAIN Pin Peak Current: TOP265 ................................... 2.72 A

DRAIN Pin Peak Current: TOP266 ................................... 4.08 A

DRAIN Pin Peak Current: TOP267 ................................... 5.44 A

DRAIN Pin Peak Current: TOP268 ................................... 6.88 A

DRAIN Pin Peak Current: TOP269 ................................... 7.73 A

DRAIN Pin Peak Current: TOP270 ................................... 9.00 A

DRAIN Pin Peak Current: TOP271 ................................. 11.10 A

CONTROL Pin Voltage ............................................ -0.3 V to 9 V

CONTROL Pin Current .................................................. 100 mA

VOLTAGE MONITOR Pin Voltage ........................... -0.3 V to 9 V

CURRENT LIMIT Pin Voltage .............................. -0.3 V to 4.5 V

FREQUENCY Pin Voltage ......................................-0.3 V to 9 V

Storage Temperature ...................................... -65 °C to 150 °C

Operating Junction Temperature ......................-40 °C to 150 °C

Lead Temperature(1) ........................................................260 °C

Notes:

1. 1/16 in. from case for 5 seconds.

2. Maximum ratings speciﬁed may be applied one at a time

without causing permanent damage to the product. Expo

sure to Absolute Maximum Rating conditions for extended

periods of time may affect product reliability.

Thermal Resistance

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

See Figure 35

(Unless Otherwise Speciﬁed)

Min Typ Max Units

Control Functions

Switching Frequency

in Full Frequency

Mode (average)

fOSC TJ = 25 °C

FREQUENCY Pin

Connected to SOURCE 119 132 145

kHz

FREQUENCY Pin

Connected to CONTROL 59.4 66 72.6

Frequency Jitter

Deviation Df132 kHz Operation ±5 kHz

66 kHz Operation ±2.5

Frequency Jitter

Modulation Rate fM250 Hz

Maximum Duty Cycle DCMAX IC = ICD1

IV ≤ IV(DC)

VV = 0 V 75 78 83

IV = 95 mA 30

Soft-Start Time tSOFT TJ = 25 °C 17 ms

PWM Gain DCREG

TJ = 25 °C

IB < IC < IC01

See Note C

TOP264-265 -62 -50 -40

%/mA

TOP266-268 -54 -44 -34

TOP269-271 -50 -40 -30

TJ = 25 °C

IC ≥ IC01

See Note A

TOP264-265 -61 -51 -41

TOP266-268 -60 -50 -40

TOP269-271 -57 -48 -38

PWM Gain

Temperature Drift See Note B -0.01 %/mA/°C

External Bias Current IB66 kHz Operation

TOP264-265 0.8 1.4 2.0

mATOP266-268 0.9 1.5 2.1

TOP269-271 1.0 1.6 2.2

Thermal Resistance: E Package

(qJA) .......................................... 105 °C/W(1)

(qJC) .............................................. 2 °C/W(2)

V Package

(qJA) ...........................68 °C/W(3), 58 °C/W(4)

(qJC) .............................................. 2 °C/W(2)

K Package

(qJA) ...........................45 °C/W(3), 38 °C/W(4)

(qJC) .............................................. 2 °C/W(2)

Notes:

1. Free standing with no heat sink.

2. Measured at the back surface of tab.

3. Soldered (including exposed pad for K package) to typical

application PCB with a heat sinking area of 0.36 sq. in. (232mm2),

2 oz. (610 g/m2) copper clad.

4. Soldered (including exposed pad for K package) to typical

application PCB with a heat sinking area of 1 sq. in. (645 mm2),

2 oz. (610 g/m2) copper clad.

Rev. F 05/15

TOP264-271

www.power.com

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

(Unless Otherwise Speciﬁed)

Min Typ Max Units

Control Functions (cont.)

External Bias Current IB132 kHz Operation

TOP264-265 0.9 1.5 2.1

mATOP266-268 1.2 1.8 2.4

TOP269-271 1.5 2.1 2.8

CONTROL Current at

0% Duty Cycle IC(OFF)

66 kHz Operation

TOP264-265 2.9 3.9

TOP266-268 3.1 4.1

TOP269-271 3.3 4.3

132 kHz Operation

TOP264-265 3.1 4.1

TOP266-268 3.4 4.4

TOP269-271 3.8 4.8

Dynamic Impedance ZCIC = 2.5 mA; TJ = 25 °C, See Figure 33 13 21 25 W

Dynamic Impedance

Temperature Drift 0.18 %/°C

CONTROL Pin Internal

Filter Pole 7 kHz

Upper Peak Current to

Set Current Limit Ratio kPS(UPPER)

TJ = 25 °C

See Note C 50 55 60 %

Lower Peak Current to

Set Current Limit Ratio kPS(LOWER)

TJ = 25 °C

See Note C 25 %

Multi-Cycle-

Modulation Switching

Frequency

fMCM(MIN) TJ = 25 °C 30 kHz

Minimum Multi-Cycle-

Modulation On Period TMCM(MIN) TJ = 25 °C 135 ms

Shutdown/Auto-Restart

CONTROL Pin

Charging Current IC(CH) TJ = 25 °C

VC = 0 V -5.0 -3.5 -1.0 mA

VC = 5 V -3.0 -1.8 -0.6

Charging Current

Temperature Drift See Note B 0.5 %/°C

Auto-Restart

Upper Threshold

Voltage

VC(AR)U 5.8 V

Auto-Restart Lower

Threshold Voltage VC(AR)L 4.5 4.8 5.1 V

Voltage Monitor (V) and External Current Limit (X) Inputs

Auto-Restart

Hysteresis Voltage VC(AR)HYST 0.8 1.0 V

Auto-Restart

Duty Cycle DCAR 2 4 %

Auto-Restart

Frequency fAR 0.5 Hz

Line Undervoltage

Threshold Current and

Hysteresis (V Pin)

IUV TJ = 25 °C

Threshold 22 25 27 mA

Hysteresis 14 mA

Line Overvoltage

Threshold Current and

Hysteresis (V Pin)

IOV TJ = 25 °C

Threshold 107 112 117 mA

Hysteresis 4 mA

Rev. F 05/15

TOP264-271

www.power.com

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

(Unless Otherwise Speciﬁed)

Min Typ Max Units

Voltage Monitor (V) and External Current Limit (X) Inputs (cont.)

Output Overvoltage

Latching Shutdown

Threshold Current

IOV(LS) TJ = 25 °C 269 336 403 mA

V Pin Remote

ON/OFF Voltage VV(TH) TJ = 25 °C 0.8 1.0 1.6 V

X Pin Remote-ON/OFF

and Latch Reset

Negative Threshold

Current and Hysteresis

IREM(N) TJ = 25 °C

Threshold -35 -27 -20

Hysteresis 5

V Pin Short-Circuit

Current IV(SC) TJ = 25 °C VV = VC300 400 500 mA

X Pin Short-Circuit

Current IX(SC) VX = 0 V

Normal Mode -260 -200 -140

Auto-Restart Mode -95 -75 -55

V Pin Voltage

(Positive Current) VVIV = IOV TOP264-TOP271 2.83 3.0 3.25 V

V Pin Voltage Hysteresis

(Positive Current) VV(HYST) IV = IOV 0.2 0.5 V

X Pin Voltage

(Negative Current) VX

IX = -50 mA 1.23 1.30 1.37 V

IX = -150 mA 1.15 1.22 1.29

Maximum Duty Cycle

Reduction Onset

Threshold Current

IV(DC) IC ≥ IB, TJ = 25 °C 18.9 22.0 24.2 mA

Maximum Duty Cycle

Reduction Slope TJ = 25 °C

IV(DC) < IV <48 mA -1.0

%/mA

IV ≥48 mA -0.25

Remote-OFF DRAIN

Supply Current ID(RMT) VDRAIN = 150 V

X or V Pin Floating 0.6 1.0

V Pin Shorted to

CONTROL 1.0 1.6

Remote-ON Delay tR(ON)

From Remote-ON to Drain

Turn-On

See Note C

66 kHz 3.0

132 kHz 1.5

Remote-OFF

Set-up Time tR(OFF)

Minimum Time Before Drain

Turn-On to Disable Cycle

See Note C

66 kHz 3.0

132 kHz 1.5

Frequency Input

FREQUENCY Pin

Threshold Voltage VFSee Note B 2.9 V

FREQUENCY Pin

Input Current IFTJ = 25 °C VF = VC10 55 90 mA

Rev. F 05/15

TOP264-271

www.power.com

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

(Unless Otherwise Speciﬁed)

Min Typ Max Units

Circuit Protection

Self Protection

Current Limit

(See Note D)

ILIMIT

TOP264

TJ = 25 °C di/dt = 270 mA/ms 1.209 1.30 1.391

TOP265

TJ = 25 °C di/dt = 350 mA/ms 1.581 1.70 1.819

TOP266

TJ = 25 °C di/dt = 530 mA/ms 2.371 2.55 2.728

TOP267

TJ = 25 °C di/dt = 625 mA/ms 2.800 3.01 3.222

TOP268

TJ = 25 °C di/dt = 675 mA/ms 3.023 3.25 3.478

TOP269

TJ = 25 °C di/dt = 720 mA/ms 3.236 3.48 3.723

TOP270

TJ = 25 °C di/dt = 870 mA/ms 3.906 4.20 4.494

TOP271

TJ = 25 °C di/dt = 1065 mA/ms 4.808 5.17 5.532

Initial Current Limit IINIT See Note C 0.70 ×

ILIMIT(MIN)

Power Coefﬁcient PCOEFF

TJ = 25 °C,

See Note E

IX ≤ - 165 mA 0.9 × I2f I2f 1.2 × I2f

A2kHz

IX ≤ - 117 mA 0.9 × I2f I2f 1.2 × I2f

Leading Edge

Blanking Time tLEB TJ = 25 °C, See Figure 34 220 ns

Current Limit Delay tIL(D) 100 ns

Thermal Shutdown

Temperature 135 142 150 °C

Thermal Shutdown

Hysteresis 75 °C

Power-Up Reset

Threshold Voltage VC(RESET) Figure 35 (S1 Open Condition) 1.75 3.0 4.25 V

Rev. F 05/15

TOP264-271

www.power.com

Parameter Symbol

Conditions

SOURCE = 0 V; TJ = -40 to 125 °C

(Unless Otherwise Speciﬁed)

Min Typ Max Units

Output

ON-State

Resistance RDS(ON)

TOP264

ID = 150 mA

TJ = 25 °C 5.4 6.25

TJ = 100 °C 8.35 9.70

TOP265

ID = 200 mA

TJ = 25 °C 4.1 4.70

TJ = 100 °C 6.3 7.30

TOP266

ID = 300 mA

TJ = 25 °C 2.8 3.20

TJ = 100 °C 4.1 4.75

TOP267

ID = 400 mA

TJ = 25 °C 2.0 2.30

TJ = 100 °C 3.1 3.60

TOP268

ID = 500 mA

TJ = 25 °C 1.7 1.95

TJ = 100 °C 2.5 2.90

TOP269

ID = 600 mA

TJ = 25 °C 1.45 1.70

TJ = 100 °C 2.25 2.60

TOP270

ID = 700 mA

TJ = 25 °C 1.20 1.40

TJ = 100 °C 1.80 2.10

TOP271

ID = 800 mA

TJ = 25 °C 1.05 1.20

TJ = 100 °C 1.55 1.80

DRAIN Supply Voltage TJ ≤ 85 °C, See Note F

OFF-State Drain

Leakage Current IDSS

VV = Floating, Device Not Switching,

VDS = 580 V, TJ = 125 °C470 mA

Breakdown

Voltage BVDSS

VV = Floating, Device Not Switching,

TJ = 25 °C, See Note G 725 V

Rise Time tRMeasured in a Typical Flyback

Converter Application

100 ns

Fall Time tF50 ns

Supply Voltage Characteristics

Control Supply/

Discharge Current

ICD1

Output

MOSFET

Enabled

VX, VV =

0 V

66 kHz

Operation

TOP264-265 0.6 1.2 2.0

TOP266-268 0.9 1.4 2.3

TOP269-271 1.1 1.6 2.5

132 kHz

Operation

TOP264-265 0.8 1.4 2.1

TOP266-268 1.2 1.7 2.4

TOP269-271 1.5 2.1 2.9

ICD2

Output MOSFET Disabled

VX, VV = 0 V 0.3 0.5 1.2

Rev. F 05/15

TOP264-271

www.power.com

NOTES:

A. Derived during test from the parameters DCMAX, IB and IC(OFF) at 132 kHz.

B. For speciﬁcations with negative values, a negative temperature coefﬁcient corresponds to an increase in magnitude with increasing

temperature, and a positive temperature coefﬁcient corresponds to a decrease in magnitude with increasing temperature.

C. Guaranteed by characterization. Not tested in production.

D. For externally adjusted current limit values, please refer to Figures 36 and 37 (Current Limit vs. External Current Limit Resistance)

in the Typical Performance Characteristics section. The tolerance speciﬁed is only valid at full current limit.

E. I2f calculation is based on typical values of ILIMIT and fOSC, i.e. ILIMIT(TYP)

2 × fOSC, where fOSC = 66 kHz or 132 kHz depending on FREQUENCY

pin connection. See fOSC speciﬁcation for detail.

F. The device will start up at 18 VDC drain voltage. The capacitance of electrolytic capacitors drops signiﬁcantly at temperatures below

0 °C. For reliable start-up at 18 V in sub-zero temperatures, designers must ensure that circuit capacitors meet recommended

capacitance values.

G. Breakdown voltage may be checked against minimum BVDSS speciﬁcation by ramping the DRAIN pin voltage up to but not exceeding

minimum BVDSS.

Rev. F 05/15

TOP264-271

www.power.com

Figure 32. Duty Cycle Measurement.

Figure 33. CONTROL Pin I-V Characteristic. Figure 34. Drain Current Operating Envelope.

Figure 35. TOPSwitch-JX General Test Circuit.

PI-2039-033001

DRAIN

VOLTAGE

0 V

90%

10%

90%

D = t1

120

100

5 6 7 8 9

CONTROL Pin Voltage (V)

CONTROL Pin Current (mA)

Slope

Dynamic

Impedance =

PI-4737-061207

PI-5534-072409

5-50 V

40 V

0.1 µF47 µF

470 Ω

5 W

(X and V Pins)

470 Ω

0-300 kΩ

0-60 kΩ

NOTES: 1. This test circuit is not applicable for current limit or output characteristic measurements.

SFX

CONTROL

0-15 V

0.8

1.3

1.2

1.1

0.9

0.8

1.0

0 1 2 6 83

Time (µs)

DRAIN Current (normalized)

PI-4758-061407

4 5 7

0.7

0.6

0.5

0.4

0.3

0.2

0.1

IINIT(MIN)

tLEB (Blanking Time)

Rev. F 05/15

TOP264-271

www.power.com

Figure 36. Normalized Current Limit vs. X Pin Current.

Figure 37. Normalized Current Limit vs. External Current Limit Resistance.

PI-5581-090309

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1.1

-200 -150 -100 -50 0

Normalized Current Limit

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1.1

Typical

Notes:

1. Maximum and Minimum levels are

based on characterization.

2. TJ = 0 OC to 125 OC.

Minimum

Maximum

Normalized Current Limit

Normalized di/dt

( µA)

PI-5582-090309

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1.1

0 5 10 15 20 25 30 35 40 45

( kΩ )

Normalized Current Limit (A)

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1.1

Normalized di/dt (mA/

Notes:

1. Maximum and Minimum levels are

based on characterization.

2. T

= 0

C to 125

3. Includes the variation of X pin voltage.

Typical

Maximum

Minimum

Normalized Current Limit

Normalized di/dt

Typical Performance Characteristics

Rev. F 05/15

TOP264-271

www.power.com

Typical Performance Characteristics (cont.)

1.1

1.0

0.9

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

Breakdown Voltage

(Normalized to 25

°C)

PI-176B-033001

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-4759-061407

Output Frequency

(Normalized to 25

°C)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-4760-061407

Current Limit

(Normalized to 25

°C)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0255075 100 125 150

Junction Temperature (°C)

PI-4739-061507

Current Limit

(Normalized to 25

ϒC)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-4761-061407

Overvoltage Threshold

(Normalized to 25

°C)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-4762-100610

Undervoltage Threshold

(Normalized to 25

°C)

Figure 38. Breakdown Voltage vs. Temperature. Figure 39. Frequency vs. Temperature.

Figure 40. Internal Current Limit vs. Temperature.

Figure 41. External Current Limit vs. Temperature with RIL = 10.5 kW.

Figure 42. Overvoltage Threshold vs. Temperature. Figure 43. Undervoltage Threshold vs. Temperature.

Rev. F 05/15

TOP264-271

www.power.com

Typical Performance Characteristics (cont.)

4.5

5.5

0 100 200 500400300

VOLTAGE-MONITOR Pin Current (µA)

VOLTAGE MONITOR Pin Voltage (V)

PI-4740-060607

2.5

3.5

1.6

1.0

1.4

1.2

-200 -150 -50-100 0

EXTERNAL CURRENT LIMIT Pin Current (µA)

EXTERNAL CURRENT LIMIT

Pin Voltage (V)

PI-4741-110907

0.4

0.2

0.6

0.8

VX = 1.354 - 1147.5 × IX + 1.759 × 106 ×

(IX)2 with -180 µA < IX < -25 µA

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-4763-072208

CONTROL Current

(Normalized to 25

°C)

1.2

1.0

0.8

0.6

0.4

0.2

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

PI-4764-061407

Onset Threshold Current

(Normalized to 25

°C)

Figure 45. EXTERNAL CURRENT LIMIT Pin Voltage vs. Current.

Figure 46. Control Current Out at 0% Duty Cycle vs. Temperature. Figure 47. Maximum Duty Cycle Reduction Onset Threshold

Current vs. Temperature.

Figure 44. VOLTAGE-MONITOR Pin vs. Current.

-0.5

0.5

-2.5

020 40 60 80 100

Drain Pin Voltage (V)

CONTROL Pin Current (mA)

PI-4744-072208

-1.5

-2

-1

VC = 5 V

Figure 48. Output Characteristics. Figure 49. IC vs. DRAIN Voltage.

02 4 6 8 10 12 14 16 18 20

Drain Voltage (V)

DRAIN Current (A)

PI-5569-110409

TCASE = 25 °C

TCASE = 100 °C

TOP271 1.62

TOP270 1.42

TOP269 1.17

TOP268 1.00

TOP267 0.85

TOP266 0.61

TOP265 0.42

TOP264 0.32

Scaling Factors:

Rev. F 05/15

TOP264-271

www.power.com

Typical Performance Characteristics (cont.)

Figure 50. COSS vs. DRAIN Voltage. Figure 51. DRAIN Capacitance Power.

Figure 52. Remote-OFF DRAIN Supply Current vs. Temperature.

0 100 200 300 400 500 600

100

1000

10000

PI-5570-090309

Drain Pin Voltage (V)

DRAIN Capacitance (pF)

TOP271 1.62

TOP270 1.42

TOP269 1.17

TOP268 1.00

TOP267 0.85

TOP266 0.61

TOP265 0.42

TOP264 0.32

Scaling Factors:

500

400

200

100

300

0 200100 400 500 600300 700

Drain Pin Voltage (V)

Power (mW)

PI-5571-090309

132 kHz

66 kHz

TOP271 1.62

TOP270 1.42

TOP269 1.17

TOP268 1.00

TOP267 0.85

TOP266 0.61

TOP265 0.42

TOP264 0.32

Scaling Factors:

1.2

0.8

1.0

-50 0-25 5025 10075 125 150

Junction Temperature (°C)

Remote OFF DRAIN Supply Current

(Normalized to 25

°C)

PI-4745-061407

0.2

0.4

0.6

Rev. F 05/15

TOP264-271

www.power.com

PI-4917-061510

MOUNTING HOLE PATTERN

(not to scale)

PIN 7

PIN 1

0.100 (2.54) 0.100 (2.54)

0.059 (1.50)

0.050 (1.27)

0.100 (2.54)

0.155 (3.93)

0.020 (0.50)

Notes:

1. Dimensioning and tolerancing per ASME Y14.5M-1994.

2. Dimensions noted are determined at the outermost

extremes of the plastic body exclusive of mold flash,

tie bar burrs, gate burrs, and interlead flash, but including

any mismatch between the top and bottom of the plastic

body. Maximum mold protrusion is 0.007 [0.18] per side.

3. Dimensions noted are inclusive of plating thickness.

4. Does not include inter-lead flash or protrusions.

5. Controlling dimensions in inches (mm).

0.403 (10.24)

0.397 (10.08)

0.325 (8.25)

0.320 (8.13)

0.050 (1.27)

FRONT VIEW

0.070 (1.78) Ref.

Pin #1

I.D.

0.016 (0.41)

Ref.

0.290 (7.37)

Ref.

0.047 (1.19)

0.100 (2.54)

0.519 (13.18)

Ref.

0.198 (5.04) Ref.

0.264 (6.70)

Ref.

0.118 (3.00)

6×

0.140 (3.56)

0.120 (3.05)

0.021 (0.53)

0.019 (0.48)

0.378 (9.60)

Ref. 0.019 (0.48) Ref.

0.060 (1.52)

Ref.

0.048 (1.22)

0.046 (1.17)

0.081 (2.06)

0.077 (1.96)

0.207 (5.26)

0.187 (4.75)

0.033 (0.84)

0.028 (0.71)

0.016 (0.41)

0.011 (0.28)

eSIP-7C (E Package)

10° Ref.

All Around

0.020 M 0.51 M C

0.010 M 0.25 M C A B

SIDE VIEW

END VIEW

BACK VIEW

0.023 (0.58)

0.027 (0.70)

DETAIL A

Detail A

Rev. F 05/15

TOP264-271

www.power.com

SIDE VIEW

END VIEW

11×

Detail A

0.059 [1.50]

Ref, typ.

Notes:

1. Dimensioning and tolerancing per

ASME Y14.5M-1994.

2. Dimensions noted are determined at the outer-

most extremes of the plastic body exclusive of

mold flash, tie bar burrs, gate burrs, and interlead

flash, but including any mismatch between the

top and bottom of the plastic body. Maximum

mold protrusion is 0.007 [0.18] per side.

3. Dimensions noted are inclusive of plating

thickness.

4. Does not include inter-lead flash or protrusions.

5. Controlling dimensions in inches [mm].

6. Datums A and B to be determined at Datum H.

7. Measured with the leads constrained to be

perpendicular to Datum C.

8. Measured with the leads unconstrained.

9. Lead numbering per JEDEC SPP-012.

10. Exposed pad is nominally located at the center-

line of Datums A and B. “Max” dimensions

noted include both size and positional tolerances.

eDIP-12B (V Package)

TOP VIEW

0.325 [8.26]

Max.

Pin #1 I.D.

(Laser Marked)

0.350 [8.89]

0.070 [1.78]

1 2 3 4 6

12 11 10 9 8 7 7 12

0.059 [1.50]

Ref, typ.

0.225 [5.72]

Max.

0.192 [4.87]

Ref.

0.436 [11.08]

0.406 [10.32]

0.023 [0.58]

0.018 [0.46]

0.092 [2.34]

0.086 [2.18]

0.049 [1.23]

0.046 [1.16] 0.022 [0.56]

Ref.

0.031 [0.80]

0.028 [0.72]

0.016 [0.41]

0.011 [0.28]

0.400 [10.16]

3 4

0.400 [10.16]

0.010 [0.25] Ref.

Seating Plane

0.412 [10.46]

Ref.

0.120 [3.05]

Ref.

0.306 [7.77]

Ref.

0.104 [2.65] Ref.

0.356 [9.04]

Ref.

0.019 [0.48]

Ref.

0.028 [0.71]

Ref.

0.020 [0.51]

Ref.

BOTTOM VIEW

0.010 [0.25] M C A B

0.004 [0.10] C B

DETAIL A (Scale = 9X)

0.004 [0.10] C A

5 ±

° 4°

PI-5556a-100311

0.07 [1.78] 0.03 [0.76]

0.400 [10.16]

Mounting

Hole Pattern

Dimensions

Drill Hole 0.03 [0.76]

Round Pad 0.05 [1.27]

Solder Mask 0.056 [1.42]

Rev. F 05/15

TOP264-271

www.power.com

SIDE VIEW END VIEW

11×

PI-5748a-100311

Notes:

1. Dimensioning and tolerancing per ASME Y14.5M-1994.

2. Dimensions noted are determined at the outermost

extremes of the plastic body exclusive of mold flash,

tie bar burrs, gate burrs, and interlead flash, but

including any mismatch between the top and bottom of

the plastic body. Maximum mold protrusion is 0.007

[0.18] per side.

3. Dimensions noted are inclusive of plating thickness.

4. Does not include interlead flash or protrusions.

5. Controlling dimensions in inches [mm].

6. Datums A and B to be determined at Datum H.

7. Exposed pad is nominally located at the centerline of

Datums A and B. “Max” dimensions noted include both

size and positional tolerances.

eSOP-12B (K Package)

TOP VIEW BOTTOM VIEW

Pin #1 I.D.

(Laser Marked)

0.023 [0.58]

0.018 [0.46]

0.006 [0.15]

0.000 [0.00]

0.098 [2.49]

0.086 [2.18]

0.092 [2.34]

0.086 [2.18]

0.032 [0.80]

0.029 [0.72]

Seating

Plane

Detail A

Seating plane to

package bottom

standoff

0.034 [0.85]

0.026 [0.65]

0.049 [1.23]

0.046 [1.16]

3 4

0.460 [11.68]

0.400 [10.16]

0.070 [1.78]

0.306 [7.77]

Ref.

0.350 [8.89]

0.010 [0.25]

Ref.

Gauge Plane

Seating Plane

0.055 [1.40] Ref.

0.010 [0.25]

0.059 [1.50]

Ref, Typ 0.225 [5.72]

Max.

0.019 [0.48]

Ref.

0.022 [0.56]

Ref.

0.020 [0.51]

Ref.

0.028 [0.71]

Ref.

0.325 [8.26]

Max.

0.356 [9.04]

Ref.

0.059 [1.50]

Ref, Typ

0.120 [3.05] Ref

0.010 (0.25) M C A B

11×

0.016 [0.41]

0.011 [0.28]

DETAIL A (Scale = 9X)

0.008 [0.20] C

2X, 5/6 Lead Tips

0.004 [0.10] C

0.004 [0.10] C A 2X

0.004 [0.10] C B

0 -

° 8°

123 4 6 6 1

7 12

0.217 [5.51]

0.321 [8.15]

0.429 [10.90]

0.028 [0.71]

0.067 [1.70] Land Pattern

Dimensions

Rev. F 05/15

TOP264-271

www.power.com

Part Ordering Information

• TOPSwitch Product Family

• JX Series Number

• Package Identiﬁer

E Plastic eSIP-7C

V Plastic eDIP-12

K Plastic eSOP-12

• Pin Finish

G Halogen Free and RoHS Compliant

• Tape & Reel and Other Options

Blank Standard Conﬁgurations

TOP 264 E G - TL

Rev. F 05/15

TOP264-271

www.power.com

Revision Notes Date

A Release data sheet. 01/10

B Added eDIP parts. 01/10

B Page 4 “latching” changed to “hysteretic”. Table 3 updated. 03/10

B Sentence in ‘Line-Sense Resistor Values’ section updated. 07/10

C Added K package parts. 11/10

D Updated K and V package drawings. 06/11

E Added eDIP-12B and eSOP-12B packages. Removed eDIP-12 and eSOP-12 packages. 10/11

E Updated Figure 2 and K package layout. 08/12

F Updated with new Brand Style. 05/15

For the latest updates, visit our website: www.power.com

Power Integrations reserves the right to make changes to its products at any time to improve reliability or manufacturability. Power

Integrations does not assume any liability arising from the use of any device or circuit described herein. POWER INTEGRATIONS MAKES

NO WARRANTY HEREIN AND SPECIFICALLY DISCLAIMS ALL WARRANTIES INCLUDING, WITHOUT LIMITATION, THE IMPLIED

WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE, AND NON-INFRINGEMENT OF THIRD PARTY RIGHTS.

Patent Information

The products and applications illustrated herein (including transformer construction and circuits external to the products) may be covered

by one or more U.S. and foreign patents, or potentially by pending U.S. and foreign patent applications assigned to Power Integrations.

A complete list of Power Integrations patents may be found at www.power.com. Power Integrations grants its customers a license under

certain patent rights as set forth at http://www.power.com/ip.htm.

Life Support Policy

POWER INTEGRATIONS PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR

SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF POWER INTEGRATIONS. As used herein:

1. A Life support device or system is one which, (i) is intended for surgical implant into the body, or (ii) supports or sustains life, and (iii)

whose failure to perform, when properly used in accordance with instructions for use, can be reasonably expected to result in signiﬁcant

injury or death to the user.

2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause

the failure of the life support device or system, or to affect its safety or effectiveness.

The PI logo, TOPSwitch, TinySwitch, LinkSwitch, LYTSwitch, InnoSwitch, DPA-Switch, PeakSwitch, CAPZero, SENZero, LinkZero,

HiperPFS, HiperTFS, HiperLCS, Qspeed, EcoSmart, Clampless, E-Shield, Filterfuse, FluxLink, StakFET, PI Expert and PI FACTS are

Power Integrations Worldwide Sales Support Locations

World Headquarters

5245 Hellyer Avenue

San Jose, CA 95138, USA.

Main: +1-408-414-9200

Customer Service:

Phone: +1-408-414-9665

Fax: +1-408-414-9765

e-mail: usasales@power.com

China (Shanghai)

Rm 1601/1610, Tower 1,

Kerry Everbright City

No. 218 Tianmu Road West,

Shanghai, P.R.C. 200070

Phone: +86-21-6354-6323

Fax: +86-21-6354-6325

e-mail: chinasales@power.com

China (Shenzhen)

17/F, Hivac Building, No. 2,

Keji Nan 8th Road, Nanshan

District, Shenzhen, China,

518057

Phone: +86-755-8672-8689

Fax: +86-755-8672-8690

e-mail: chinasales@power.com

Germany

Lindwurmstrasse 114

80337 Munich

Germany

Phone: +49-895-527-39110

Fax: +49-895-527-39200

e-mail: eurosales@power.com

India

#1, 14th Main Road

Vasanthanagar

Bangalore-560052 India

Phone: +91-80-4113-8020

Fax: +91-80-4113-8023

e-mail: indiasales@power.com

Italy

Via Milanese 20, 3rd. Fl.

20099 Sesto San Giovanni (MI)

Italy

Phone: +39-024-550-8701

Fax: +39-028-928-6009

e-mail: eurosales@power.com

Japan

Kosei Dai-3 Bldg.

2-12-11, Shin-Yokohama,

Kohoku-ku

Yokohama-shi Kanagwan

222-0033 Japan

Phone: +81-45-471-1021

Fax: +81-45-471-3717

e-mail: japansales@power.com

Korea

RM 602, 6FL

Korea City Air Terminal B/D, 159-6

Samsung-Dong, Kangnam-Gu,

Seoul, 135-728, Korea

Phone: +82-2-2016-6610

Fax: +82-2-2016-6630

e-mail: koreasales@power.com

Singapore

51 Newton Road

#19-01/05 Goldhill Plaza

Singapore, 308900

Phone: +65-6358-2160

Fax: +65-6358-2015

e-mail: singaporesales@power.com

Taiwan

5F, No. 318, Nei Hu Rd., Sec. 1

Nei Hu Dist.

Taipei 11493, Taiwan R.O.C.

Phone: +886-2-2659-4570

Fax: +886-2-2659-4550

e-mail: taiwansales@power.com

First Floor, Unit 15, Meadway

Court, Rutherford Close,

Stevenage, Herts. SG1 2EF

United Kingdom

Phone: +44 (0) 1252-730-141

Fax: +44 (0) 1252-727-689

e-mail: eurosales@power.com