LM26480SQ-AAEV - National Semiconductor

September 12, 2008

LM26480

Externally Programmable Dual High-Current Step-Down

DC/DC and Dual Linear Regulators

General Description

The LM26480 is a multi-functional Power Management Unit,

optimized for low-power digital applications. This device inte-

grates two highly efficient 1.5A step-down DC/DC converters

and two 300 mA linear regulators. The LM26480 is offered in

a tiny 4 x 4 x 0.8mm LLP-24 pin package.

Key Specifications

Step-Down DC/DC Converter (Buck)

■1.5A output current

■VOUT from:

—Buck1 : 0.8V–2.0V @ 1.5A

—Buck2 : 1.0V–3.3V @ 1.5A

■Up to 96% efficiency

■±3% FB voltage accuracy

■2 MHz PWM switching frequency

■PWM - PFM automatic mode change under low loads

■Automatic soft start

Linear Regulators (LDO)

■VOUT of 1.0V–3.5V

■±3% FB voltage accuracy

■300 mA output current

■25 mV (typ) dropout

Features

■Compatible with advanced applications processors and

FPGAs

■2 LDOs for powering Internal processor functions and I/Os

■Precision internal reference

■Thermal overload protection

■Current overload protection

■24-lead 4 × 4 × 0.8mm LLP package

■External Power-On-Reset function for Buck1 and Buck2

■Undervoltage lock-out detector to monitor input supply

voltage

Applications

■Core digital power

■Applications processors

■Peripheral I/O power

Typical Application Circuit

30040401

LM26480 Externally Programmable Dual High-Current Step-Down DC/DC and Dual Linear

Regulators

30040402

FIGURE 1. Application Circuit

www.national.com 2

LM26480

Connection Diagrams and Package Mark Information

30040403

FIGURE 2. 24-Lead LLP Package (top view)

Note: The physical placement of the package marking will vary from part to part.

(*) UZXYTT format: ‘U’ – wafer fab code; ‘Z’ – assembly code; ’XY’ 2 digit date code; ‘TT” – die run code. See http://www.national.com/quality/

marking_conventions.html for more information on marking information.

Part Number Spec Quantity

LM26480SQ-AA NOPB 1000 tape and reel

LM26480SQX-AA NOPB 4500 tape and reel

3 www.national.com

LM26480

Pin Descriptions

LLP Pin No. Name I/O Type Description

1 VINLDO12 I PWR Analog Power for Internal Functions (VREF, BIAS, I2C, Logic)

2 SYNC I G/(D) Frequency Synchronization pin which allows the user to connect an

external clock signal to synchronize the PMIC internal oscillator.

Default OFF and must be grounded when not used. Contact National

Sales office to enable.

3 NPOR O D nPOR Power on reset pin for both Buck1 and Buck 2. Open drain logic

output 100K pullup resistor. nPOR is pulled to ground when the

voltages on these supplies are not good. See nPOR section for more

info.

4 GND_SW1 G G Buck1 NMOS Power Ground

5 SW1 O PWR Buck1 switcher output pin

6 VIN1 I PWR Power in from either DC source or Battery to Buck1

7 ENSW1 I D Enable Pin for Buck1 switcher, a logic HIGH enables Buck1. Pin cannot

be left floating.

8 FB1 I A Buck1 input feedback terminal

9 GND_C G G Non-switching core ground pin

10 AVDD I PWR Analog Power for Buck converters

11 FB2 I A Buck2 input feedback terminal

12 ENSW2 I D Enable Pin for Buck2 switcher, a logic HIGH enables Buck2. Pin cannot

be left floating.

13 VIN2 I PWR Power in from either DC source or Battery to Buck2

14 SW2 O PWR Buck2 switcher output pin

15 GND_SW2 G G Buck2 NMOS

16 ENLDO2 I D LDO2 enable pin, a logic HIGH enables LDO2. Pin cannot be left

floating.

17 ENLDO1 I D LDO1 enable pin, a logic HIGH enables LDO1. Pin cannot be left

floating.

18 GND_L G G LDO ground

19 VINLDO1 I PWR Power in from either DC source or battery to LDO1

20 LDO1 O PWR LDO1 Output

21 FBL1 I A LDO1 Feedback Terminal

22 FBL2 I A LDO2 Feedback Terminal

23 LDO2 O PWR LDO Output

24 VINLDO2 I PWR Power in from either DC source or battery to LDO2.

DAP DAP GND GND Connection isn't necessary for electrical performance, but it is

recommended for better thermal dissipation.

A: Analog Pin D: Digital Pin G: Ground Pin PWR: Power Pin I: Input Pin I/O: Input/Output Pin O: Output Pin

Power Block Operation Note

Power Block Input Enabled Disabled

VINLDO12 VIN+ VIN+ Always Powered

AVDD VIN+ VIN+ Always Powered

VIN1 VIN+ VIN+ or 0V

VIN2 VIN+ VIN+ or 0V

VINLDO1 ≤ VIN+ ≤ VIN+ If Enabled, Min VIN is 1.74V

VINLDO2 ≤ VIN+ ≤ VIN+ If Enabled, Min VIN is 1.74V

VIN+ is the largest potential voltage on the device.

www.national.com 4

LM26480

Absolute Maximum Ratings (Notes 1, 2)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

VINLDO12, VIN1, AVDD, VIN2,

VINLDO1, VINLDO2, ENSW1,

FB1, FB2, ENSW2, ENLDO1,

ENLDO2, SYNC, FBL1, FBL2 −0.3V to +6V

GND to GND SLUG ±0.3V

Power Dissipation (PD_MAX)

(TA=85°C, TMAX=125°C )

(Note 5) 1.17W

Junction Temperature (TJ-MAX)150°C

Storage Temperature Range −65°C to +150°C

Maximum Lead Temperature (Soldering) 260°C

ESD Ratings

Human Body Model (Note

4) 2 kV

Operating Ratings: Bucks (Notes 1, 2, 7)

VIN 2.8V to 5.5V

VEN 0 to (VIN + 0.3V)

Junction Temperature (TJ) Range –40°C to +125°C

Ambient Temperature (TA) Range

(Note 6)

−40°C to +85°C

Thermal Properties (Notes 3, 5, 6)

Junction-to-Ambient Thermal

Resistance (θJA) SQA024AG

34.1°C/W

General Electrical Characteristics (Notes 1, 2, 7, 13, 16)

Unless otherwise noted, VIN = 3.6V. Typical values and limits appearing in normal type apply for TJ = 25°C. Limits appearing in

boldface type apply over the entire junction temperature range for operation, −40°C to +125°C.

Symbol Parameter Conditions Min Typ Max Units

IQVINLDO12 Shutdown Current VIN = 3.6V 0.5 µA

VPOR Power-On Reset Threshold VDD Falling Edge(Note 16) 1.9 V

TSD Thermal Shutdown Threshold (Note 13) 160 °C

TSDH Themal Shutdown Hysteresis (Note 13) 20 °C

UVLO Under Voltage Lock Out Rising 2.9 V

Failing 2.7 V

Low Drop Out Regulators, LDO1 and LDO2

Unless otherwise noted, VIN = 3.6V, CIN = 1.0 µF, COUT = 0.47 µF. Typical values and limits appearing in normal type apply for

TJ = 25°C. Limits appearing in boldface type apply over the entire junction temperature range for operation, −40°C to +125°C.

(Notes 2, 7, 8, 9)

Symbol Parameter Conditions Min Typ Max Units

VIN Operational Voltage Range VINLDO1 and VINLDO2 PMOS

pins (Note 15) 1.74 5.5 V

VFB FB Voltage Accuracy −3 3%

ΔVOUT Line Regulation VIN = (VOUT + 0.3V) to 5.0V

(Note 12) Load Current = 1 mA 0.15 %/V

Load Regulation VIN = 3.6V,

Load Current = 1 mA to IMAX

0.011 %/mA

ISC Short Circuit Current Limit LDO1-2, VOUT = 0V 500 mA

VIN – VOUT Dropout Voltage Load Current = 50 mA

(Note 10) 25 200 mV

PSRR Power Supply Ripple Rejection F = 10 kHz, Load Current = IMAX 45 dB

θnSupply Output Noise 10 Hz < F < 100 kHz 150 µVrms

IQQuiescent Current “On” IOUT = 0 mA 40 150 µA

Quiescent Current “On” IOUT = 300 mA 60 200 µA

Quiescent Current “Off” EN is de-asserted 0.03 1 µA

TON Turn On Time Start up from shut-down 300 µsec

5 www.national.com

LM26480

Symbol Parameter Conditions Min Typ Max Units

COUT Output Capacitor Capacitance for stability

0°C ≤ TJ ≤ 125°C 0.33 0.47 µF

−40°C ≤ TJ ≤ 125°C 0.68 1.0 µF

ESR (Equivalent Series

Resistance) 5 500 mΩ

Buck Converters SW1, SW2

Unless otherwise noted, VIN = 3.6V, CIN = 10 µF, COUT = 10 µF, LOUT = 2.2 µH. Typical values and limits appearing in normal type

apply for TJ = 25°C. Limits appearing in boldface type apply over the entire junction temperature range for operation, −40°C to

+125°C. ((Notes 2, 7, 8, 9, 11, 14)

Symbol Parameter Conditions Min Typ Max Units

VFB (Note 14) Feedback Voltage −3 +3 %

VOUT Line Regulation 2.8 < VIN < 5.5

IO =10 mA

0.089 %/V

Load Regulation 100 mA < IO < IMAX 0.0013 %/mA

Eff Efficiency Load Current = 250 mA 96 %

ISHDN Shutdown Supply Current EN is de-asserted 0.01 1 µA

fOSC Internal Oscillator Frequency 2.0 2.4 MHz

IPEAK Buck1 Peak Switching Current Limit 2.0 2.4 A

Buck2 Peak Switching Current Limit 2.0 2.4

IQQuiescent Current “On” No load PFM Mode 33 µA

RDSON (P) Pin-Pin Resistance PFET 200 400 mΩ

RDSON (N) Pin-Pin Resistance NFET 180 400 mΩ

TON Turn On Time Start up from shut-down 500 µsec

CIN Input Capacitor Capacitance for stability 10 µF

COOutput Capacitor Capacitance for stability 10 µF

I/O Electrical Characteristics

Unless otherwise noted: Typical values and limits appearing in normal type apply for TJ = 25°C. Limits appearing in boldface

type apply over the entire junction temperature range for operation, TJ = 0°C to +125°C.

Symbol Parameter Conditions Limit Units

Min Max

VIL Input Low Level 0.4 V

VIH Input High Level 0.7*VDD V

Power On Reset Threshold/Function (POR)

Symbol Parameter Conditions Min Typ Max Units

nPOR nPOR = Power on reset for Buck1 and

Buck2

Default 60 msec

nPOR

Threshold

Percentage of Target voltage Buck1 or

Buck2

VBUCK1 AND VBUCK2 rising 92 %

VBUCK1 OR VBUCK2 falling 82

VOL Output Level Low Load = IOL = 500 µA 0.23 0.5 V

www.national.com 6

LM26480

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Operating Ratings are conditions under which operation

of the device is guaranteed. Operating Ratings do not imply guaranteed performance limits. For guaranteed performance limits and associated test conditions,

see the Electrical Characteristics.

Note 2: All voltages are with respect to the potential at the GND pin.

Note 3: Internal thermal shutdown circuitry protects the device from permanent damage. Thermal shutdown engages at TJ = 160°C (typ.) and disengages at TJ

= 140°C (typ.)

Note 4: The Human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. (MILSTD - 883 3015.7)

Note 5: In applications where high power dissipation and/or poor package thermal resistance is present, the maximum ambient temperature may have to be

derated. Maximum ambient temperature (TA-MAX) is dependent on the maximum operating junction temperature (TJ-MAX-OP = 125°C), the maximum power

dissipation of the device in the application (PD-MAX), and the junction-to-ambient thermal resistance of the part/package in the application (θJA), as given by the

following equation: TA-MAX = TJ-MAX-OP − (θJA × PD-MAX). See Applications section.

Note 6: Junction-to-ambient thermal resistance is highly application and board-layout dependent. In applications where high maximum power dissipation exists,

special care must be paid to thermal dissipation issues in board design.

Note 7: Min and Max limits are guaranteed by design, test, or statistical analysis. Typical numbers are not guaranteed, but do represent the most likely norm.

Note 8: CIN, COUT: Low-ESR Surface-Mount Ceramic Capacitors (MLCCs) used in setting electrical characteristics.

Note 9: The device maintains a stable, regulated output voltage without a load.

Note 10: Dropout voltage is the voltage difference between the input and the output at which the output voltage drops to 100 mV below its nominal value.

Note 11: Quiescent current is defined here as the difference in current between the input voltage source and the load at VOUT.

Note 12: VIN minimum for line regulation values is 1.8V.

Note 13: This specification is guaranteed by design.

Note 14: VIN ≥ VOUT + RDSON(P) (IOUT + 1/2 IRIPPLE). If these conditions are not met, voltage regulation will degrade as load increases.

Note 15: Pins 24, 19 can operate from VIN min of 1.74V to a VIN max of 5.5V. This rating is only for the series pass PMOS power FET. It allows the system design

to use a lower voltage rating if the input voltage comes from a buck output.

Note 16: VPOR is voltage at which the EPROM resets. This is different from the UVLO on VINLDO12, which is the voltage at which the regulators shut off; and

is also different from the nPOR function, which signals if the regulators are in a specified range.

7 www.national.com

LM26480

Typical Performance Characteristics — LDO

Output Voltage Change vs Temperature (LDO1)

VIN = 3.6V, VOUT = 2.5V, 100 mA load

30040466

Output Voltage Change vs Temperature (LDO2)

VIN = 3.6V, VOUT = 1.8V, 100 mA load

30040455

Load Transient

3.6 VIN, 2.5VOUT, 0 – 150 mA load

30040437

Load Transient

3.6 VIN, 2.5VOUT, 150–300 mA load

30040438

Line Transient (LDO1)

3.6 - 4.2 VIN, 2.5 VOUT, 100 mA load

30040439

Line Transient (LDO2)

3.6 – 4.2 VIN, 1.8VOUT, 150 mA load

30040440

www.national.com 8

LM26480

Enable Start-up time (LDO1)

0-3.6 VIN, 2.5 VOUT, 1 mA load

30040441

Enable Start-up time (LDO2)

0 – 3.6 VIN, 1.8VOUT, 1 mA load

30040442

9 www.national.com

LM26480

Typical Performance Characteristics — Buck VIN = 2.8V to 5.5V, TA = 25°C

Shutdown Current vs. Temp

30040443

Output Voltage vs. Supply Voltage

(VOUT = 1.2V)

30040444

Output Voltage vs. Supply Voltage

(VOUT = 2.0V)

30040445

Output Voltage vs. Supply Voltage

(VOUT = 3.0V)

30040446

www.national.com 10

LM26480

Typical Performance Characteristics — Buck Output Current transitions from PFM mode to PWM

mode for Buck 1

Efficiency vs. Output Current

(VOUT = 1.2V, L = 2.2 µH)

30040447

Efficiency vs. Output Current

(VOUT = 2.0V, L = 2.2 µH)

30040448

Output Current transitions from PWM mode to PFM mode

for Buck 2

Efficiency vs. Output Current

(VOUT = 3.0V, L = 2.2 µH)

30040449

Efficiency vs. Output Current

(VOUT = 3.5V, L = 2.2 µH)

30040450

11 www.national.com

LM26480

Typical Performance Characteristics — Buck

VIN= 3.6V, TA = 25°C, VOUT = 1.2V unless otherwise noted

Load Transient Response

VOUT = 1.2V (PWM Mode)

30040456

Mode Change by Load Transients

VOUT = 1.2V (PWM to PFM)

30040457

Line Transient Response

VIN = 3.6 – 4.2V, VOUT = 1.2V, 250 mA load

30040458

Line Transient Response

VIN = 3.0 – 3.6V, VOUT = 3.0V, 250 mA load

30040459

Start up into PWM Mode

VOUT = 1.2V, 1.5A load

30040460

Start up into PWM Mode

VOUT = 3.0 V, 1.5A load

30040461

www.national.com 12

LM26480

Start up into PFM Mode

VOUT = 1.2V, 30 mA load

30040462

Start up into PFM Mode

VOUT = 3.0V, 30 mA load

30040470

13 www.national.com

LM26480

DC/DC Converters

OVERVIEW

The LM26480 provides the DC/DC converters that supply the

various power needs of the application by means of two linear

low dropout regulators, LDO1 and LDO2, and two buck con-

verters, SW1 and SW2. The table here under lists the output

characteristics of the various regulators.

Supply Specification

Supply Load

Output

VOUT Range (V)

IMAX

Maximum Output

Current (mA)

LDO1 analog 1.0 to 3.5 300

LDO2 analog 1.0 to 3.5 300

SW1 digital 0.8 to 2.0 1500

SW2 digital 1.0 to 3.3 1500

LINEAR LOW DROPOUT REGULATORS (LDOs)

LDO1 and LDO2 are identical linear regulators targeting ana-

log loads characterized by low noise requirements. LDO1 and

LDO2 are enabled through the ENLDO pin.

30040404

NO-LOAD STABILITY

The LDOs will remain stable and in regulation with no external

load. This is an important consideration in some circuits, for

example, CMOS RAM keep-alive applications.

SW1, SW2: Synchronous Step-

Down Magnetic DC/DC Converters

FUNCTIONAL DESCRIPTION

The LM26480 incorporates two high-efficiency synchronous

switching buck regulators, SW1 and SW2, that deliver a con-

stant voltage from a single Li-Ion battery to the portable

system processors. Using a voltage mode architecture with

synchronous rectification, both bucks have the ability to de-

liver up to 1500 mA depending on the input voltage and output

voltage (voltage head room), and the inductor chosen (max-

imum current capability).

There are three modes of operation depending on the current

required - PWM, PFM, and shutdown. PWM mode handles

current loads of approximately 70 mA or higher, delivering

voltage precision of +/-3% with 90% efficiency or better.

Lighter output current loads cause the device to automatically

switch into PFM for reduced current consumption (IQ = 15 µA

typ.) and a longer battery life. The Standby operating mode

turns off the device, offering the lowest current consumption.

PWM or PFM mode is selected automatically or PWM mode

can be forced through the setting of the buck control register.

Both SW1 and SW2 can operate up to a 100% duty cycle

(PMOS switch always on) for low drop out control of the output

voltage. In this way the output voltage will be controlled down

to the lowest possible input voltage.

Additional features include soft-start, under-voltage lock-out,

current overload protection, and thermal overload protection.

CIRCUIT OPERATION DESCRIPTION

A buck converter contains a control block, a switching PFET

connected between input and output, a synchronous rectify-

ing NFET connected between the output and ground

(BCKGND pin) and a feedback path. During the first portion

of each switching cycle, the control block turns on the internal

PFET switch. This allows current to flow from the input

through the inductor to the output filter capacitor and load. The

inductor limits the current to a ramp with a slope of

by storing energy in a magnetic field. During the second por-

tion of each cycle, the control block turns the PFET switch off,

blocking current flow from the input, and then turns the NFET

synchronous rectifier on. The inductor draws current from

ground through the NFET to the output filter capacitor and

load, which ramps the inductor current down with a slope of

The output filter stores charge when the inductor current is

high, and releases it when low, smoothing the voltage across

the load.

SYNC FUNCTION

The LM26480 has the ability to use an external oscillator

source. The source must be 13MHz nominal and operate

within a range of 15.6 MHz and 10.4 MHz, proportionally the

same limits as the 2.0 MHz internal oscillator. The LM26480

has an internal divider which will divide the speed down by

6.5 to the nominal 2 MHz and use it for the regulators. This

SYNC function replaces the internal oscillator. When the

LM26480 is sold with this feature enabled the part will not

function without the external oscillator present. The feature

must be enabled by manufacturing before it can be used.

Contact National Sales for more information.

PWM OPERATION

During PWM operation the converter operates as a voltage-

mode controller with input voltage feed forward. This allows

the converter to achieve excellent load and line regulation.

The DC gain of the power stage is proportional to the input

voltage. To eliminate this dependence, feed forward voltage

inversely proportional to the input voltage is introduced.

INTERNAL SYNCHRONOUS RECTIFICATION

While in PWM mode, the buck uses an internal NFET as a

synchronous rectifier to reduce rectifier forward voltage drop

and associated power loss. Synchronous rectification pro-

vides a significant improvement in efficiency whenever the

output voltage is relatively low compared to the voltage drop

across an ordinary rectifier diode.

www.national.com 14

LM26480

CURRENT LIMITING

A current limit feature allows the converter to protect itself and

external components during overload conditions. PWM mode

implements current limiting using an internal comparator that

trips at 2.0A for both bucks (typ). If the output is shorted to

ground the device enters a timed current limit mode where the

NFET is turned on for a longer duration until the inductor cur-

rent falls below a low threshold, ensuring inductor current has

more time to decay, thereby preventing runaway.

PFM OPERATION

At very light loads, the converter enters PFM mode and op-

erates with reduced switching frequency and supply current

to maintain high efficiency.

The part will automatically transition into PFM mode when ei-

ther of two conditions occurs for a duration of 32 or more clock

cycles:

A. The inductor current becomes discontinuous

B. The peak PMOS switch current drops below the IMODE

level

During PFM operation, the converter positions the output volt-

age slightly higher than the nominal output voltage during

PWM operation, allowing additional headroom for voltage

drop during a load transient from light to heavy load. The PFM

comparators sense the output voltage via the feedback pin

and control the switching of the output FETs such that the

output voltage ramps between 0.8% and 1.6% (typical) above

the nominal PWM output voltage. If the output voltage is be-

low the ‘high’ PFM comparator threshold, the PMOS power

switch is turned on. It remains on until the output voltage ex-

ceeds the ‘high’ PFM threshold or the peak current exceeds

the IPFM level set for PFM mode. The typical peak current in

PFM mode is:

Once the PMOS power switch is turned off, the NMOS power

switch is turned on until the inductor current ramps to zero.

When the NMOS zero-current condition is detected, the

NMOS power switch is turned off. If the output voltage is be-

low the ‘high’ PFM comparator threshold (see following fig-

ure), the PMOS switch is again turned on and the cycle is

repeated until the output reaches the desired level. Once the

output reaches the ‘high’ PFM threshold, the NMOS switch is

turned on briefly to ramp the inductor current to zero and then

both output switches are turned off and the part enters an

extremely low power mode. Quiescent supply current during

this ‘sleep’ mode is less than 30 µA, which allows the part to

achieve high efficiencies under extremely light load condi-

tions. When the output drops below the ‘low’ PFM threshold,

the cycle repeats to restore the output voltage to ~1.6% above

the nominal PWM output voltage.

If the load current should increase during PFM mode (see

figure below) causing the output voltage to fall below the

‘low2’ PFM threshold, the part will automatically transition into

fixed-frequency PWM mode.

SW1, SW2 CONTROL

SW1 and SW2 are enabled/disabled through the external en-

able pins.

The Modulation mode PWM/PFM is by default automatic and

depends on the load as described above in the functional de-

scription. The modulation mode can be factory trimmed, forc-

ing the buck to operate in PWM mode regardless of the load

condition.

15 www.national.com

LM26480

30040405

SHUTDOWN MODE

During shutdown the PFET switch, reference, control and

bias circuitry of the converters are turned off. The NFET

switch will be on in shutdown to discharge the output. When

the converter is enabled, soft start is activated. It is recom-

mended to disable the converter during the system power up

and under voltage conditions when the supply is less than

2.8V.

SOFT START

The soft-start feature allows the power converter to gradually

reach the initial steady state operating point, thus reducing

start-up stresses and surges. The two LM26480 buck con-

verters have a soft-start circuit that limits in-rush current dur-

ing start-up. During start-up the switch current limit is

increased in steps. Soft start is activated only if EN goes from

logic low to logic high after VIN reaches 2.8V. Soft start is im-

plemented by increasing switch current limit in steps of 250

mA, 500 mA, 950 mA and 2A for both bucks (typ. switch cur-

rent limit). The start-up time thereby depends on the output

capacitor and load current demanded at start-up.

LOW DROPOUT OPERATION

The LM26480 can operate at 100% duty cycle (no switching;

PMOS switch completely on) for low dropout support of the

output voltage. In this way the output voltage will be controlled

down to the lowest possible input voltage. When the device

operates near 100% duty cycle, output voltage ripple is ap-

proximately 25 mV. The minimum input voltage needed to

support the output voltage is

VIN, MIN = ILOAD * (RDSON, PFET + RINDUCTOR) + VOUT

—ILOAD Load current

—RDSON, PFET Drain to source resistance of

PFET switch in the triode region

—RINDUCTOR Inductor resistance

FLEXIBLE POWER-ON RESET (i.e., POWER GOOD WITH

DELAY)

The LM26480 is equipped with an internal Power-On-Reset

(“POR”) circuit which monitors the output voltage levels on

bucks 1 and 2. The nPOR is an open drain logic output which

is logic LOW when either of the buck outputs are below 91%

of the rising value , or when one or both outputs fall below

82% of the desired value. The time delay between output

voltage level and nPOR is enabled is (50 µs, 50 ms, 100 ms,

200 ms), 50 ms by default. For any other delay option, other

than the default, please consult a National Sales Represen-

tative. The system designer can choose the external pull-up

resistor (i.e. 100 kΩ ) for the nPOR pin.

www.national.com 16

LM26480

NPOR with Counter Delay

30040406

The above diagram shows the simplest application of the

Power-On Reset, where both switcher enables are tied to-

gether. In Case 1, EN1 causes nPOR to transition LOW and

triggers the nPOR delay counter. If the power supply for

Buck2 does not come on within that period, nPOR will stay

LOW, indicating a power fail mode. Case 2 indicates the vice

versa scenario if Buck1 supply did not come on. In both cases

the nPOR remains LOW. Case 3 shows a typical application

of the Power-On Reset, where both switcher enables are tied

together. Even if RDY1 ramps up slightly faster than RDY2

(or vice versa), the nPOR signal will trigger a programmable

delay before going HIGH, as explained below.

17 www.national.com

LM26480

Faults Occurring in Counter Delay After Startup

30040407

The above timing diagram details the Power Good with delay

with respect to the enable signals EN1, and EN2. The RDY1,

RDY2 are internal signals derived from the output of two com-

parators. Each comparator has been trimmed as follows:

Comparator Level Buck Supply Level

HIGH Greater than 91%

LOW Less than 82%

The circuits for EN1 and RDY1 are symmetrical to EN2 and

RDY2, so each reference to EN1 and RDY1 will also work for

EN2 and RDY2 and vice versa.

If EN1 and RDY1 signals are High at time t1, then the RDY1

signal rising edge triggers the programmable delay counter

(50 μs, 50 ms, 100 ms, 200 ms). This delay forces nPOR LOW

between time interval t1 and t2. NPOR is then pulled high after

the programmable delay is completed. Now if EN2 and RDY2

are initiated during this interval the nPOR signal ignores this

event.

If either RDY1or RDY2 were to go LOW at t3 then the pro-

grammable delay is triggered again.

www.national.com 18

LM26480

NPOR Mask Window

30040408

In Case 1, we see that case where EN2 and RDY2 are initi-

ated after triggered programmable delay. To prevent the

nPOR being asserted again, a masked window (5 ms) counter

delay is triggered off the EN2 rising edge. NPOR is still held

HIGH for the duration of the mask, whereupon the nPOR sta-

tus afterwards will depend on the status of both RDY1 and

RDY2 lines.

In Case 2, we see the case where EN2 is initiated after the

RDY1 triggered programmable delay, but RDY2 never goes

HIGH (Buck2 never turns on). Normal operation operation of

nPOR occurs wilth respect to EN1 and RDY1, and the nPOR

signal is held HIGH for the duration of the mask window. We

see that nPOR goes LOW after the masking window has

timed out because it is now dependent on RDY1 and RDY2,

where RDY2 is LOW.

19 www.national.com

LM26480

Design Implementation of the Flexible Power-On Reset

30040409

Design implementation of the flexible power-on reset. An in-

ternal power-on reset of the IC is used with EN1 and EN2 to

produce a reset signal (LOW) to the delay timer nPOR. EN1

and RDY1 or EN2 and RDY2 are used to generate the set

signal (HIGH) to the delay timer. S=R=1 never occurs. The

mask timers are triggered off EN1 and EN2 which are gated

with RDY1, and RDY2 to generate outputs to the final AND

gate to generate the nPOR.

UNDER VOLTAGE LOCK OUT

The LM26480 features an “under voltage lock out circuit”. The

function of this circuit is to continuously monitor the raw input

supply voltage (VINLDO12) and automatically disables the

four voltage regulators whenever this supply voltage is less

than 2.8 VDC.

The circuit incorporates a bandgap based circuit that estab-

lishes the reference used to determine the 2.8 VDC trip point

for a VIN OK – Not OK detector. This VIN OK signal is then

used to gate the enable signals to the four regulators of the

LM26480. When VINLDO12 is greater than 2.8 VDC the four

enables control the four regulators, when VINLDO12 is less

than 2.8 VDC the four regulators are disabled by the VIN de-

tector being in the “Not OK” state. The circuit has built in

hysteresis to prevent chattering occurring.

www.national.com 20

LM26480

Application Notes

EXTERNAL COMPONENT SELECTION

30040410

Actual VOUT W/

Com/R (V)

Actual

VOUTDelta from

Target (V)

Ideal Resistor Values Common R Values Feedback Capacitors

Target

Vout (V)

R1 (KΩ) R2 (KΩ) R1 (KΩ) R2 (KΩ) C1(pF) C2(pF)

0.8 120 200 121 200 0.803 0.002 15 none Buck1

0.9 160 200 162 200 0.905 0.005 15 none Only

1 200 200 200 200 1 0 15 none ^

1.1 240 200 240 200 1.1 0 15 none |

1.2 280 200 280 200 1.2 0 12 none |

1.3 320 200 324 200 1.31 0.01 12 none Buck1

1.4 360 200 357 200 1.393 -0.008 10 none And

1.5 400 200 402 200 1.505 0.005 10 none Buck2

1.6 440 200 442 200 1.605 0.005 8.2 none |

1.7 427 178 432 178 1.713 0.013 8.2 none |

1.8 463 178 464 178 1.803 0.003 8.2 none |

1.9 498 178 499 178 1.902 0.002 8.2 none |

2 450 150 453 150 2.01 0.01 8.2 none >

2.1 480 150 475 150 2.083 -0.017 8.2 none ^

2.2 422 124 422 124 2.202 0.002 8.2 none |

2.3 446 124 442 124 2.282 -0.018 8.2 none |

2.4 471 124 475 124 2.415 0.015 8.2 none |

2.5 400 100 402 100 2.51 0.01 8.2 none |

2.6 420 100 422 100 2.61 0.01 8.2 none |

2.7 440 100 442 100 2.71 0.01 8.2 33 Buck2

2.8 460 100 464 100 2.82 0.02 8.2 33 Only

2.9 480 100 475 100 2.875 -0.025 8.2 33 |

3 500 100 499 100 2.995 -0.005 6.8 33 |

3.1 520 100 523 100 3.115 0.015 6.8 33 |

3.2 540 100 536 100 3.18 -0.02 6.8 33 |

3.3 560 100 562 100 3.31 0.01 6.8 33 |

The output voltages of the bucks of the LM26480 are estab-

lished by the feed back resistor divider R1 shown on the

application circuit above. The equation for determining V is:

VOUT = VFB (R1+R2)/R2 where VFB is the voltage on the Buck

FBx pin.

The Buck control loop will force the voltage on VFB to be 0.50

V +/- 3%.

The above table shows ideal resistor values to establish buck

voltages from 0.8V to 3.3 V along with common resistor val-

ues to establish these voltages. Common resistors do not

always produce the target value, error is given in the delta

column.

In addition to the resistor feedback, capacitor feedback C1 is

always required, and depending on the output voltage capac-

itor C2 is also required. See the application diagram below

and the above table for these requirements.

21 www.national.com

LM26480

Inductor Value Unit Description Notes

LSW1,2 2.2 µH SW1,2 inductor D.C.R. 70 mΩ

OUTPUT INDUCTORS & CAPACITORS FOR SW1 AND

SW2

There are several design considerations related to the selec-

tion of output inductors and capacitors:

•Load transient response;

•Stability;

•Efficiency;

•Output ripple voltage; and

•Over-current ruggedness.

The LM26480 has been optimized for use with nominal values

2.2 µH and 10 µF. If other values are needed for the design,

please contact National Semiconductor sales with any con-

cerns.

INDUCTOR SELECTION FOR SW1 AND SW2

A nominal inductor value of 2.2 µH is recommended. It is im-

portant to guarantee the inductor core does not saturate

during any foreseeable operational situation.

Care should be taken when reviewing the different saturation

current ratings that are specified by different manufacturers.

Saturation current ratings are typically specified at 25ºC, so

ratings at maximum ambient temperature of the application

should be requested from the manufacturer.

There are two methods to choose the inductor saturation cur-

rent rating:

Recommended method:

The best way to guarantee the inductor does not saturate is

to choose an inductor that has saturation current rating

greater than the maximum LM26480 current limit of 2.4A. In

this case the device will prevent inductor saturation.

Alternate method:

If the recommended approach cannot be used, care must be

taken to guarantee that the saturation current is greater than

the peak inductor current:

30040471

ISAT:Inductor saturation current at operating tempera-

ture

ILPEAK:Peak inductor current during worst case conditions

IOUTMAX:Maximum average inductor current

IRIPPLE:Peak-to-Peak inductor current

VOUT:Output voltage

VIN:Input voltage

L: Inductor value in Henries at IOUTMAX

F: Switching frequency, Hertz

D: Estimated duty factor

EFF: Estimated power supply efficiency

ISAT may not be exceeded during any operation, including

transients, startup, high temperature, worst case conditions,

etc.

SUGGESTED INDUCTORS AND THEIR SUPPLIERS

Model Vendor Dimension

s (mm)

DCR

(max)

ISATURATION

DO3314-22

2MX

Coilcraft 3.3 x 3.3 x

1.4

200 mΩ≈1.8A

LPO3310-2

22MX

Coilcraft 3.3 x 3.3 x 1 150 mΩ≈1.3A

ELL6PG2R

Panaso

nic

6.0 x 6.0 x

2.0

37mΩ≈2.2A

ELC6GN2R

Panaso

nic

6.0 x 6.0 x

1.5

53 mΩ≈1.9A

CDRH2D14

NP-2R2NC

Sumida 3.2 x 3.2 x

1.5

94 mΩ≈1.5A

Note: Inductor Current Saturation values are estimates; inductor

manufacturer should be contacted for guaranteed values.

OUTPUT CAPACITOR SELECTION FOR SW1 AND SW2

A ceramic output capacitor of 10 µF, 6.3V is recommended

with an ESR of about 2 mΩ or less.

Output ripple can be estimated from the vector sum of the

reactive (Capacitor) voltage component and the real (ESR)

voltage component of the output capacitor.

VCOUT:Estimated reactive output ripple

VROUT:Estimated real output ripple

VPPOUT:Estimated peak-to-peak output ripple

The output capacitor needs to be mounted as close as pos-

sible to the output pin of the device. For better temperature

performance, X7R or X5R types are recommended. DC bias

characteristics of ceramic capacitors must be considered

when selecting case sizes like 0805 and 0603.

DC bias characteristics vary from manufacturer to manufac-

turer and by case size. DC bias curves should be requested

from them as part of the capacitor selection process. ESR is

typically higher for smaller packages.

The output filter capacitor smooths out current flow from the

inductor to the load, helps maintain a steady output voltage

during transient load changes and reduces output voltage

ripple. These capacitors must be selected with sufficient ca-

pacitance and sufficiently low ESR to perform these functions.

Note that the output voltage ripple is dependent on the induc-

tor current ripple and the equivalent series resistance of the

output capacitor (ESRCOUT). ESRCOUT is frequency depen-

dent as well as temperature dependent. The RESR should be

calculated with the applicable switching frequency and ambi-

ent temperature.

www.national.com 22

LM26480

INPUT CAPACITOR SELECTION FOR SW1 AND SW2

It is required to use a ceramic input capacitor of at least 4.7

μF and 6.3V with an ESR of under 10 mΩ.

The input power source supplies average current continu-

ously. During the PFET switch on-time, however, the de-

manded di/dt is higher than can be typically supplied by the

input power source. This delta is supplied by the input capac-

itor.

A simplified “worst case” assumption is that all of the PFET

current is supplied by the input capacitor. This will result in

conservative estimates of input ripple voltage and capacitor

RMS current. Input ripple voltage is estimated as follows:

VPPIN:Estimated peak-to-peak input ripple voltage

IOUT:Output current, Amps

CIN: Input capacitor value, Farads

ESRIN: Input capacitor ESR, Ohms

This capacitor is exposed to significant RMS current, so it is

important to select a capacitor with an adequate RMS current

rating. Capacitor RMS current estimated as follows:

IRSCIN Estimated input capacitor RMS current

Model Type Vendor Voltage Rating Case Size

4.7 µF for CIN

C2012X5R0J475K Ceramic, X5R TDK 6.3V 0805, (2012)

JMK212BJ475K Ceramic, X5R Taiyo-Yuden 6.3V 0805, (2012)

GRM21BR60J475K Ceramic, X5R Murata 6.3V 0805, (2012)

C1608X5R0J475K Ceramic, X5R TDK 6.3V 0603, (1608)

10 µF for COUT

GRM21BR60J106K Ceramic, X5R Murata 6.3V 0805, (2012)

JMK212BJ106K Ceramic, X5R Taiyo-Yuden 6.3V 0805, (2012)

C2012X5R0J106K Ceramic, X5R TDK 6.3V 0805, (2012)

C1608X5R0J106K Ceramic, X5R TDK 6.3V 0603, (1608)

23 www.national.com

LM26480

FEEDBACK RESISTORS FOR LDOs

30040410

Target VOUT (V) Ideal Resistor Values Common R Values Actual VOUT W/

Com/R (V)

R1 (KΩ) R2 (KΩ) R1 (KΩ) R2 (KΩ)

1 200 200 200 200 1

1.1 240 200 240 200 1.1

1.2 280 200 280 200 1.2

1.3 320 200 324 200 1.31

1.4 360 200 357 200 1.393

1.5 400 200 402 200 1.505

1.6 440 200 442 200 1.605

1.7 480 200 562 232 1.711

1.8 520 200 604 232 1.802

1.9 560 200 562 200 1.905

2 600 200 604 200 2.01

2.1 640 200 715 221 2.118

2.2 680 200 681 200 2.203

2.3 720 200 806 226 2.283

2.4 760 200 845 221 2.412

2.5 800 200 750 187 2.505

2.6 840 200 909 215 2.614

2.7 880 200 1100 249 2.709

2.8 920 200 1150 249 2.809

2.9 960 200 1210 255 2.873

3 1000 200 1000 200 3

3.1 1040 200 1000 191 3.118

3.2 1080 200 1000 187 3.174

3.3 1120 200 1210 215 3.314

3.4 1160 200 1210 210 3.381

3.5 1200 200 1210 200 3.525

The output Voltages of the LDOs of the LM26480 are estab-

lished by the feed back resistor divider R1 shown on the

application circuit above. The equation for determining VOUT

is: VOUT = VFB(R1+R2)/R2, where Vfb is the voltage on the

LDOX_FB pin.

The LDO control loop will force the voltage on VFBo be 0.50

V +/- 3%. The above table shows ideal resistor values to es-

tablish LDO voltages from 1.0 V to 3.5 V along with common

resistor values to establish these voltages. Common resistors

do not always produce the target value, error is given in the

final column.

To keep the power consumed by the feedback network low it

is recommended that R2 be established as about 200 KΩ.

Lesser values of R2 are OK at the users discretion..

www.national.com 24

LM26480

LDO CAPACITOR SELECTION

Input Capacitor

An input capacitor is required for stability. It is recommended

that a 1.0 μF capacitor be connected between the LDO input

pin and ground (this capacitance value may be increased

without limit). This capacitor must be located a distance of not

more than 1 cm from the input pin and returned to a clean

analog ground. Any good quality ceramic, tantalum, or film

capacitor may be used at the input.

Warning: Important: Tantalum capacitors can suffer catastrophic failures

due to surge currents when connected to a low impedance

source of power (like a battery or a very large capacitor). If a

tantalum capacitor is used at the input, it must be guaranteed

by the manufacturer to have a surge current rating sufficient for

the application.

There are no requirements for the ESR on the input capacitor,

but tolerance and temperature coefficient must be considered

when selecting the capacitor to ensure the capacitance will re-

main approximately 1.0 μF over the entire operating tempera-

ture range.

Output Capacitor

The LDOs on the LM26480 are designed specifically to work

with very small ceramic output capacitors. A 1.0 μF ceramic

capacitor (temperature types Z5U, Y5V or X7R) with ESR be-

tween 5 mΩ to 500 mΩ, are suitable in the application circuit.

It is also possible to use tantalum or film capacitors at the

device output COUT (or VOUT), but these are not as attrac-

tive for reasons of size and cost. The output capacitor must

meet the requirement for the minimum value of capacitance

and also have an ESR value that is within the range 5 mΩ to

500 mΩ for stability.

Capacitor Characteristics

The LDOs are designed to work with ceramic capacitors on

the output to take advantage of the benefits they offer. For

capacitance values in the range of 0.47 μF to 4.7 μF, ceramic

capacitors are the smallest, least expensive and have the

lowest ESR values, thus making them best for eliminating

high frequency noise. The ESR of a typical 1.0 μF ceramic

capacitor is in the range of 20 mΩ to 40 mΩ, which easily

meets the ESR requirement for stability for the LDOs.

For both input and output capacitors, careful interpretation of

the capacitor specification is required to ensure correct device

operation. The capacitor value can change greatly, depend-

ing on the operating conditions and capacitor type.

In particular, the output capacitor selection should take ac-

count of all the capacitor parameters, to ensure that the

specification is met within the application. The capacitance

can vary with DC bias conditions as well as temperature and

frequency of operation. Capacitor values will also show some

decrease over time due to aging. The capacitor parameters

are also dependent on the particular case size, with smaller

sizes giving poorer performance figures in general. As an ex-

ample, the graph below shows a typical graph comparing

different capacitor case sizes in a capacitance vs. DC bias

plot.

30040416

As shown in the graph, increasing the DC bias condition can

result in the capacitance value that falls below the minimum

value given in the recommended capacitor specifications ta-

ble. Note that the graph shows the capacitance out of spec

for the 0402 case size capacitor at higher bias voltages. It is

therefore recommended that the capacitor manufacturers’

specifications for the nominal value capacitor are consulted

for all conditions, as some capacitor sizes (e.g. 0402) may not

be suitable in the actual application.

The ceramic capacitor’s capacitance can vary with tempera-

ture. The capacitor type X7R, which operates over a temper-

ature range of −55°C to +125°C, will only vary the capacitance

to within ±15%. The capacitor type X5R has a similar toler-

ance over a reduced temperature range of −55°C to +85°C.

Many large value ceramic capacitors, larger than 1 μF are

manufactured with Z5U or Y5V temperature characteristics.

Their capacitance can drop by more than 50% as the tem-

perature varies from 25°C to 85°C. Therefore X7R is recom-

mended over Z5U and Y5V in applications where the ambient

temperature will change significantly above or below 25°C.

Tantalum capacitors are less desirable than ceramic for use

as output capacitors because they are more expensive when

comparing equivalent capacitance and voltage ratings in the

0.47 μF to 4.7 μF range. Another important consideration is

that tantalum capacitors have higher ESR values than equiv-

alent size ceramics. This means that while it may be possible

to find a tantalum capacitor with an ESR value within the sta-

ble range, it would have to be larger in capacitance (which

means bigger and more costly) than a ceramic capacitor with

the same ESR value. It should also be noted that the ESR of

a typical tantalum will increase about 2:1 as the temperature

goes from 25°C down to −40°C, so some guard band must

be allowed.

Capacitor Min Value Unit Description Recommended Type

CLDO1 0.47 µF LDO1 output capacitor Ceramic, 6.3V, X5R

CLDO2 0.47 µF LDO2 output capacitor Ceramic, 6.3V, X5R

CSW1 10 µF SW1 output capacitor Ceramic, 6.3V, X5R

CSW2 10 µF SW2 output capacitor Ceramic, 6.3V, X5R

Analog Power Signal Routing

All power inputs should be tied to the main VDD source (i.e.

battery), unless the user wishes to power it from another

source. (i.e. powering LDO from Buck output).

The analog VDD inputs power the internal bias and error am-

plifiers, so they should be tied to the main VDD. The analog

VDD inputs must have an input voltage between 2.8 and 5.5

V, as specified in the Electrical Characteristics section of this

datasheet.

25 www.national.com

LM26480

The other Vins (VINLDO1, VINLDO2, VIN1, VIN2) can actu-

ally have inputs lower than 2.8V, as long as it's higher than

the programmed output (+0.3V, to be safe). The analog and

digital grounds should be tied together outside of the chip to

reduce noise coupling.

For more information on board layout techniques, refer to Ap-

plication Note AN–1187 “Leadless Lead frame Package

(LLP).” on http://www.national.com This application note also

discusses package handling, solder stencil and the assembly

process.

www.national.com 26

LM26480

Board Layout Considerations

PC board layout is an important part of DC-DC converter de-

sign. Poor board layout can disrupt the performance of a DC-

DC converter and surrounding circuitry by contributing to EMI,

ground bounce, and resistive voltage loss ii the traces. These

can send erroneous signals to the DC-DC converter IC, re-

sulting in poor regulation or instability. Poor layout can also

result in re-flow problems leading to poor solder joints, which

can result in erratic or degraded performance.

Good layout for the LM26480 bucks can be implemented by

following a few simple design rules, as illustrated in Figure 6.

30040468

FIGURE 3. Board Layout Design Rules for the LM26480

1. Place the buck inductor and filter capacitors close

together and make the trace short. The traces between

these components carry relatively high switching

currents and act as antennas. Following this rule reduces

radiated noise. Place the capacitors and inductor close

to the buck.

2. Arrange the components so that the switching current

loops curl in the same direction. During the first halt of

each cycle, current flows from the input filter capacitor,

through the buck and inductor to the output filter

capacitor and back through ground, forming a current

loop. In the second half of each cycle, current is pulled

up from ground, through the buck by the inductor, to the

output filter capacitor and then back through ground,

forming a second current loop. Routing these loops so

the current curls in the same direction prevents magnetic

field reversal between the two half-cycles and reduces

radiated noise.

3. Connect the ground pins of the buck, and filter capacitors

together using generous component-side copper fill as a

pseudo-ground plane. Then connect this to the ground-

plane (if one is used) with several vias. This reduces

ground—plane noise by preventing the switching

currents from circulating through the ground plane. it also

reduces ground bounce at the buck by giving it a low-

impedance ground connection.

4. Use wide traces between the power components and for

power connections to the DC-DC converter circuit. This

reduces voltage errors caused by resistive losses across

the traces

5. Rout noise sensitive traces, such as the voltage

feedback path, away from noisy traces between the

power components. The voltage feedback trace must

remain close to the buck circuit and should be routed

directly from FB to VOUT at the output capacitor and

should be routed opposite to noise components. This

reduces EMI radiated onto the DC-DC converter’s own

voltage feedback trace.

In mobile phones, for example, a common practice is to place

the DC-DC converter on one corner of the board, arrange the

CMOS digital circuitry around it (since this also generates

noise), and then place sensitive preamplifiers and IF stages

on the diagonally opposing corner. Often, the sensitive cir-

cuitry is shielded with a metal pan and power to it is post-

regulated to reduce conducted noise, using low-dropout

linear regulators.

27 www.national.com

LM26480

High VIN-High Load Operation

Additional inforamtion is provided when the IC is operated at

extremes of VIN and regulator loads. These are described in

terms of the junction temperature and buck output ripple man-

agement.

Junction Temperature

The maximum junction temperature TJ-MAX-OP of 125°C of the

IC package.

The following equations demonstrate junction temperature

determination, ambient temperature TA-MAX and total chip

power ust be controlled to keep TJ below this maximum:

TJ-MAX-OP = TA-MAX + (θJA) [°C/Watt] * (PD-MAX) [Watts]

Total IC power dissipation PD-MAX is the sum of the individual

power dissipation of the four regulators plus a minor amount

for chip overhead. Chip overhead is bias, TSD and LDO ana-

log.

PD-MAX = PLOD1 + PLDO2 +PBUCK1 + PBUCK2 + (0.0001A

* VIN) [Watts].

Power dissipation of LDO1 (PLDO1) = (VINLDO1 − VOUTL-

DO1) * IOUTLDO1 [V*A]

Power dissipation of LDO2 (PLDO2) = (VINLDO2 − VOUTL-

DO2) * IOUTLDO2 [V*A]

Power dissipation of Buck1 (PBuck1) = POUT − PIN = VOUT-

BUCK1 − IOUTBUCK1 * (1 − η2)/ η2 [V*A]

η1 = efficiency of Buck1

Power dissipation of Buck2 (PBuck2) = POUT − PIN = VOUT-

BUCK2 − IOUTBUCK2 * (1 − η2)/ η2 [V*A]

η2 = efficiency of Buck2

Where η is the efficiency for the specific condition is taken

from efficiency graphs.

If VIN and ILOADincrease, the output ripple associated with

the Buck Regulators also increases. This mainly occurs with

VIN > 5.2V and a load current greater than 1.20A. To ensure

operation in this area of operation, it is recommended that the

system designer circumvents the output ripple issues by in-

stalling Schottky diodes on the bucks(s) that are expected to

perform under these extreme conditions.

www.national.com 28

LM26480

Physical Dimensions inches (millimeters) unless otherwise noted

4 X 4 X 0.8 mm 24-Pin LLP Package

NS Package SQA24A

For ordering, refer to Ordering Information table

29 www.national.com

LM26480

Notes

LM26480 Externally Programmable Dual High-Current Step-Down DC/DC and Dual Linear

Regulators

For more National Semiconductor product information and proven design tools, visit the following Web sites at:

Products Design Support

Amplifiers www.national.com/amplifiers WEBENCH www.national.com/webench

Audio www.national.com/audio Analog University www.national.com/AU

Clock Conditioners www.national.com/timing App Notes www.national.com/appnotes

Data Converters www.national.com/adc Distributors www.national.com/contacts

Displays www.national.com/displays Green Compliance www.national.com/quality/green

Ethernet www.national.com/ethernet Packaging www.national.com/packaging

Interface www.national.com/interface Quality and Reliability www.national.com/quality

LVDS www.national.com/lvds Reference Designs www.national.com/refdesigns

Power Management www.national.com/power Feedback www.national.com/feedback

Switching Regulators www.national.com/switchers

LDOs www.national.com/ldo

LED Lighting www.national.com/led

PowerWise www.national.com/powerwise

Serial Digital Interface (SDI) www.national.com/sdi

Temperature Sensors www.national.com/tempsensors

Wireless (PLL/VCO) www.national.com/wireless

THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION

(“NATIONAL”) PRODUCTS. NATIONAL MAKES NO REPRESENTATIONS OR WARRANTIES WITH RESPECT TO THE ACCURACY

OR COMPLETENESS OF THE CONTENTS OF THIS PUBLICATION AND RESERVES THE RIGHT TO MAKE CHANGES TO

SPECIFICATIONS AND PRODUCT DESCRIPTIONS AT ANY TIME WITHOUT NOTICE. NO LICENSE, WHETHER EXPRESS,

IMPLIED, ARISING BY ESTOPPEL OR OTHERWISE, TO ANY INTELLECTUAL PROPERTY RIGHTS IS GRANTED BY THIS

DOCUMENT.

TESTING AND OTHER QUALITY CONTROLS ARE USED TO THE EXTENT NATIONAL DEEMS NECESSARY TO SUPPORT

NATIONAL’S PRODUCT WARRANTY. EXCEPT WHERE MANDATED BY GOVERNMENT REQUIREMENTS, TESTING OF ALL

PARAMETERS OF EACH PRODUCT IS NOT NECESSARILY PERFORMED. NATIONAL ASSUMES NO LIABILITY FOR

APPLICATIONS ASSISTANCE OR BUYER PRODUCT DESIGN. BUYERS ARE RESPONSIBLE FOR THEIR PRODUCTS AND

APPLICATIONS USING NATIONAL COMPONENTS. PRIOR TO USING OR DISTRIBUTING ANY PRODUCTS THAT INCLUDE

NATIONAL COMPONENTS, BUYERS SHOULD PROVIDE ADEQUATE DESIGN, TESTING AND OPERATING SAFEGUARDS.

EXCEPT AS PROVIDED IN NATIONAL’S TERMS AND CONDITIONS OF SALE FOR SUCH PRODUCTS, NATIONAL ASSUMES NO

LIABILITY WHATSOEVER, AND NATIONAL DISCLAIMS ANY EXPRESS OR IMPLIED WARRANTY RELATING TO THE SALE

AND/OR USE OF NATIONAL PRODUCTS INCLUDING LIABILITY OR WARRANTIES RELATING TO FITNESS FOR A PARTICULAR

PURPOSE, MERCHANTABILITY, OR INFRINGEMENT OF ANY PATENT, COPYRIGHT OR OTHER INTELLECTUAL PROPERTY

RIGHT.

LIFE SUPPORT POLICY

NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR

SYSTEMS WITHOUT THE EXPRESS PRIOR WRITTEN APPROVAL OF THE CHIEF EXECUTIVE OFFICER AND GENERAL

COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:

Life support devices or systems are devices which (a) are intended for surgical implant into the body, or (b) support or sustain life and

whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected

to result in a significant injury to the user. A critical component is any component in a life support device or system whose failure to perform

can be reasonably expected to cause the failure of the life support device or system or to affect its safety or effectiveness.

National Semiconductor and the National Semiconductor logo are registered trademarks of National Semiconductor Corporation. All other

brand or product names may be trademarks or registered trademarks of their respective holders.

For the most current product information visit us at www.national.com

National Semiconductor

Americas Technical

Support Center

Email: support@nsc.com

Tel: 1-800-272-9959

National Semiconductor Europe

Technical Support Center

Email: europe.support@nsc.com

German Tel: +49 (0) 180 5010 771

English Tel: +44 (0) 870 850 4288

National Semiconductor Asia

Pacific Technical Support Center

Email: ap.support@nsc.com

National Semiconductor Japan

Technical Support Center

Email: jpn.feedback@nsc.com