LM4960SQ/NOPB - NATIONAL SEMICONDUCTOR

LM4960

LM4960 Piezoelectric Speaker Driver

Literature Number: SNAS221A

LM4960

Piezoelectric Speaker Driver

General Description

The LM4960 utilizes a switching regulator to drive a dual

audio power amplifier. It delivers 24V

P-P

mono-BTL to a

ceramic speaker with less than 1.0% THD+N while operating

on a 3.0V power supply.

The LM4960’s switching regulator is a current-mode boost

converter operating at a fixed frequency of 1.6MHz.

Boomer audio power amplifiers were designed specifically to

provide high quality output power with a minimal amount of

external components. The LM4960 does not require output

coupling capacitors or bootstrap capacitors, and therefore is

ideally suited for mobile phone and other low voltage appli-

cations where minimal power consumption is a primary re-

quirement.

The LM4960 features a low-power consumption externally

controlled micropower shutdown mode. Additionally, the

LM4960 features and internal thermal shutdown protection

mechanism along with a short circuit protection.

The LM4960 is unity-gain stable and can be configured by

external gain-setting resistors.

Key Specifications

OUT

= 3.0 THD+N ≤1% 24V

P-P

(typ)

nPower supply range 3.0 to 7V

nSwitching Frequency 1.6MHz (typ)

Features

nLow current shutdown mode

n"Click and pop" suppression circuitry

nLow Quiescent current

nUnity-gain stable audio amplifiers

nExternal gain configuration capability

nThermal shutdown protection circuitry

nWide input voltage range (3.0V - 7V)

n1.6MHz switching frequency

Applications

nMobile phone

nPDA’s

Connection Diagram

LM4960SQ

20076582

Top View

Order Number LM4960SQ

See NS Package Number

Boomer®is a registered trademark of National Semiconductor Corporation.

April 2005

LM4960 Piezoelectric Speaker Driver

Typical Application

20076581

FIGURE 1. Typical Audio Amplifier Application Circuit

LM4960

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Absolute Maximum Ratings (Notes 1, 2)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Supply Voltage (V

) 8.5V

Supply Voltage (V1)

(Pin 27 referred to GND) 18V

Storage Temperature −65˚C to +150˚C

Input Voltage −0.3V to V

+ 0.3V

Power Dissipation (Note 3) Internally limited

ESD Susceptibility (Note 4) 2000V

ESD Susceptibility (Note 5) 200V

Junction Temperature 150˚C

Thermal Resistance

(LLP) ˚C/W

See AN-1187 ’Leadless Leadframe Packaging (LLP).’

Operating Ratings

Temperature Range

MIN

≤T

MAX

−40˚C ≤T

≤+85˚C

Supply Voltage (V

) 3.0V ≤V

≤7V

Supply Voltage (V1) 9.6V ≤V

≤16V

Electrical Characteristics V

= 3.0V (Notes 1, 2)

The following specifications apply for V

= 3V, A

= 10, R

= 800nF+20Ω, V1 = 12V unless otherwise specified. Limits apply

for T

= 25˚C.

Symbol Parameter Conditions LM4960 Units

(Limits)

Typical

(Note 6)

Limit

(Notes 7, 8)

Quiescent Power Supply Current V

= GND, No Load 85 150 mA (max)

Shutdown Current V

SHUTDOWN

= GND (Note 9) 30 100 µA (max)

Output Offset Voltage 5 40 mV (max)

SDIH

Shutdown Voltage Input High 2 V (max)

SDIL

Shutdown Voltage Input Low 0.4 V (min)

Wake-up Time C

= 0.22µF 50 ms

TSD Thermal Shutdown Temperature 170 150

190

˚C (min)

˚C (max)

Output Voltage THD = 1% (max); f = 1kHz 24 20 V

P-P

(min)

THD+N Total Harmomic Distortion + Noise V

= 3Wrms; f = 1kHz 0.04 %

Output Noise A-Weighted Filter, V

=0V 90 µV

PSRR Power Supply Rejection Ratio V

RIPPLE

= 200mV

p-p

, f = 1kHz 55 50 dB (min)

Feedback Pin Reference Voltage 1.23 V (max)

Electrical Characteristics V

= 5.0V (Notes 1, 2)

The following specifications apply for V

= 5V, A

= 10, R

= 800nF+20Ωunless otherwise specified. Limits apply for T

25˚C.

Symbol Parameter Conditions LM4960 Units

(Limits)

Typical

(Note 6)

Limit

(Notes 7, 8)

Quiescent Power Supply Current V

= GND, No Load 45 mA (max)

Shutdown Current V

SHUTDOWN

= GND (Note 9) 55 100 µA (max)

SDIH

Shutdown Voltage Input High 2 V (max)

SDIL

Shutdown Voltage Input Low 0.4 V (min)

Wake-up Time C

= 0.22µF 50 s

TSD Thermal Shutdown Temperature 170 150

190

˚C (min)

˚C (max)

Output Voltage THD = 1% (max); f = 1kHz

= Ceramic Speaker 24 20 V

P-P

(min)

THD+N Total Harmomic Distortion + Noise V

= 3Wrms; f = 1kHz 0.04 %

Output Noise A-Weighted Filter, V

=0V 90 µV

PSRR Power Supply Rejection Ratio V

RIPPLE

= 200mV

p-p

, f = 1kHz 60 dB (min)

Feedback Pin Reference Voltage 1.23 V (max)

Note 1: All voltages are measured with respect to the GND pin, unless otherwise specified.

LM4960

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Electrical Characteristics V

= 5.0V (Notes 1, 2) (Continued)

Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which

guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit

is given, however, the typical value is a good indication of device performance.

Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX,θJA, and the ambient temperature, TA. The maximum

allowable power dissipation is PDMAX =(T

JMAX −T

A)/θJA or the given in Absolute Maximum Ratings, whichever is lower. For the LM4960 typical application (shown

in Figure 1) with VDD = 12V, RL=4Ωstereo operation the total power dissipation is 3.65W. θJA = 35˚C/W.

Note 4: Human body model, 100pF discharged through a 1.5kΩresistor.

Note 5: Machine Model, 220pF–240pF discharged through all pins.

Note 6: Typicals are measured at 25˚C and represent the parametric norm.

Note 7: Limits are guaranteed to National’s AOQL (Average Outgoing Quality Level).

Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.

Note 9: Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to VDD for minimum shutdown

current.

LM4960

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Typical Performance Characteristics

THD+N vs Frequency

= 3V, V

= 9.6V, V

= 3Vrms

THD+N vs Frequency

= 3V, V

= 12V, V

= 3Vrms

20076514 20076515

THD+N vs Frequency

= 3V, V

= 15V, V

= 3Vrms

THD+N vs Frequency

= 5V, V

= 9.6V, V

= 3Vrms

20076516 20076517

THD+N vs Frequency

= 5V, V

=12V, V

= 3Vrms

THD+N vs Frequency

= 5V, V

=15V, V

= 3Vrms

20076518 20076519

LM4960

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Typical Performance Characteristics (Continued)

THD+N vs Output Power

= 3V, V

= 9.6V,

f = 100Hz, 1kHz, 10kHz

THD+N vs Output Power

= 3V, V

= 12V,

f = 100Hz, 1kHz, 10kHz

20076520 20076521

THD+N vs Output Power

= 3V, V

= 15V,

f = 100Hz, 1kHz, 10kHz

THD+N vs Output Power

= 5V, V

= 9.6V,

f = 100Hz, 1kHz, 10kHz

20076522 20076523

THD+N vs Output Power

= 5V, V

= 12V,

f = 100Hz, 1kHz, 10kHz

THD+N vs Output Power

= 5V, V

= 15V,

f = 100Hz, 1kHz, 10kHz

20076524 20076525

LM4960

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Typical Performance Characteristics (Continued)

Power Dissipation vs Output Voltage

= 3V, from top to bottom:

= 15V, V

= 12V, V

= 9.6V

Power Dissipation vs Output Voltage

= 5V, from top to bottom:

= 15V, V

= 12V, V

= 9.6V

20076509 20076510

Supply Current vs Supply Voltage

from top to bottom:

= 15V, V

= 12V, V

= 9.6V

Power Supply Rejection Ratio

=3V

20076513 20076511

Power Supply Rejection Ratio

=5V

20076512

LM4960

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Application Information

BRIDGE CONFIGURATION EXPLANATION

The Audio Amplifier portion of the LM4960 has two internal

amplifiers allowing different amplifier configurations. The first

amplifier’s gain is externally configurable, whereas the sec-

ond amplifier is internally fixed in a unity-gain, inverting

configuration. The closed-loop gain of the first amplifier is set

by selecting the ratio of Rf to Ri while the second amplifier’s

gain is fixed by the two internal 20kΩresistors. Figure 1

shows that the output of amplifier one serves as the input to

amplifier two. This results in both amplifiers producing sig-

nals identical in magnitude, but out of phase by 180˚. Con-

sequently, the differential gain for the Audio Amplifier is

= 2 *(Rf/Ri)

By driving the load differentially through outputs Vo1 and

Vo2, an amplifier configuration commonly referred to as

“bridged mode” is established. Bridged mode operation is

different from the classic single-ended amplifier configura-

tion where one side of the load is connected to ground.

A bridge amplifier design has a few distinct advantages over

the single-ended configuration. It provides differential drive

to the load, thus doubling the output swing for a specified

supply voltage. Four times the output power is possible as

compared to a single-ended amplifier under the same con-

ditions. This increase in attainable output power assumes

that the amplifier is not current limited or clipped. In order to

choose an amplifier’s closed-loop gain without causing ex-

cessive clipping, please refer to the Audio Power Amplifier

Design section.

The bridge configuration also creates a second advantage

over single-ended amplifiers. Since the differential outputs,

Vo1 and Vo2, are biased at half-supply, no net DC voltage

exists across the load. This eliminates the need for an output

coupling capacitor which is required in a single supply,

single-ended amplifier configuration. Without an output cou-

pling capacitor, the half-supply bias across the load would

result in both increased internal IC power dissipation and

also possible loudspeaker damage.

AMPLIFIER POWER DISSIPATION

Power dissipation is a major concern when designing a

successful amplifier, whether the amplifier is bridged or

single-ended. A direct consequence of the increased power

delivered to the load by a bridge amplifier is an increase in

internal power dissipation. Since the amplifier portion of the

LM4960 has two operational amplifiers, the maximum inter-

nal power dissipation is 4 times that of a single-ended am-

plifier. The maximum power dissipation for a given BTL

application can be derived from Equation 1.

DMAX(AMP)

= 4(V

)

/(2π

) (1)

where

1+R

2 +1/2πfc

BOOST CONVERTER POWER DISSIPATION

At higher duty cycles, the increased ON-time of the switch

FET means the maximum output current will be determined

by power dissipation within the LM2731 FET switch. The

switch power dissipation from ON-time conduction is calcu-

lated by Equation 2.

DMAX(SWITCH)

=DCxI

IND

(AVE)

(ON) (2)

where DC is the duty cycle.

There will be some switching losses as well, so some derat-

ing needs to be applied when calculating IC power dissipa-

tion.

TOTAL POWER DISSIPATION

The total power dissipation for the LM4960 can be calculated

by adding Equation 1 and Equation 2 together to establish

Equation 3:

DMAX(TOTAL)

= [4*(V

)

/2π

]+[DCxI

IND

(AVE)

D-

(ON)] (3)

The result from Equation 3 must not be greater than the

power dissipation that results from Equation 4:

DMAX

=(T

JMAX

-T

)/θJA (4)

For the LQA28A, θ

= 59˚C/W. T

JMAX

= 125˚C for the

LM4960. Depending on the ambient temperature, T

,ofthe

system surroundings, Equation 4 can be used to find the

maximum internal power dissipation supported by the IC

packaging. If the result of Equation 3 is greater than that of

Equation 4, then either the supply voltage must be in-

creased, the load impedance increased or T

reduced. For

the typical application of a 3V power supply, with V1 set to

12V and a 800nF + 20Ωload, the maximum ambient tem-

perature possible without violating the maximum junction

temperature is approximately 118˚C provided that device

operation is around the maximum power dissipation point.

Thus, for typical applications, power dissipation is not an

issue. Power dissipation is a function of output power and

thus, if typical operation is not around the maximum power

dissipation point, the ambient temperature may be increased

accordingly. Refer to the Typical Performance Characteris-

tics curves for power dissipation information for lower output

levels.

EXPOSED-DAP PACKAGE PCB MOUNTING

CONSIDERATIONS

The LM4960’s exposed-DAP (die attach paddle) package

(LD) provides a low thermal resistance between the die and

the PCB to which the part is mounted and soldered. The low

thermal resistance allows rapid heat transfer from the die to

the surrounding PCB copper traces, ground plane, and sur-

rounding air. The LD package should have its DAP soldered

to a copper pad on the PCB. The DAP’s PCB copper pad

may be connected to a large plane of continuous unbroken

copper. This plane forms a thermal mass, heat sink, and

radiation area. Further detailed and specific information con-

cerning PCB layout, fabrication, and mounting an LD (LLP)

package is found in National Semiconductor’s Package En-

gineering Group under application note AN1187.

LM4960

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Application Information (Continued)

SHUTDOWN FUNCTION

In many applications, a microcontroller or microprocessor

output is used to control the shutdown circuitry to provide a

quick, smooth transition into shutdown. Another solution is to

use a single-pole, single-throw switch, and a pull-up resistor.

One terminal of the switch is connected to GND. The other

side is connected to the two shutdown pins and the terminal

of the pull-up resistor. The remaining resistance terminal is

connected to V

. If the switch is open, then the external

pull-up resistor connected to V

will enable the LM4960.

This scheme guarantees that the shutdown pins will not float

thus preventing unwanted state changes.

PROPER SELECTION OF EXTERNAL COMPONENTS

Proper selection of external components in applications us-

ing integrated power amplifiers, and switching DC-DC con-

verters, is critical for optimizing device and system perfor-

mance. Consideration to component values must be used to

maximize overall system quality.

The best capacitors for use with the switching converter

portion of the LM4960 are multi-layer ceramic capacitors.

They have the lowest ESR (equivalent series resistance)

and highest resonance frequency, which makes them opti-

mum for high frequency switching converters.

When selecting a ceramic capacitor, only X5R and X7R

dielectric types should be used. Other types such as Z5U

and Y5F have such severe loss of capacitance due to effects

of temperature variation and applied voltage, they may pro-

vide as little as 20% of rated capacitance in many typical

applications. Always consult capacitor manufacturer’s data

curves before selecting a capacitor. High-quality ceramic

capacitors can be obtained from Taiyo-Yuden, AVX, and

Murata.

POWER SUPPLY BYPASSING

As with any amplifier, proper supply bypassing is critical for

low noise performance and high power supply rejection. The

capacitor location on both V1 and V

pins should be as

close to the device as possible.

SELECTING INPUT CAPACITOR FOR AUDIO

AMPLIFIER

One of the major considerations is the closedloop bandwidth

of the amplifier. To a large extent, the bandwidth is dictated

by the choice of external components shown in Figure 1. The

input coupling capacitor, C

, forms a first order high pass filter

which limits low frequency response. This value should be

chosen based on needed frequency response for a few

distinct reasons.

High value input capacitors are both expensive and space

hungry in portable designs. Clearly, a certain value capacitor

is needed to couple in low frequencies without severe at-

tenuation. But ceramic speakers used in portable systems,

whether internal or external, have little ability to reproduce

signals below 100Hz to 150Hz. Thus, using a high value

input capacitor may not increase actual system perfor-

mance.

In addition to system cost and size, click and pop perfor-

mance is affected by the value of the input coupling capaci-

tor, C

. A high value input coupling capacitor requires more

charge to reach its quiescent DC voltage (nominally 1/2

). This charge comes from the output via the feedback

and is apt to create pops upon device enable. Thus, by

minimizing the capacitor value based on desired low fre-

quency response, turn-on pops can be minimized.

SELECTING BYPASS CAPACITOR FOR AUDIO

AMPLIFIER

Besides minimizing the input capacitor value, careful consid-

eration should be paid to the bypass capacitor value. Bypass

capacitor, C

, is the most critical component to minimize

turn-on pops since it determines how fast the amplifer turns

on. The slower the amplifier’s outputs ramp to their quies-

cent DC voltage (nominally 1/2 V

), the smaller the turn-on

pop. Choosing C

equal to 1.0µF along with a small value of

(in the range of 0.039µF to 0.39µF), should produce a

virtually clickless and popless shutdown function. Although

the device will function properly, (no oscillations or motor-

boating), with C

equal to 0.1µF, the device will be much

more susceptible to turn-on clicks and pops. Thus, a value of

equal to 1.0µF is recommended in all but the most cost

sensitive designs.

SELECTING FEEDBACK CAPACITOR FOR AUDIO

AMPLIFIER

The LM4960 is unity-gain stable which gives the designer

maximum system flexability. However, to drive ceramic

speakers, a typical application requires a closed-loop differ-

ential gain of 10. In this case a feedback capacitor (C

2) will

be needed as shown in Figure 2 to bandwidth limit the

amplifier.

This feedback capacitor creates a low pass filter that elimi-

nates possible high frequency oscillations. Care should be

taken when calculating the -3dB frequency because an in-

correct combination of R

and C

2 will cause rolloff before the

desired frequency

SELECTING OUTPUT CAPACITOR (C

) FOR BOOST

CONVERTER

A single 4.7µF to 10µF ceramic capacitor will provide suffi-

cient output capacitance for most applications. If larger

amounts of capacitance are desired for improved line sup-

port and transient response, tantalum capacitors can be

used. Aluminum electrolytics with ultra low ESR such as

Sanyo Oscon can be used, but are usually prohibitively

expensive. Typical AI electrolytic capacitors are not suitable

for switching frequencies above 500 kHz because of signifi-

cant ringing and temperature rise due to self-heating from

ripple current. An output capacitor with excessive ESR can

also reduce phase margin and cause instability.

In general, if electrolytics are used, we recommended that

they be paralleled with ceramic capacitors to reduce ringing,

switching losses, and output voltage ripple.

SELECTING INPUT CAPACITOR (Cs1) FOR BOOST

CONVERTER

An input capacitor is required to serve as an energy reservoir

for the current which must flow into the coil each time the

switch turns ON. This capacitor must have extremely low

ESR, so ceramic is the best choice. We recommend a

nominal value of 4.7µF, but larger values can be used. Since

this capacitor reduces the amount of voltage ripple seen at

the input pin, it also reduces the amount of EMI passed back

along that line to other circuitry.

LM4960

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Application Information (Continued)

SETTING THE OUTPUT VOLTAGE (V

) OF BOOST

CONVERTER

The output voltage is set using the external resistors R1 and

R2 (see Figure 1). A value of approximately 13.3kΩis rec-

ommended for R2 to establish a divider current of approxi-

mately 92µA. R1 is calculated using the formula:

R1=R2X(V

/1.23 − 1) (5)

FEED-FORWARD COMPENSATION FOR BOOST

CONVERTER

Although the LM4960’s internal Boost converter is internally

compensated, the external feed-forward capacitor C

is re-

quired for stability (see Figure 1). Adding this capacitor puts

a zero in the loop response of the converter. The recom-

mended frequency for the zero fz should be approximately

6kHz. C

1 can be calculated using the formula:

1=1/(2XR1Xfz) (6)

SELECTING DIODES

The external diode used in Figure 1 should be a Schottky

diode. A 20V diode such as the MBR0520 is recommended.

The MBR05XX series of diodes are designed to handle a

maximum average current of 0.5A. For applications exceed-

ing 0.5A average but less than 1A, a Microsemi UPS5817

can be used.

DUTY CYCLE

The maximum duty cycle of the boost converter determines

the maximum boost ratio of output-to-input voltage that the

converter can attain in continuous mode of operation. The

duty cycle for a given boost application is defined as:

Duty Cycle = V

OUT

DIODE

-V

OUT

DIODE

-V

This applies for continuous mode operation.

INDUCTANCE VALUE

The first question we are usually asked is: “How small can I

make the inductor.” (because they are the largest sized

component and usually the most costly). The answer is not

simple and involves trade-offs in performance. Larger induc-

tors mean less inductor ripple current, which typically means

less output voltage ripple (for a given size of output capaci-

tor). Larger inductors also mean more load power can be

delivered because the energy stored during each switching

cycle is:

E = L/2 X (lp)2

Where “lp” is the peak inductor current. An important point to

observe is that the LM4960 will limit its switch current based

on peak current. This means that since lp(max) is fixed,

increasing L will increase the maximum amount of power

available to the load. Conversely, using too little inductance

may limit the amount of load current which can be drawn

from the output.

Best performance is usually obtained when the converter is

operated in “continuous” mode at the load current range of

interest, typically giving better load regulation and less out-

put ripple. Continuous operation is defined as not allowing

the inductor current to drop to zero during the cycle. It should

be noted that all boost converters shift over to discontinuous

operation as the output load is reduced far enough, but a

larger inductor stays “continuous” over a wider load current

range.

To better understand these trade-offs, a typical application

circuit (5V to 12V boost with a 10µH inductor) will be ana-

lyzed. We will assume:

=5V,V

OUT

= 12V, V

DIODE

= 0.5V, V

= 0.5V

Since the frequency is 1.6MHz (nominal), the period is ap-

proximately 0.625µs. The duty cycle will be 62.5%, which

means the ON-time of the switch is 0.390µs. It should be

noted that when the switch is ON, the voltage across the

inductor is approximately 4.5V. Using the equation:

V = L (di/dt)

We can then calculate the di/dt rate of the inductor which is

found to be 0.45 A/µs during the ON-time. Using these facts,

we can then show what the inductor current will look like

during operation:

During the 0.390µs ON-time, the inductor current ramps up

0.176A and ramps down an equal amount during the OFF-

time. This is defined as the inductor “ripple current”. It can

also be seen that if the load current drops to about 33mA,

the inductor current will begin touching the zero axis which

means it will be in discontinuous mode. A similar analysis

can be performed on any boost converter, to make sure the

ripple current is reasonable and continuous operation will be

maintained at the typical load current values.

MAXIMUM SWITCH CURRENT

The maximum FET switch current available before the cur-

rent limiter cuts in is dependent on duty cycle of the appli-

cation. This is illustrated in a graph in the typical perfor-

mance characterization section which shows typical values

of switch current as a function of effective (actual) duty cycle.

20076583

FIGURE 2. 10µH Inductor Current

5V - 12V Boost (LM4960)

LM4960

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Application Information (Continued)

CALCULATING OUTPUT CURRENT OF BOOST

CONVERTER (I

AMP

)

As shown in Figure 2 which depicts inductor current, the load

current is related to the average inductor current by the

relation:

LOAD

IND

(AVG) x (1 - DC) (7)

Where "DC" is the duty cycle of the application. The switch

current can be found by:

IND

(AVG) + 1/2 (I

RIPPLE

) (8)

Inductor ripple current is dependent on inductance, duty

cycle, input voltage and frequency:

RIPPLE

=DCx(V

-V

)/(fxL) (9)

combining all terms, we can develop an expression which

allows the maximum available load current to be calculated:

LOAD

(max) = (1–DC)x(I

(max)–DC(V

-V

))/fL (10)

The equation shown to calculate maximum load current

takes into account the losses in the inductor or turn-OFF

switching losses of the FET and diode.

DESIGN PARAMETERS V

AND I

The value of the FET "ON" voltage (referred to as V

equations 7 thru 10) is dependent on load current. A good

approximation can be obtained by multiplying the "ON Re-

sistance" of the FET times the average inductor current.

FET on resistance increases at V

values below 5V, since

the internal N-FET has less gate voltage in this input voltage

range (see Typical Performance Characteristics curves).

Above V

= 5V, the FET gate voltage is internally clamped

to 5V.

The maximum peak switch current the device can deliver is

dependent on duty cycle. For higher duty cycles, see Typical

Performance Characteristics curves.

INDUCTOR SUPPLIERS

Recommended suppliers of inductors for the LM4960 in-

clude, but are not limited to Taiyo-Yuden, Sumida, Coilcraft,

Panasonic, TDK and Murata. When selecting an inductor,

make certain that the continuous current rating is high

enough to avoid saturation at peak currents. A suitable core

type must be used to minimize core (switching) losses, and

wire power losses must be considered when selecting the

current rating.

PCB LAYOUT GUIDELINES

High frequency boost converters require very careful layout

of components in order to get stable operation and low

noise. All components must be as close as possible to the

LM4802 device. It is recommended that a 4-layer PCB be

used so that internal ground planes are available.

Some additional guidelines to be observed:

1. Keep the path between L1, D1, and Co extremely short.

Parasitic trace inductance in series with D1 and Co will

increase noise and ringing.

2. The feedback components R1, R2 and C

1 must be kept

close to the FB pin of U1 to prevent noise injection on the FB

pin trace.

3. If internal ground planes are available (recommended)

use vias to connect directly to ground at pin 2 of U1, as well

as the negative sides of capacitors C

1 and Co.

GENERAL MIXED-SIGNAL LAYOUT

RECOMMENDATION

This section provides practical guidelines for mixed signal

PCB layout that involves various digital/analog power and

ground traces. Designers should note that these are only

"rule-of-thumb" recommendations and the actual results will

depend heavily on the final layout.

Power and Ground Circuits

For 2 layer mixed signal design, it is important to isolate the

digital power and ground trace paths from the analog power

and ground trace paths. Star trace routing techniques (bring-

ing individual traces back to a central point rather than daisy

chaining traces together in a serial manner) can have a

major impact on low level signal performance. Star trace

routing refers to using individual traces to feed power and

ground to each circuit or even device. This technique will

take require a greater amount of design time but will not

increase the final price of the board. The only extra parts

required may be some jumpers.

Single-Point Power / Ground Connection

The analog power traces should be connected to the digital

traces through a single point (link). A "Pi-filter" can be helpful

in minimizing high frequency noise coupling between the

analog and digital sections. It is further recommended to

place digital and analog power traces over the correspond-

ing digital and analog ground traces to minimize noise cou-

pling.

Placement of Digital and Analog Components

All digital components and high-speed digital signals traces

should be located as far away as possible from analog

components and circuit traces.

Avoiding Typical Design / Layout Problems

Avoid ground loops or running digital and analog traces

parallel to each other (side-by-side) on the same PCB layer.

When traces must cross over each other do it at 90 degrees.

Running digital and analog traces at 90 degrees to each

other from the top to the bottom side as much as possible will

minimize capacitive noise coupling and crosstalk.

LM4960

www.national.com11

Physical Dimensions inches (millimeters) unless otherwise noted

LLP, Plastic, Quad

Order Number LM4960SQ

NS Package Number SQA28A

National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves

the right at any time without notice to change said circuitry and specifications.

For the most current product information visit us at www.national.com.

LIFE SUPPORT POLICY

NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS

WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR

CORPORATION. As used herein:

1. Life support devices or systems are devices or systems

which, (a) are intended for surgical implant into the body, or

(b) support or sustain life, and whose failure to perform when

properly used in accordance with instructions for use

provided in the labeling, can be reasonably expected to result

in a significant injury to the user.

2. A critical component is any component of a life support

device or system whose failure to perform can be reasonably

expected to cause the failure of the life support device or

system, or to affect its safety or effectiveness.

BANNED SUBSTANCE COMPLIANCE

National Semiconductor manufactures products and uses packing materials that meet the provisions of the Customer Products

Stewardship Specification (CSP-9-111C2) and the Banned Substances and Materials of Interest Specification (CSP-9-111S2) and contain

no ‘‘Banned Substances’’ as defined in CSP-9-111S2.

National Semiconductor

Americas Customer

Support Center

Email: new.feedback@nsc.com

Tel: 1-800-272-9959

National Semiconductor

Europe Customer Support Center

Fax: +49 (0) 180-530 85 86

Email: europe.support@nsc.com

Deutsch Tel: +49 (0) 69 9508 6208

English Tel: +44 (0) 870 24 0 2171

Français Tel: +33 (0) 1 41 91 8790

National Semiconductor

Asia Pacific Customer

Support Center

Email: ap.support@nsc.com

National Semiconductor

Japan Customer Support Center

Fax: 81-3-5639-7507

Email: jpn.feedback@nsc.com

Tel: 81-3-5639-7560

www.national.com

LM4960 Piezoelectric Speaker Driver

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