LM34923 www.ti.com SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 80-V 600-mA Constant On-Time Buck Switching Regulator Check for Samples: LM34923 FEATURES DESCRIPTION * * * * * * The LM34923 Step Down Switching Regulator features all of the functions needed to implement a low cost, efficient Buck bias regulator. This high voltage regulator contains an 80V N-Channel MOSFET Switch and a startup regulator. The device is easy to implement and is provided in an 10-pin VSSOP package. The regulator's control scheme uses an on-time inversely proportional to VIN. This feature results in the operating frequency remaining relatively constant with line and load variations. The control scheme requires no loop compensation, resulting in fast transient response. An intelligent current limit is implemented with a forced off-time which is inversely proportional to VOUT. This scheme ensures short circuit control while providing minimum foldback. Other features include: Thermal Shutdown, VCC Under Voltage Lock-out, Max Duty Cycle Limiter, a Pre-charge Switch, and a programmable Under Voltage Detector with a status flag output. 1 2 * * * * * * * Operating Input Voltage Range: 6V to 75V Integrated 80V, N-Channel Buck Switch Internal Start-up Regulator No Loop Compensation Required Ultra-Fast Transient Response Operating Frequency Remains Constant with Line and Load Variations Adjustable Output voltage From 2.5V Precision Internal Reference, 2.5% Intelligent Current Limit Reduces Foldback Programmable Input UV Detector with Status Flag Output Pre-charge Switch Enables Bootstrap Gate Drive with No Load Thermal Shutdown 10-Pin VSSOP Package APPLICATIONS * * * Non-Isolated Telecommunication Buck Regulator Secondary High Voltage Post Regulator +42V Automotive Systems Typical Application, Basic Step-Down Regulator 6V - 75V Input VCC VIN VIN LM34923 C1 RT GND C3 BST C4 L1 RT/SD SW SHUTDOWN VOUT RUV2 D1 UV RFB2 R3 RUV1 C2 FB RUVO GND UVO RTN RFB1 UV STATUS 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright (c) 2011-2013, Texas Instruments Incorporated LM34923 SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 www.ti.com Connection Diagram SW 1 10 BST 2 9 VCC N/C 3 8 RT RTN 4 7 FB UV 5 6 UVO VIN Figure 1. Top View 10-Lead VSSOP Table 1. Pin Descriptions Pin No. Name 1 SW Switching Node Power switching node. Connect to the output inductor, re-circulating diode or synchronous FET, and bootstrap capacitor. 2 BST Boost Pin An external capacitor is required between the BST and the SW pins (0.01uF or greater ceramic). The BST pin capacitor is charged from VCC through an internal diode when SW is low. 3 N/C Do not connect 4 RTN Ground pin Ground for the entire circuit. 5 UV Input pin for the under voltage indicator A resistor divider from VIN, or some other system voltage, programs the under-voltage detection threshold. An internal current sink is enabled when UV is below 2.5V to provide hysteresis. 6 UVO Under voltage status indicator This open drain output is high when the UV pin voltage is below 2.5V, or when the VCCUVLO function or the shutdown function is invoked. 7 FB Feedback Input from Regulated Output This pin is connected to the inverting input of the internal regulation comparator. The regulation level is 2.5V. 8 RT/SD On-time set pin and shutdown input A resistor between this pin and Vin sets the switch on-time as a function of Vin, and the frequency. The minimum recommended on-time is 200 ns at max input voltage. Taking this pin to ground shuts off the regulator. 9 VCC Output from the internal high voltage series pass regulator. Regulated at 7.5V. The internal regulator provides bias supply for the Buck switch gate driver and other internal circuitry. A 1uF ceramic capacitor to ground is required. The regulator is current limited to 30 mA. 10 VIN Input Voltage The operating input range is 6V to 75V 2 Description Application Information Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 LM34923 www.ti.com SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. Absolute Maximum Ratings (1) (2) VIN, UV to RTN -0.3V to 80V BST to RTN -0.3V to 88V SW to RTN (Steady State) -1V to VIN + 0.3V BST to VCC 80V BST to SW 10V VCC, UVO, RT to RTN -0.3V to 10V FB, RT, to RTN -0.3 to 5V ESD Rating, Human Body Model (3) 2kV For soldering specifications see: Application Note SNOA549. Junction Temperature 150C Storage Temperature Range (1) (2) (3) -55C to +150C If Military/Aerospace specified devices are required, please contact the TI Sales Office/Distributors for availability and specifications. Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device is intended to be functional. For specifications and test conditions, see the Electrical Characteristics. The human body model is a 100pF capacitor discharged through a 1.5k resistor into each pin. Operating Ratings (1) VIN 6V to 75V -40C to + 125C Operating Junction Temperature (1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device is intended to be functional. For specifications and test conditions, see the Electrical Characteristics. Electrical Characteristics Specifications with standard type are for TJ = 25C only; limits in boldface type apply over the full Operating Junction Temperature (TJ) range. Minimum and Maximum limits are specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25C, and are provided for reference purposes only. Unless otherwise stated the following conditions apply: VIN = 48V (1). Symbol Parameter Conditions Min Typ Max 7.5 7.9 Unit VCC Supply Vcc Reg Vcc Regulator Output Vin - Vcc Vcc Output Impedance Vcc Current Limit Vcc UVLO Vin = 48V 7.1 VIN = 6V, ICC = 5mA 240 Vin =6V Vin = 48V (2) 20 Vcc Increasing 45 30 mA 4 Vcc UVLO hysteresis V mV 4.8 450 V mV Iin Operating current FB = 3V, Vin = 48V 1 1.32 mA Iin Shutdown Current RT/SD = 0V 20 70 A 0.56 1.1 3 3.8 Switch Characteristics Buck switch Rds(on) Gate Drive UVLO Itest = 200 mA Vbst - Vsw Rising Gate Drive UVLO hysteresis Pre-charge switch voltage 250 At 1 mA Pre-charge switch on-time (1) (2) 2.15 V mV 0.8 V 150 ns All electrical characteristics having room temperature limits are tested during production with TA = TJ = 25C. All hot and cold limits are specified by correlating the electrical characteristics to process and temperature variations and applying statistical process control. The VCC output is intended as a self bias for the internal gate drive power and control circuits. Device thermal limitations limit external loading. Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 3 LM34923 SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 www.ti.com Electrical Characteristics (continued) Specifications with standard type are for TJ = 25C only; limits in boldface type apply over the full Operating Junction Temperature (TJ) range. Minimum and Maximum limits are specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25C, and are provided for reference purposes only. Unless otherwise stated the following conditions apply: VIN = 48V(1). Symbol Parameter Conditions Min Typ Max Unit 700 1175 1500 mA Current Limit Current Limit Threshold Current Limit Response Time Iswitch = 1.24A, Time to Switch Off 190 ns TOFF-1 OFF time generator (test 1) TOFF-2 OFF time generator (test 2) FB=0V, VIN = 75V 37 s FB=2.3V, VIN = 75V 7.2 TOFF-3 s OFF time generator (test 3) FB=0V, VIN = 10V 5.7 s TOFF-4 OFF time generator (test 4) FB=2.3V, VIN = 10V 1.25 s On Time Generator TON - 1 On-Time Vin = 10V Ron = 250K 2.2 3.3 4.51 s TON - 2 On-Time Vin = 75V Ron = 250K 300 450 565 ns Remote Shutdown Threshold Voltage at RT/SD rising 0.46 0.9 1.4 V Remote Shutdown Hysteresis 60 mV Minimum Off Time Minimum Off Time VIN = 6V 260 347 ns 2.5 2.5625 V Regulation and OV Comparators FB Reference Threshold Internal reference Trip point for switch ON FB Over-Voltage Threshold Trip point for switch OFF 2.4365 FB Bias Current 2.85 V 1 nA Under Voltage Sensing UVTH UV Threshold 2.4 2.5 2.6 V UVHYS UV Hysteresis Current UV = 2V UVBIAS UV Bias Current UV = 3V 2.7 5 7.3 A UVOVOL UVO Output Low Voltage UV = 3V, IUVO = 5mA 360 600 mV UVOIOH UVO Leakage Current UV = 2V, VUVO = 7.8V 1 nA Thermal Shutdown Temp. 165 C Thermal Shutdown Hysteresis 20 C 200 C/W 1 nA Thermal Shutdown Tsd Thermal Resistance JA 4 Junction to Ambient VSSOP Package Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 LM34923 www.ti.com SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 Typical Performance Characteristics Efficiency at 300 kHz, 10V Efficiency Comparison at 200 kHz 100 EFFICIENCY (%) 90 6V, D1 80 24V, D1 70 7.5V, D1 60 VOUT=5V, D1=DFLS1100 50 0 100 200 300 400 500 LOAD CURRENT (mA) Figure 2. Figure 3. VCC vs. VIN VCC vs. ICC Figure 4. Figure 5. ICC vs. Externally Applied VCC On-Time vs. VIN and RT 600 5 ICC (mA) 4 3 900 kHz, D1 2 200 kHz, D1 1 DCM 0 7.5 8.0 8.5 9.0 APPLIED VCC (V) 9.5 10.0 Figure 6. Figure 7. Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 5 LM34923 SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 www.ti.com Typical Performance Characteristics (continued) 4.5 Current Limit Off-Time vs. VFB Maximum Switching Frequency Figure 8. Figure 9. Voltage at the RT Pin Operating Current into VIN VOLTAGE AT THE RT PIN (V) 77 k: 4.0 34 k: 200 k: 3.5 300 k: 3.0 2.5 RT = 500 k: 2.0 1.5 1.0 5 15 25 35 45 55 65 75 VIN (V) 6 Figure 10. Figure 11. Shutdown Current into VIN UVO Pin Low Voltage vs. Sink Current Figure 12. Figure 13. Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 LM34923 www.ti.com SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 Typical Performance Characteristics (continued) VCC UVLO vs. Temperature Gate Drive UVLO vs. Temperature Figure 14. Figure 15. VCC vs. Temperature VCC Dropout vs. Temperature Figure 16. Figure 17. VCC Output Impedance vs. Temperature VCC Current Limit vs. Temperature Figure 18. Figure 19. Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 7 LM34923 SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 www.ti.com Typical Performance Characteristics (continued) 8 Reference Voltage vs. Temperature On-time vs. Temperature Figure 20. Figure 21. Minimum Off-time vs. Temperature Current Limit Threshold vs. Temperature Figure 22. Figure 23. Current Limit Off-Time vs. Temperature Operating Current vs. Temperature Figure 24. Figure 25. Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 LM34923 www.ti.com SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 Typical Performance Characteristics (continued) Shutdown Current vs. Temperature RT Pin Shutdown Threshold vs. Temperature Figure 26. Figure 27. UV Pin Threshold vs. Temperature UV Hysteresis Current vs. Temperature Figure 28. Figure 29. Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 9 LM34923 SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 www.ti.com BLOCK DIAGRAM LM34923 Input C1 VIN VCC START -UP REGULATOR C5 C3 VCC UVLO THERMAL SHUTDOWN RT ON/OFF TIMERS 0.9 V BST RT/SD VIN 2.5V C4 FEEDBACK FB L1 Delay OVER- VOLTAGE SW DRIVER LOGIC VOUT PRE CHARGE 2.85V Current Limit One- Shot D1 CURRENT LIMIT + - R3 C2 RFB2 Vth RFB1 RUV2 2.5V UVO UV RUV1 RTN 5 PA FUNCTIONAL DESCRIPTION The LM34923 Step Down Switching Regulator features all the functions needed to implement a low cost, efficient, Buck bias power converter. This high voltage regulator contains an 80 V N-Channel Buck Switch, is easy to implement and is provided in the VSSOP-10 package. The regulator is based on a control scheme using an on-time inversely proportional to VIN. The control scheme requires no loop compensation. Current limit is implemented with forced off-time, which is inversely proportional to VOUT. This scheme ensures short circuit control while providing minimum foldback. The LM34923 can be applied in numerous applications to efficiently regulate down higher voltages. This regulator is well suited for 48 Volt Telecom and the new 42V Automotive power bus ranges. Features include: Thermal Shutdown, VCC under-voltage lockout, Gate drive under-voltage lockout, Max Duty Cycle limit timer, intelligent current limit off timer, a pre-charge switch, and a programmable under voltage detector with status flag. Control Circuit Overview The LM34923 is a Buck DC-DC regulator that uses a control scheme in which the on-time varies inversely with line voltage (VIN). Control is based on a comparator and the on-time one-shot, with the output voltage feedback (FB) compared to an internal reference (2.5V). If the FB level is below the reference the buck switch is turned on for a fixed time determined by the line voltage and a programming resistor (RT). Following the ON period the switch remains off for at least the minimum off-timer period of 260 ns. If FB is still below the reference at that time the switch turns on again for another on-time period. This continues until regulation is achieved. 10 Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 LM34923 www.ti.com SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 The LM34923 operates in discontinuous conduction mode at light load currents, and continuous conduction mode at heavy load current. In discontinuous conduction mode, current through the output inductor starts at zero and ramps up to a peak during the on-time, then ramps back to zero before the end of the off-time. The next ontime period starts when the voltage at FB falls below the internal reference - until then the inductor current remains zero. In this mode the operating frequency is lower than in continuous conduction mode, and varies with load current. Therefore at light loads the conversion efficiency is maintained, since the switching losses reduce with the reduction in load and frequency. The discontinuous operating frequency can be calculated as follows: F= VOUT2 x L x 1.28 x 1020 RL x (RT)2 (1) where RL = the load resistance In continuous conduction mode, current flows continuously through the inductor and never ramps down to zero. In this mode the operating frequency is greater than the discontinuous mode frequency and remains relatively constant with load and line variations. The approximate continuous mode operating frequency can be calculated as follows: VOUT x (Vin - 0.5V) F= 1.25 x 10-10 x VIN x (RT + 500:) (2) The buck switch duty cycle is approximately equal to: DC = tON V = OUT tON + tOFF VIN (3) The output voltage (VOUT) is programmed by two external resistors as shown in the Block Diagram. The regulation point can be calculated as follows: VOUT = 2.5 x (RFB1 + RFB2) / RFB1 (4) The LM34923 regulates the output voltage based on ripple voltage at the feedback input, requiring a minimum amount of ESR for the output capacitor C2. A minimum of 25mV to 50mV of ripple voltage at the feedback pin (FB) is required for the LM34923. In cases where the capacitor ESR is too small, additional series resistance may be required (R3 in the Block Diagram). For applications where lower output voltage ripple is required the output can be taken directly from a low ESR output capacitor, as shown in Figure 30. However, R3 slightly degrades the load regulation. L1 SW RFB2 LM34923 R3 FB VOUT RFB1 C2 Figure 30. Low Ripple Output Configuration Start-Up Regulator (VCC) The high voltage bias regulator is integrated within the LM34923. The input pin (VIN) can be connected directly to line voltages between 6V and 75V, with transient capability to 80V. The VCC output is regulated at 7.5V. The VCC regulator output current is limited at approximately 30 mA. Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 11 LM34923 SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 www.ti.com C3 must be located as close as possible to the VCC and RTN pins. In applications with a relatively high input voltage, power dissipation in the bias regulator is a concern. An auxiliary voltage of between 7.5V and 10V can be diode connected to the VCC pin to shut off the VCC regulator, thereby reducing internal power dissipation. The current required into the VCC pin depends on the voltage applied to VCC and the switching frequency. See the graph "ICC vs. Externally Applied VCC." Internally a diode connects VCC to VIN requiring that the auxiliary voltage be less than VIN. The turn-on sequence is shown in Figure 31. During the initial delay (t1) VCC ramps up at a rate determined by its current limit and C3 while internal circuitry stabilizes. When VCC reaches the upper threshold of its undervoltage lock-out, the buck switch is enabled. The inductor current increases to the current limit threshold (ILIM) and during t2 VOUT increases as the output capacitor charges up. When VOUT reaches the intended voltage the average inductor current decreases (t3) to the nominal load current (IO). VIN t1 UVLO VCC Vin SW Pin 0V I LIM Inductor Current IO t2 t3 VOUT Figure 31. Startup Sequence Regulation Comparator The feedback voltage at FB is compared to an internal 2.5V reference. In normal operation (the output voltage is regulated), an on-time period is initiated when the voltage at FB falls below 2.5V. The buck switch stays on for the on-time, causing the FB voltage to rise above 2.5V. After the on-time period, the buck switch stays off until the FB voltage again falls below 2.5V. During start-up, the FB voltage will be below 2.5V at the end of each ontime, resulting in the minimum off-time of 260 ns. 12 Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 LM34923 www.ti.com SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 Over-Voltage Comparator The feedback voltage at FB is compared to an internal 2.85V reference. If the voltage at FB rises above 2.85V the on-time pulse is immediately terminated. This condition can occur if the input voltage, or the output load, change suddenly. The buck switch will not turn on again until the voltage at FB falls below 2.5V. On-Time Generator and Shutdown The on-time for the LM34923 is determined by the RT resistor, and is inversely proportional to the input voltage (Vin), resulting in a nearly constant frequency as Vin is varied over its range. The on-time equation for the LM34923 is: TON = 1.25 x 10-10 x (RT + 500:) + 30 ns (VIN - 0.5V) (5) RT should be selected for a minimum on-time (at maximum VIN) greater than 200 ns, for proper current limit operation. This requirement limits the maximum frequency for each application, depending on VIN and VOUT. The LM34923 can be remotely disabled by taking the RT/SD pin to ground. See Figure 32. The voltage at the RT/SD pin is between 1.5 and 5.0 volts, depending on Vin and the value of the RT resistor. Input Voltage VIN LM34923 RT RT/SD STOP RUN Figure 32. Shutdown Implementation Current Limit The LM34923 contains an intelligent current limit OFF timer. If the current in the Buck switch reaches the current limit threshold, the present cycle is immediately terminated, and a non-resetable OFF timer is triggered. The length of off-time is controlled by the FB voltage and VIN (see the graph Current Limit Off-Time vs. VFB). When FB = 0V, a maximum off-time is required. This condition occurs when the output is shorted, and during the initial part of start-up. This amount of time ensures safe short circuit operation up to the maximum input voltage of 75V. In cases of overload where the FB voltage is above zero volts (not a short circuit) the required current limit offtime is less. Reducing the off-time during less severe overloads reduces the amount of foldback, recovery time, and the start-up time. The off-time in microseconds is calculated from the following equation: (VIN + 1.83V) x 0.28 TOFF = (VFB x 1.05) + 0.58 (6) The current limit sensing circuit is blanked for the first 50-70 ns of each on-time so it is not falsely tripped by the current surge which occurs at turn-on. The current surge is required by the re-circulating diode (D1) for its turnoff recovery. N-Channel Buck Switch and Driver The LM34923 integrates an N-Channel Buck switch and associated floating high voltage gate driver. The gate driver circuit works in conjunction with an external bootstrap capacitor and an internal high voltage diode. A 0.01 F ceramic capacitor (C4) connected between the BST pin and SW pin provides the voltage to the driver during the on-time. Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 13 LM34923 SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 www.ti.com During each off-time, the SW pin is at approximately 0V, and the bootstrap capacitor charges from Vcc through the internal diode. The minimum OFF timer, set to 260 ns, ensures a minimum time each cycle to recharge the bootstrap capacitor. The internal pre-charge switch at the SW pin is turned on for 150 ns during the minimum off-time period, ensuring sufficient voltage exists across the bootstrap capacitor for the on-time. This feature helps prevent operating problems which can occur during very light load conditions, involving a long off-time, during which the voltage across the bootstrap capacitor could otherwise reduce below the Gate Drive UVLO threshold. The precharge switch also helps prevent startup problems which can occur if the output voltage is pre-charged prior to turn-on. After current limit detection, the pre-charge switch is turned on for the entire duration of the forced offtime . Under Voltage Detector The Under Voltage Detector can be used to monitor the input voltage, or any other system voltage as long as the voltage at the UV pin does not exceed its maximum rating. The Under Voltage Output indicator pin (UVO) is connected to the drain of an internal N-channel MOSFET capable of sustaining 10V in the off-state. An external pull-up resistor is required at UVO to an appropriate voltage to indicate the status to downstream circuitry. The off-state voltage at the UVO pin can be higher or lower than the voltage at VIN, but must not exceed 10V. The UVO pin switches low when the voltage at the UV input pin is above its threshold. Typically the monitored voltage threshold is set with a resistor divider (RUV1, RUV2) as shown in the Block Diagram. When the voltage at the UV pin is below its threshold, the internal 5 A current source at UV is enabled. As the input voltage increases, taking UV above its threshold, the current source is disabled, raising the voltage at UV to provide threshold hysteresis. The UVO output is high when the VCC voltage is below its UVLO threshold, or when the LM34923 is shutdown using the RT/SD pin (see Figure 32), regardless of the voltage at the UV pin. Thermal Protection The LM34923 should be operated so the junction temperature does not exceed 125C during normal operation. An internal Thermal Shutdown circuit is provided to shutdown the LM34923 in the event of a higher than normal junction temperature. When activated, typically at 165C, the controller is forced into a low power reset state by disabling the buck switch. This feature prevents catastrophic failures from accidental device overheating. When the junction temperature reduces below 145C (typical hysteresis = 20C) normal operation is resumed. 14 Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 LM34923 www.ti.com SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 APPLICATIONS INFORMATION SELECTION OF EXTERNAL COMPONENTS A guide for determining the component values is illustrated with a design example. Refer to the Block Diagram. The following steps will configure the LM34923 for: * Input voltage range (Vin): 15V to 75V * Output voltage (VOUT): 10V * Load current (for continuous conduction mode): 100 mA to 400 mA * Switching Frequency: 300 kHz RFB1, RFB2: VOUT = VFB x (RFB1 + RFB2) / RFB1, and since VFB = 2.5V, the ratio of RFB2 to RFB1 calculates as 3:1. Standard values of 3.01 k and 1.00 k are chosen. Other values could be used as long as the 3:1 ratio is maintained. Fs and RT: Unless the application requires a specific frequency, the choice of frequency is generally a compromise. A higher frequency allows for a smaller inductor, input capacitor, and output capacitor (both in value and physical size), while providing a lower conversion efficiency. A lower frequency provides higher efficiency, but generally requires higher values for the inductor, input capacitor and output capacitor. The maximum allowed switching frequency for the LM34923 is limited by the minimum on-time (200 ns) at the maximum input voltage, and by the minimum off-time (260 ns) at the minimum input voltage. The maximum frequency limit for each application is defined by the following two calculations: FS(max)1 = FS(max)2 = VOUT VIN(max) x 200 ns (7) VIN(min) - VOUT VIN(min) x 260 ns (8) The maximum allowed frequency is the lesser of the two above calculations. See the graph "Maximum Switching Frequency". For this exercise, Fs(max)1 calculates to 667 kHz, and Fs(max)2 calculates to 1.28 MHz. Therefore the maximum allowed frequency for this example is 667 kHz, which is greater than the 300 kHz specified for this design. Using Equation 1, RT calculates to 258 k. A standard value 261 k resistor is used. The minimum ontime calculates to 469 ns, and the maximum on-time calculates to 2.28 s. L1: The main parameter affected by the inductor is the output current ripple amplitude. The choice of inductor value therefore depends on both the minimum and maximum load currents, keeping in mind that the maximum ripple current occurs at maximum Vin. a) Minimum load current: To maintain continuous conduction at minimum Io (100 mA) if a flyback diode is used, the ripple amplitude (IOR) must be less than 200 mA p-p so the lower peak of the waveform does not reach zero. L1 is calculated using the following equation: L1 = VOUT x (VIN - VOUT) IOR x Fs x VIN (9) At Vin = 75V, L1(min) calculates to 146H. The next larger standard value (150 H) is chosen and with this value IOR calculates to 195 mA p-p at Vin = 75V, and 75 mA p-p at Vin = 15V. b) Maximum load current: At a load current of 400 mA, the peak of the ripple waveform must not reach the minimum specified value of the LM34923's current limit threshold (700 mA). Therefore the ripple amplitude must be less than 600 mA p-p, which is already satisfied in the above calculation. With L1 = 150 H, at maximum Vin and Io, the peak of the ripple is 498 mA. While L1 must carry this peak current without saturating or exceeding its temperature rating, it also must be capable of carrying the maximum specified value of the LM34923's current limit threshold without saturating, since the current limit is reached during startup. The DC resistance of the inductor should be as low as possible. For example, if the inductor's DCR is 0.5 ohm, the power dissipated at maximum load current is 0.08W. While small, it is not insignificant compared to the load power of 4W. Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 15 LM34923 SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 www.ti.com C3: The capacitor on the VCC output provides not only noise filtering and stability, but its primary purpose is to prevent false triggering of the VCC UVLO at the buck switch on/off transitions. C3 should be no smaller than 1 F. C2 and R3: When selecting the output filter capacitor C2, the items to consider are ripple voltage due to its ESR, ripple voltage due to its capacitance, and the nature of the load. A low ESR for C2 is generally desirable so as to minimize power losses and heating within the capacitor. However, the regulator requires a minimum amount of ripple voltage at the feedback input for proper loop operation. For the LM34923 the minimum ripple required at pin 7 is 25 mV p-p, requiring a minimum ripple at VOUT of 100 mV for this example. Since the minimum ripple current (at minimum Vin) is 75 mA p-p, the minimum ESR required at VOUT is 100 mV/75 mA = 1.33. Since quality capacitors for SMPS applications have an ESR considerably less than this, R3 is inserted as shown in the Block Diagram. R3's value, along with C2's ESR, must result in at least 25 mV p-p ripple at pin 7. See LOW OUTPUT RIPPLE CONFIGURATIONS for techniques to reduce the output ripple voltage. D1: A power Schottky diode is recommended. Ultra-fast recovery diodes are not recommended as the high speed transitions at the SW pin may inadvertently affect the IC's operation through external or internal EMI. The important parameters are reverse recovery time and forward voltage. The reverse recovery time determines how long the reverse current surge lasts with each turn-on of the internal buck switch. The forward voltage drop affects efficiency. The diode's reverse voltage rating must be at least as great as the maximum input voltage, plus ripple and transients, and its current rating must be at least as great as the maximum current limit specification. The diode's average power dissipation is calculated from: PD1 = VF x IOUT x (1-D) (10) Where VF is the diode's forward voltage drop, and D is the on-time duty cycle. C1: This capacitor's purpose is to supply most of the switch current during the on-time, and limit the voltage ripple at Vin, on the assumption that the voltage source feeding Vin has an output impedance greater than zero. At maximum load current, when the buck switch turns on, the current into the VIN pin suddenly increases to the lower peak of the output current waveform, ramp up to the peak value, then drop to zero at turn-off. The average input current during this on-time is the load current (400 mA). For a worst case calculation, C1 must supply this average load current during the maximum on-time. To keep the input voltage ripple to less than 1V (for this exercise), C1 calculates to: C1 = I x tON 0.4A x 2.28 Ps = 0.91 PF = 'V 1V (11) Quality ceramic capacitors in this value have a low ESR which adds only a few millivolts to the ripple. It is the capacitance which is dominant in this case. To allow for the capacitor's tolerance, temperature effects, and voltage effects, a 1.0 F, 100V, X7R capacitor is used. C4: The recommended value is 0.01F for C4, as this is appropriate in the majority of applications. A high quality ceramic capacitor, with low ESR is recommended as C4 supplies the surge current to charge the buck switch gate at turn-on. A low ESR also ensures a quick recharge during each off-time. C5: This capacitor helps avoid supply voltage transients and ringing due to long lead inductance at VIN. A low ESR, 0.1F ceramic chip capacitor is recommended, located close to the LM34923. UV and UVO pins: The Under Voltage Detector function is used to monitor a system voltage, such as the input voltage at VIN, by connecting the UV pin to two resistors (RUV1, RUV2) as shown in the Block Diagram. When the voltage at the UV pin increases above its threshold the UVO pin switches low. The UVO pin is high when the voltage at the UV input pin is below its threshold. Hysteresis is provided by the internal 5A current source which is enabled when the voltage at the UV pin is below its threshold. The resistor values are calculated using the following procedure: Choose the upper and lower thresholds (VUVH and VUVL) at VIN. RUV2 = RUV1 = 16 VUVH - VUVL VUV(HYS) = 5 PA 5 PA (12) RUV2 x 2.5V VUVL - 2.5V (13) Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 LM34923 www.ti.com SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 As an example, assume the application requires the following thresholds: VUVH = 15V and VUVL = 14V. Therefore VUV(HYS) = 1V. The resistor values calculate to: (14) (15) RUV2 = 200k, RUV1 = 43.5k Capacitor C6 is added to filter noise and ripple, which may be present on the VIN line. Where the resistor values are known, the threshold voltages and hysteresis are calculated from the following: VUVH = 2.5V + [RUV2 x ( VUVL = 2.5V x 2.5V + 5 PA)] RUV1 (16) (RUV1 + RUV2) RUV1 (17) VUV(HYS) = RUV2 x 5 A (18) The pull-up voltage for the UVO output can be any voltage under 10V. The maximum continuous current into the UVO output pin should not exceed 5 mA. FINAL CIRCUIT The final circuit is shown in Figure 33. The circuit was tested, and the resulting performance is shown in Figure 34 and Figure 35. 15V - 75V Input VIN VIN C1 1 PF GND C5 0.1 PF VCC LM34923 RT 261 k: BST RT/SD SHUT DOWN RUV1 43.2 k: RUV2 200 k: C6 1000 pF 0.01 PF C4 L1 150 PH 10V D1 RFB2 3.01 k: FB UVO VOUT SW UV RUVO 100 k: UV STATUS C3 1 PF RTN RFB1 1 k: R3 1.4: C2 10 PF GND Figure 33. LM34923 Example Circuit Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 17 LM34923 SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 www.ti.com Figure 34. Efficiency vs. Load Current and VIN Figure 35. Efficiency vs. VIN LOW OUTPUT RIPPLE CONFIGURATIONS For applications where low output ripple is required, the following options can be used to reduce or nearly eliminate the ripple. a) Reduced ripple configuration: In Figure 36, Cff is added across RFB2 to AC-couple the ripple at VOUT directly to the FB pin. This allows the ripple at VOUT to be reduced to a minimum of 25 mVp-p by reducing R3, since the ripple at VOUT is not attenuated by the feedback resistors. The minimum value for Cff is determined from: 3 x tON (max) Cff = (RFB1//RFB2) (19) where tON(max) is the maximum on-time, which occurs at the minimum input voltage. The next larger standard value capacitor should be used for Cff. L1 SW VOUT Cff LM34923 RFB2 R3 FB RFB1 C2 Figure 36. Reduced Ripple Configuration b) Minimum ripple configuration: If the application requires a lower value of ripple (<10 mVp-p), the circuit of Figure 37 can be used. R3 is removed, and the resulting output ripple voltage is determined by the inductor's ripple current and C2's characteristics. RA and CA are chosen to generate a sawtooth waveform at their junction, and that voltage is AC-coupled to the FB pin via CB. To determine the values for RA, CA and CB, use the following procedure: Calculate VA = VOUT - (VSW x (1 - (VOUT/VIN(min)))) (20) where VSW is the absolute value of the voltage at the SW pin during the off-time. If a Schottky diode is used for the flyback function, the off-time voltage is in the range of 0.5V to 1V, depending on the specific diode used, and the maximum load current. VA is the DC voltage at the RA/CA junction, and is used in the next equation. - Calculate RA x CA = (VIN(min) - VA) x tON/V 18 Submit Documentation Feedback (21) Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 LM34923 www.ti.com SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 where tON is the maximum on-time (at minimum input voltage), and V is the desired ripple amplitude at the RA/CA junction (typically 40-50 mV). RA and CA are then chosen from standard value components to satisfy the above product. Typically CA is 1000 pF to 5000 pF, and RA is 10 k to 300 k. CB is then chosen large compared to CA, typically 0.1 F. L1 SW VOUT LM34923 RA FB CA C2 RFB2 CB RFB1 Figure 37. Minimum Output Ripple Using Ripple Injection c) Alternate minimum ripple configuration: The circuit in Figure 38 is the same as that in the Block Diagram, except the output voltage is taken from the junction of R3 and C2. The ripple at VOUT is determined by the inductor's ripple current and C2's characteristics. However, R3 slightly degrades the load regulation. This circuit may be suitable if the load current is fairly constant. L1 SW LM34923 RFB2 R3 FB VOUT RFB1 C2 Figure 38. Alternate Minimum Output Ripple PC Board Layout The LM34923 regulation, over-voltage, and current limit comparators are very fast, and respond to short duration noise pulses. Layout considerations are therefore critical for optimum performance. The layout must be as neat and compact as possible, and all of the components must be as close as possible to the associated pins. The two major current loops have currents which switch very fast, and so the loops should be as small as possible to minimize conducted and radiated EMI. The first loop is formed by C1, through VIN to the SW pin, L1, C2, and back to C1. The second loop is formed by L1, C2, D1, and back to L1. Since a current equal to the load current switches between these two loops with each transition from on-time to off-time and back to on-time, it is imperative that the ground end of C1 have a short and direct connection to D1's anode, without going through vias or a lengthy route. The power dissipation in the LM34923 can be approximated by determining the total conversion loss (PIN - POUT), and then subtracting the power losses in D1, and in the inductor. The power loss in the diode is approximately: PD1 = IOUT x VF x (1-D) (22) where VF is the diode's forward voltage drop, and D is the on-time duty cycle. PL1 = IOUT2 x RL x 1.1 (23) Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 19 LM34923 SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 www.ti.com where RL is the inductor's DC resistance, and the 1.1 factor is an approximation for the AC losses. If it is expected that the internal dissipation of the LM34923 will produce excessive junction temperatures during normal operation, good use of the PC board's ground plane can help to dissipate heat. Additionally the use of wide PC board traces, where possible, can help conduct heat away from the IC. Judicious positioning of the PC board within the end product, along with the use of any available air flow (forced or natural convection) can help reduce the junction temperature. 20 Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 LM34923 www.ti.com SNVS695A - MARCH 2011 - REVISED FEBRUARY 2013 REVISION HISTORY Changes from Original (February 2013) to Revision A * Page Changed layout of National Data Sheet to TI format .......................................................................................................... 20 Submit Documentation Feedback Copyright (c) 2011-2013, Texas Instruments Incorporated Product Folder Links: LM34923 21 PACKAGE OPTION ADDENDUM www.ti.com 11-Apr-2013 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish (2) MSL Peak Temp Op Temp (C) Top-Side Markings (3) (4) LM34923MM/NOPB ACTIVE VSSOP DGS 10 1000 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 SB5B LM34923MMX/NOPB ACTIVE VSSOP DGS 10 3500 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 SB5B (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) Multiple Top-Side Markings will be inside parentheses. Only one Top-Side Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation of the previous line and the two combined represent the entire Top-Side Marking for that device. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. 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Addendum-Page 1 Samples PACKAGE MATERIALS INFORMATION www.ti.com 30-Oct-2015 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant LM34923MM/NOPB VSSOP DGS 10 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 LM34923MMX/NOPB VSSOP DGS 10 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 30-Oct-2015 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LM34923MM/NOPB VSSOP DGS 10 1000 210.0 185.0 35.0 LM34923MMX/NOPB VSSOP DGS 10 3500 367.0 367.0 35.0 Pack Materials-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest issue. 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