MIC2176-1/-2/-3
Wide Input Voltage, Synchronous Buck
Controllers Featuring Adaptive On-Time
Control
Hyper Speed Control™ Family
Hype
MLF and
4-0800 • fax + 1 (
r Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc.
MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 94 408) 474-1000 • http://www.micrel.com
Novem M9999-111710-A
ber 2010
General Description
The Micrel MIC2176-1/-2/-3 is a family of constant-frequency,
synchronous buck controllers featuring a unique digitally
modified adaptive ON-time control architecture. The MIC2176
family operates over an input supply range of 4.5V to 75V
and can be used to supply up to 15A of output current. The
output voltage is adjustable down to 0.8V with a guaranteed
accuracy of ±1%, and the device operates at a constant
switching frequency of 100kHz, 200kHz, and 300kHz.
Micrel’s Hyper Speed ControlTM architecture allows for ultra-
fast transient response while reducing the output capacitance
and also makes (High VIN)/(Low VOUT) operation possible.
This digitally modified adaptive tON ripple control architecture
combines the advantages of fixed-frequency operation and
fast transient response in a single device.
The MIC2176 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper operation
under power-sag conditions, internal soft-start to reduce
inrush current, fold-back current limit, “hiccup” mode short-
circuit protection and thermal shutdown.
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
Features
• Hyper Speed ControlTM architecture enables
- High delta V operation (VIN = 75V and VOUT = 1.2V)
- Small output capacitance
• 4.5V to 75V input voltage
• Output down to 0.8V with ±1% accuracy
• Any CapacitorTM Stable
- Zero-ESR to high-ESR output capacitance
• 100kHz/200kHz/300kHz switching frequency
• Internal compensation
• 6ms Internal soft-start
• Foldback current limit and “hiccup” mode short-circuit
protection
• Thermal shutdown
• Supports safe start-up into a pre-biased output
• –40°C to +125°C junction temperature range
• Available in 10-pin MSOP package
Applications
• Distributed power systems
• Networking/Telecom Infrastructure
• Printers, scanners, graphic cards and video cards
___________________________________________________________________________________________________________
Typical Application
MIC2176-2 Adjustable Output 200KHz Buck Converter
Efficiency
vs. Output Current
40
45
50
55
60
65
70
75
80
85
90
95
012345
O UTP UT CURRENT (A)
EFFICI ENCY (%)
28V
IN
48V
IN
60V
IN
MIC2176-2
V
OUT
= 3.3V
V
DD
= 5V LINEAR
Micrel, Inc. MIC2176
November 2010 2 M9999-111710-A
Ordering Information
Part Number Output
Voltage Switching Frequency Junction Temperature Range Package Lead Finish
MIC2176-1YMM Adjustable 100kHz –40°C to +125°C 10-pin MSOP Pb-Free
MIC2176-2YMM Adjustable 200kHz –40°C to +125°C 10-pin MSOP Pb-Free
MIC2176-3YMM Adjustable 300kHz –40°C to +125°C 10-pin MSOP Pb-Free
Pin Configur ation
10-Pin MSOP (MM)
Pin Description
Pin
Number Pin Name Pin Function
1 HSD
High-Side MOSFET Drain Connection (Input): The HSD pin in the input of the adaptive On-time
control circuitry. A 0.1uF ceramic capacitor between the HSD pin and the power ground (PGND) is
required and must be place as close as possible to the IC.
2 EN
Enable (Input): A logic level control of the output. The EN pin is CMOS compatible. Logic high or
floating = enable, logic low = shutdown. In the off state, the VDD supply current of the device is
reduced (typically 0.7mA). Do not connect the EN pin to the HSD pin.
3 FB
Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated
to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output
voltage.
4 GND
Signal ground. GND is the ground path for the device bias voltage VDD and the control circuitry. The
loop for the signal ground should be separate from the power ground (PGND) loop.
5 VDD
VDD Bias (Input): Power to the internal reference and control sections of the MIC2176. The VDD
operating voltage range is from 4.5V to 5.5V. A 1µF ceramic capacitor from the VDD pin to the PGND
pin must be placed next to the IC.
6 DL
Low-Side Drive (output): High-current driver output for external low-side MOSFET. The DL driving
voltage swings from ground to VDD.
7 PGND
Power Ground. PGND is the ground path for the buck converter power stage. The PGND pin
connects to the sources of low-side N-Channel external MOSFETs, the negative terminals of input
capacitors, and the negative terminals of output capacitors. The loop for the power ground should be
as small as possible and separate from the Signal ground (GND) loop.
8 DH
High-Side Drive (output): High-current driver output for external high-side MOSFET. The DH driving
voltage is floating on the switch node voltage (VSW). Adding a small resistor between DH pin and the
gate of the high-side N-channel MOSFETs can slow down the turn-on and turn-off time of the
MOSFETs.
Micrel, Inc. MIC2176
November 2010 3 M9999-111710-A
Pin Description (Continued)
Pin
Number Pin Name Pin Function
9 SW
Switch Node and Current-Sense input (Input): High current output driver return. The SW pin connects
directly to the switch node. Due to the high-speed switching on this pin, the SW pin should be routed
away from sensitive nodes. The SW pin also senses the current by monitoring the voltage across the
low-side MOSFET during OFF time. In order to sense the current accurately, connect the low-side
MOSFET drain to the SW pin using a Kelvin connection.
10 BST
Boost (Output): Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A Schottky
diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1F is connected
between the BST pin and the SW pin. Adding a small resistor in series with the BST pin can slow
down the turn-on time of high-side N-Channel MOSFETs.
Micrel, Inc. MIC2176
November 2010 4 M9999-111710-A
Absolute Maximum Ratings(1, 2)
VHSD to PGND................................................ −0.3V to +76V
VDD to PGND ................................................... −0.3V to +6V
VSW to PGND......................................−0.3V to (VHSD +0.3V)
VBST to VSW ........................................................ −0.3V to 6V
VBST to PGND .................................................. −0.3V to 82V
VEN to PGND ...................................... −0.3V to (VDD + 0.3V)
VFB to PGND....................................... −0.3V to (VDD + 0.3V)
PGND to GND.............................................. −0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10sec)........................ 260°C
Operating Ratings(3)
Supply Voltage (VHSD) ....................................... 4.5V to 75V
Bias Voltage (VDD)............................................ 4.5V to 5.5V
Enable Input (VEN)................................................. 0V to VDD
Junction Temperature (TJ) ........................ −40°C to +125°C
Junction Thermal Resistance
MSOP (θJA) ..................................................130.5°C/W
Continuous Power Dissipation
(derate 5.6mW/°C above 70°C)
(TA = 70°C) ........................................................421mW
ESD (Human Body Mode) .......................................... 1.5kV
Electrical Characteristics(4)
VIN = VHSD = 48V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C TJ +125°C.
Parameter Condition Min. Typ. Max. Units
Power Supply Input
HSD Voltage Range (VHSD)(5) 4.5 75 V
VDD Bias Voltage
Operating Bias Voltage (VDD) 4.5 5 5.5 V
Undervoltage Lockout Trip Level VDD Rising 3.2 3.85 4.45 V
UVLO Hysteresis 370 mV
Quiescent Supply Current VFB = 1.5V 1.4 3 mA
Shutdown Supply Current SW = unconnected, VEN = 0V 0.7 2 mA
Reference
0°C TJ 85°C (±1.0%) 0.792 0.8 0.808
Feedback Reference Voltage -40°C TJ 125°C (±1.5%) 0.788 0.8 0.812 V
FB Bias Current VFB = 0.8V 5 500 nA
Enable Control
EN Logic Level High 4.5V < VDD < 5.5V 1.2 0.85 V
EN Logic Level Low 4.5V < VDD < 5.5V 0.78 0.4 V
EN Bias Current VEN = 0V 50 100 µA
Oscillator
MIC2176-1 75 100 125
MIC2176-2 150 200 250
Switching Frequency (6)
MIC2176-3 225 300 375
kHz
MIC2176-1, VFB = 0V, HSD=4V, VO = 3.3V 96
MIC2176-2, VFB = 0V, HSD=4V, VO = 3.3V 93
Maximum Duty Cycle (7)
MIC2176-3, VFB = 0V, HSD=4V, VO = 3.3V 89
%
Minimum Duty Cycle VFB > 0.8V 0 %
Minimum Off-Time 360 ns
Minimum On-Time 60 ns
Micrel, Inc. MIC2176
November 2010 5 M9999-111710-A
Electrical Characteristics(4) (Continued)
VIN = VHSD = 48V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C TJ +125°C.
Parameter Condition Min. Typ. Max. Units
Soft Start
Soft-Start time 6 ms
Short Circuit Protection
Current-Limit Threshold VFB = 0.8V 103 130 162 mV
Short-Circuit Threshold VFB = 0V 19 48 77 mV
FET Drivers
DH, DL Output Low Voltage ISINK = 10mA 0.1 V
DH, DL Output High Voltage ISOURCE = 10mA
VDD - 0.1V
Or
VBST - 0.1V
V
DH On-Resistance, High State 2.1 3.3
DH On-Resistance, Low State 1.8 3.3
DL On-Resistance, High State 1.8 3.3
DL On-Resistance, Low State 1.2 2.3
SW Leakage Current VSW = 48V, VDD = 5V, VBST = 53V 55 µA
HSD Leakage Current VSW = 48V, VDD = 5V, VBST = 53V 55 µA
Thermal Protection
Over-Temperature Shutdown TJ Rising 160 °C
Over-Temperature Shutdown Hysteresis 25 °C
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. Specification for packaged product only.
5. The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have enough low voltage VTH.
6. Measured in test mode.
7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 360ns.
Micrel, Inc. MIC2176
November 2010 6 M9999-111710-A
Typical Characteristics
VIN Operati ng Supply Current
vs. Input Voltage
0
5
10
15
20
25
30
0 10203040506070
INPUT VO LTA GE (V)
V
IN
Shut dow n Current
vs. Input Voltage
0
5
10
15
20
25
30
35
40
0 10203040506070
INPUT VO LTAGE ( V)
V
DD
Operating Supply Current
vs. Input Voltage
0
2
4
6
8
10
SHUTDOWN CUR RENT (µA )
SUPPLY CURRENT (mA)
SUPPL Y CURRE NT ( m A)
MIC2176-2
V
OUT
= 3.3V
I
OUT
= 0A
V
DD
= 5V
SW ITCHING
MIC2176-2
V
OUT
= 3.3V
I
OUT
= 0A
V
DD
= 5V
SWITCHING
V
DD
= 5V
V
EN
= 0V
0 10203040506070
I NPUT VO L TAG E ( V)
Feedback Voltage
vs. I nput Voltage
0.792
0.794
0.796
0.798
0.800
0.802
0.804
0.806
0.808
0 10203040506070
INPUT VO LTAGE ( V)
Curre n t Limit
vs. I nput Voltage
0
5
10
15
20
CURRENT L IM IT ( A)
Total Regulation
vs. Input Voltage
0.0%
0.2%
0.4%
0.6%
0.8%
1.0%
TOTAL REG ULATIO N ( % )
FEEDBACK V OLTAGE ( V)
0 10203040506070
I NPUT VOLTA GE (V)
V
OUT
= 3.3V
V
DD
= 5V
I
OUT
= 0A to 5A
V
OUT
= 3.3V
V
DD
= 5V
I
OUT
= 0A
0 10203040506070
INPUT VO LTAGE (V)
V
OUT
= 1.2V
V
DD
= 5V
Switching Frequency
vs. Input Voltage
160
180
200
220
240
SW ITCHI NG FREQUENCY ( kHz)
0 10203040506070
I NPUT VO L TAG E (V)
MIC2176-2
V
OUT
= 3.3V
I
OUT
= 0A
V
DD
= 5V
Micrel, Inc. MIC2176
November 2010 7 M9999-111710-A
Typical Characteristics (Continued)
V
DD
Operating Supply Current
vs. Temperat ure
0
2
4
6
8
10
-50 -20 10 40 70 100 130
TEMPERA TURE (° C)
V
DD
Shutdow n Current
vs. Temperature
0
0.2
0.4
0.6
0.8
1
-50 -20 10 40 70 100 130
TEM P ERATURE (°C)
V
DD
UVLO Threshold
vs. Temperature
3.2
3.4
3.6
3.8
4.0
4.2
VDD THRESHOLD ( V)
Rising
SUPPLY CURRENT ( m A)
SUPPLY CURRENT (mA)
MIC2176-2
V
IN
= 48V
V
OUT
= 3.3V
I
OUT
= 0A
V
DD
= 5V
SWITCHING
V
IN
= 48V
I
OUT
= 0A
V
DD
= 5V
V
EN
= 0V
Falling VIN = 48V
3.0
-50 -20 10 40 70 100 130
TEMPERATURE (° C)
V
IN
Shutdown Current
vs. Temperature
0
5
10
15
20
25
30
-50 -20 10 40 70 100 130
TEM PERATURE ( ° C)
Current Lim it
vs. Temperature
0
5
10
15
20
V
IN
Operating Supply Current
vs. Temperature
10
12
14
16
18
20
22
-50 -20 10 40 70 100 130
TEM PERATURE ( °C)
SUPPLY CURRENT ( mA)
SUPPLY CURRENT (µA)
CURRENT LIM IT (A)
MIC2176-2
V
IN
= 48V
V
OUT
= 3.3V
I
OUT
= 0A
V
DD
= 5V
SW ITCHING
V
IN
= 48V
V
DD
= 5V
I
OUT
= 0A
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
-50 -20 10 40 70 100 130
TEMPERA T URE (° C)
Load Regul ation
vs. Temperat u re
-0.2%
0.0%
0.2%
0.4%
0.6%
0.8%
1.0%
-50 -20 10 40 70 100 130
TEMPERATURE (°C)
Feedback Voltage
vs. Temperature
0.792
0.794
0.796
0.798
0.800
0.802
0.804
0.806
0.808
-50 -20 10 40 70 100 130
TEM P ERATURE ( °C)
Line Regul ation
vs. Temperature
-0.2%
0.0%
0.2%
0.4%
0.6%
0.8%
1.0%
LINE REGULATION (% )
FEEBACK VOLTAGE (V)
-50 -20 10 40 70 100 130
TEM P ERATURE ( °C)
V
IN
= 6V to 70V
V
OUT
= 3.3V
I
OUT
= 0A
V
DD
= 5V
V
IN
= 48V
V
OUT
= 3.3V
I
OUT
= 0A
V
DD
= 5V
LO AD REGULATION ( % )
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
I
OUT
= 0A to 5A
Switching Frequency
vs. Temperature
160
180
200
220
240
SW I T CHING F REQUENCY (kHz )
-50 -20 10 40 70 100 130
TEMPERA TURE ( ° C)
MIC2176-1
V
IN
= 48V
V
OUT
= 3.3V
I
OUT
= 0A
V
DD
= 5V
EN Bi as Current
vs. Temperature
0
20
40
60
80
100
EN BIAS CURRENT (µA)
-50 -20 10 40 70 100 130
TEM P ERATURE ( °C)
V
IN
= 48V
V
EN
= 0V
V
DD
= 5V
Micrel, Inc. MIC2176
November 2010 8 M9999-111710-A
Typical Characteristics (Continued)
Efficiency
vs. Output Current
40
45
50
55
60
65
70
75
80
85
90
95
012345
OUTPUT CURRENT (A)
Feedback Voltage
vs. Output Current
0.792
0.794
0.796
0.798
0.800
0.802
0.804
0.806
0.808
012345
O UTPUT CURRENT ( A)
Out put Voltage
vs. Output C urrent
3.267
3.278
3.289
3.300
3.311
3.322
3.333
FEE DBACK V OLTAGE ( V )
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
OUTPUT VO LTAG E ( V)
28V
IN
EFFI C I ENCY (%)
48V
IN
60V
IN
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
MIC2176-2
V
OUT
= 3.3V
V
DD
= 5V LINEAR
012345
O UTPUT CURRENT ( A)
Li ne Regulation
vs. Output Current
-0.2%
0.0%
0.2%
0.4%
0.6%
0.8%
1.0%
012345
O U TP UT CURRE NT (A)
Switching Frequency
vs. Output Current
160
180
200
220
240
012345
O UT PUT CU RRENT (A )
IC Case Temperature* (V
IN
= 28V)
vs. Output Current
0
20
40
60
CASE TEM PERATURE ( ° C)
V
IN
= 6V to 70V
V
OUT
= 3.3V
V
DD
= 5V
SW ITCHI NG FREQUENCY (kHz )
LINE REGULATIO N (% )
MIC2176-2
VIN = 48V
VOUT = 3.3V
VDD = 5V
012345
O UT PUT CURRENT (A )
MIC2176-2
V
IN
= 28V
V
OUT
= 3.3V
V
DD
= 5V
IC Case Temperature* (V
IN
= 48V)
vs. Output Current
0
20
40
60
80
CASE T EMPERATURE (° C)
012345
O UT PUT CURRE NT (A )
MIC2176-2
V
IN
= 48V
V
OUT
= 3.3V
V
DD
= 5V
IC C ase Temperature* ( V
IN
= 60V)
vs. Output Current
0
20
40
60
80
100
CASE TEMPERATURE (° C)
012345
O UT PUT CURRENT (A)
MIC2176-2
V
IN
= 60V
V
OUT
= 3.3V
V
DD
= 5V
Case Temperature* : The temperature measurement was taken at the hottest point on the MIC2176 case mounted on a 5 square inch PCB, see
Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting
components.
Micrel, Inc. MIC2176
November 2010 9 M9999-111710-A
Typical Characteristics (Continued)
Efficiency ( V
IN
= 28V)
vs. O u tput Cu rrent
60
65
70
75
80
85
90
95
EFFI CIENCY (%)
Ef fi ciency ( V
IN
= 60V )
vs. Output Current
30
35
40
45
50
55
60
65
70
75
80
85
90
EFFICIENCY ( %)
01234567
O UT PUT CURRENT ( A)
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
MIC2176-2
VDD = 5V LINEA
R
Ef fici ency (V
IN
= 48V)
vs. O utput Current
40
45
50
55
60
65
70
75
80
85
90
EFFI CIENCY (%)
01234567
O UT PUT CU RRENT (A )
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
MIC2176-2
VDD = 5V LINEA
R
01234567
O UT PUT CURRENT (A )
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
MIC2176-2
VDD = 5V LINEA
R
Micrel, Inc. MIC2176
November 2010 10 M9999-111710-A
Functional Characteristics
Micrel, Inc. MIC2176
November 2010 11 M9999-111710-A
Functional Characteristics (Continued)
Micrel, Inc. MIC2176
November 2010 12 M9999-111710-A
Functional Characteristics (Continued)
Micrel, Inc. MIC2176
November 2010 13 M9999-111710-A
Functional Diagram
Figure 1. MIC2176 Functional Diagram
Micrel, Inc. MIC2176
November 2010 14 M9999-111710-A
Functional Description
The MIC2176 is an adaptive on-time synchronous buck
controller family built for high-input voltage and low
output voltage applications. It is designed to operate
over a wide input voltage range from, 4.5V to 75V and
the output is adjustable with an external resistive divider.
A digitally modified adaptive on-time control scheme is
employed in to obtain a constant switching frequency
and to simplify the control compensation. Over-current
protection is implemented by sensing low-side
MOSFET’s RDS(ON). The device features internal soft-
start, enable, UVLO, and thermal shutdown.
Theory of Operation
Figure 1 illustrates the block diagram of the MIC2176.
The output voltage is sensed by the MIC2176 feedback
pin FB via the voltage divider R1 and R2, and compared
to a 0.8V reference voltage VREF at the error comparator
through a low-gain transconductance (gm) amplifier. If
the feedback voltage decreases and the amplifier output
is below 0.8V, then the error comparator will trigger the
control logic and generate an ON-time period. The ON-
time period length is predetermined by the “Fixed tON
Estimator” circuitry:
SWIN
OUT
ed)ON(estimat fV
V
×
=t (1)
where VOUT is the output voltage, VIN is the power stage
input voltage, and fSW is the switching frequency
(100kHz for MIC2176-1, 200kHz for MIC2176-2, and
300kHz for MIC2176-3).
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
360ns, the MIC2176 control logic will apply the tOFF(min)
instead. tOFF(min) is required to maintain enough energy in
the boost capacitor (CBST) to drive the high-side
MOSFET.
The maximum duty cycle is obtained from the 360ns
tOFF(min):
SS
OFF(min)S
max t
360ns
1
t
tt
D−=
−
= (2)
where tS = 1/fSW. It is not recommended to use MIC2176
with a OFF-time close to tOFF(min) during steady-state
operation.
The adpative ON-time control scheme results in a
constant switching frequency in the MIC2176. The actual
ON-time and resulting switching frequency will vary with
the different rising and falling times of the external
MOSFETs. Also, the minimum tON results in a lower
switching frequency in high VIN to VOUT applications,
such as 48V to 1.0V. The minimum tON measured on the
MIC2176 evaluation board is about 60ns. During load
transients, the switching frequency is changed due to the
varying OFF-time.
To illustrate the control loop operation, we will analyze
both the steady-state and load transient scenarios. For
easy analysis, the gain of the gm amplifier is assumed to
be 1. With this assumption, the inverting input of the
error comparator is the same as the feedback voltage.
Figure 2 shows the MIC2176 control loop timing during
steady-state operation. During steady-state, the gm
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple plus injected
voltage ripple, to trigger the ON-time period. The ON-
time is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
Figure 2. MIC2176 Control Loop Timing
Micrel, Inc. MIC2176
November 2010 15 M9999-111710-A
Figure 3 shows the operation of the MIC2176 during a
load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC2176 converter.
Figure 3. MIC2176 Load Transien t Response
Unlike true current-mode control, the MIC2176 uses the
output voltage ripple to trigger an ON-time period. The
output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough. The MIC2176 control loop has the advantage of
eliminating the need for slope compensation.
In order to meet the stability requirements, the MIC2176
feedback voltage ripple should be in phase with the
inductor current ripple and large enough to be sensed by
the gm amplifier and the error comparator. The
recommended feedback voltage ripple is 20mV~100mV.
If a low ESR output capacitor is selected, then the
feedback voltage ripple may be too small to be sensed
by the gm amplifier and the error comparator. Also, the
output voltage ripple and the feedback voltage ripple are
not necessarily in phase with the inductor current ripple if
the ESR of the output capacitor is very low. In these
cases, ripple injection is required to ensure proper
operation. Please refer to “Ripple Injection” subsection in
Application Information for more details about the ripple
injection technique.
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC2176 implements an internal digital soft-start by
making the 0.8V reference voltage VREF ramp from 0 to
100% in about 6ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a stair-
case VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
Current Limit
The MIC2176 uses the RDS(ON) of the low-side power
MOSFET to sense over-current conditions. This method
will avoid adding cost, board space and power losses
taken by discrete current sense resistors. The low-side
MOSFET is used because it displays much lower
parasitic oscillations during switching than the high-side
MOSFET.
In each switching cycle of the MIC2176 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. The sensed voltage is
compared with a current-limit threshold voltage VCL after
a blanking time of 150ns. If the sensed voltage is over
VCL, which is 133mV typical at 0.8V VFB, then the
MIC2176 turns off the high-side and low-side MOSFETs
and a soft-start sequence is triggered. This mode of
operation is called “hiccup mode” and its purpose is to
protect the downstream load in case of a hard short. The
current limit threshold VCL has a foldback characteristic
related to the FB voltage. Please refer to the “Typical
Characteristics” for the curve of current limit threshold vs.
FB voltage percentage. The circuit in Figure 4 illustrates
the MIC2176 current limiting circuit.
Figure 4. MIC2176 Current Limiting Circuit
Micrel, Inc. MIC2176
November 2010 16 M9999-111710-A
Using the typical VCL value of 130mV, the current-limit
value is roughly estimated as: MOSFET Gate Drive
DS(ON)
CL R
130mV
I≈ (3)
For designs where the current ripple is significant
compared to the load current IOUT, or for low duty cycle
operation, calculating the current limit ICL should take
into account that one is sensing the peak inductor
current and that there is a blanking delay of
approximately 150ns.
2
I
L
T
R
L(pp)
DLY
DS(ON)
−
V
130mV
IOUT
CL
×
+= (4)
L f
D)(1V
I
SW
OUT
L(pp) ×
−×
= (5)
The MIC2176 high-side drive circuit is designed to
switch an N-Channel MOSFET. Figure 1 shows a
bootstrap circuit, consisting of D1 (a Schottky diode is
recommended) and CBST. This circuit supplies energy to
the high-side drive circuit. Capacitor CBST is charged
while the low-side MOSFET is on and the voltage on the
SW pin is approximately 0V. When the high-side
MOSFET driver is turned on, energy from CBST is used to
turn the MOSFET on. As the high-side MOSFET turns
on, the voltage on the SW pin increases to
approximately VIN. Diode D1 is reverse biased and CBST
floats high while continuing to keep the high-side
MOSFET on. The bias current of the high-side driver is
less than 10mA so a 0.1F to 1F is sufficient to hold
the gate voltage with minimal droop for the power stroke
(high-side switching) cycle, i.e. BST = 10mA x
3.33s/0.1F = 333mV. When the low-side MOSFET is
turned back on, CBST is recharged through D1. A small
resistor RG, which is in series with CBST, can be used to
slow down the turn-on time of the high-side N-channel
MOSFET.
where The drive voltage is derived from the VDD supply voltage.
The nominal low-side gate drive voltage is VDD and the
nominal high-side gate drive voltage is approximately
VDD – VDIODE, where VDIODE is the voltage drop across
D1. An approximate 30ns delay between the high-side
and low-side driver transitions is used to prevent current
from simultaneously flowing unimpeded through both
MOSFETs.
VOUT = The output voltage
TDLY = Current-limit blanking time, 150ns typical
IL(pp) = Inductor current ripple peak-to-peak value
D = Duty Cycle
fSW = Switching frequency
The MOSFET RDS(ON) varies 30% to 40% with
temperature; therefore, it is recommended to add a 50%
margin to ICL in the above equation to avoid false current
limiting due to increased MOSFET junction temperature
rise. It is also recommended to connect SW pin directly
to the drain of the low-side MOSFET to accurately sense
the MOSFETs RDS(ON).
Micrel, Inc. MIC2176
November 2010 17 M9999-111710-A
Application Information
MOSFET Selection
SWGside]-G[high f Q(avg)I ×=
SWGSISSside]-G[low f V C (avg)I ××
The MIC2176 controller works from power stage input
voltages of 4.5V to 73V and has an external 4.5V to 5.5V
VIN to provide power to turn the external N-Channel
power MOSFETs for the high- and low-side switches.
For applications where VDD < 5V, it is necessary that the
power MOSFETs used are sub-logic level and are in full
conduction mode for VGS of 2.5V. For applications when
VDD > 5V; logic-level MOSFETs, whose operation is
specified at VGS = 4.5V must be used.
There are different criteria for choosing the high-side and
low-side MOSFETs. These differences are more
significant at lower duty cycles. In such an application,
the high-side MOSFET is required to switch as quickly
as possible to minimize transition losses, whereas the
low-side MOSFET can switch slower, but must handle
larger RMS currents. When the duty cycle approaches
50%, the current carrying capability of the high-side
MOSFET starts to become critical.
It is important to note that the on-resistance of a
MOSFET increases with increasing temperature. A 75°C
rise in junction temperature will increase the channel
resistance of the MOSFET by 50% to 75% of the
resistance specified at 25°C. This change in resistance
must be accounted for when calculating MOSFET power
dissipation and in calculating the value of current limit.
Total gate charge is the charge required to turn the
MOSFET on and off under specified operating conditions
(VDS and VGS). The gate charge is supplied by the
MIC2176 gate-drive circuit. At 300kHz switching
frequency, the gate charge can be a significant source of
power dissipation in the MIC2176. At low output load,
this power dissipation is noticeable as a reduction in
efficiency. The average current required to drive the
high-side MOSFET is:
(6)
where:
IG[high-side](avg) = Average high-side MOSFET gate
current
QG = Total gate charge for the high-side MOSFET taken
from the manufacturer’s data sheet for VGS = VDD.
fSW = Switching Frequency
The low-side MOSFET is turned on and off at VDS = 0
because an internal body diode or external freewheeling
diode is conducting during this time. The switching loss
for the low-side MOSFET is usually negligible. Also, the
gate-drive current for the low-side MOSFET is more
accurately calculated using CISS at VDS = 0 instead of
gate charge.
For the low-side MOSFET:
(7)
=
Since the current from the gate drive comes from the
VDD, the power dissipated in the MIC2176 due to gate
drive is:
(8)
(avg))I (avg)(I V P side]-G[lowside]G[high-DDGATEDRIVE +×=
ACCONDUCTIONSW P P P +
A convenient figure of merit for switching MOSFETs is
the on resistance times the total gate charge RDS(ON) ×
QG. Lower numbers translate into higher efficiency. Low
gate-charge logic-level MOSFETs are a good choice for
use with the MIC2176. Also, the RDS(ON) of the low-side
MOSFET will determine the current-limit value. Please
refer to “Current Limit” subsection is Functional
Description for more details.
Parameters that are important to MOSFET switch
selection are:
• Voltage rating
• On-resistance
• Total gate charge
The voltage ratings for the high-side and low-side
MOSFETs are essentially equal to the power stage input
voltage VHSD. A safety factor of 20% should be added to
the VDS(max) of the MOSFETs to account for voltage
spikes due to circuit parasitic elements.
The power dissipated in the MOSFETs is the sum of the
conduction losses during the on-time (PCONDUCTION) and
the switching losses during the period of time when the
MOSFETs turn on and off (PAC).
(9)
=
DS(ON)
2
SW(RMS)CONDUCTION R I P ×=
AC(on)) AC(off
AC P P P +=
(10)
(11)
where:
RDS(ON) = On-resistance of the MOSFET switch
D = Duty Cycle = VOUT / VHSD
Micrel, Inc. MIC2176
November 2010 18 M9999-111710-A
Making the assumption that the turn-on and turn-off
transition times are equal; the transition times can be
approximated by:
G
HSDOSSINISS
TI
VCVC
t×+×
=
SWTPKDHSD AC f t I) V(V P ×××+=
(12)
where:
CISS and COSS are measured at VDS = 0
IG = Gate-drive current
The total high-side MOSFET switching loss is:
(13)
where:
tT = Switching transition time
VD = Body diode drop (0.5V)
fSW = Switching Frequency
The high-side MOSFET switching losses increase with
the switching frequency and the input voltage VHSD. The
low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor.
A good compromise between size, loss and cost is to set
the inductor ripple current to be equal to 20% of the
maximum output current.
The inductance value is calculated by Equation 14:
OUT(max)swIN(max)
OUTIN(max)OUT
I20% f V
)V(VV
L×××
−×
= (14)
where:
fSW = Switching frequency, 300kHz
20% = Ratio of AC ripple current to DC output current
VIN(max) = Maximum power stage input voltage
The peak-to-peak inductor current ripple is:
L f V
)V(VV
I
swIN(max)
OUTIN(max)OUT
L(pp) ××
−×
=Δ (15)
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
IL(pk) =IOUT(max) + 0.5 × IL(pp) (16)
The RMS inductor current is used to calculate the I2R
losses in the inductor.
12
I
II
2
L(PP)
2
OUT(max)L(RMS) += (17)
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC2176 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor.
Micrel, Inc. MIC2176
November 2010 19 M9999-111710-A
Copper loss in the inductor is calculated by Equation 18:
PINDUCTOR(Cu) = IL(RMS)
2 × RWINDING (18)
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
PWINDING(Ht) = RWINDING(20°C) ×
(1 + 0.0042 × (TH – T20°C)) (19)
where:
TH = temperature of wire under full load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
Output Capacitor Selection
The type of the output capacitor is usually determined by
its ESR (equivalent series resistance). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are tantalum, low-ESR aluminum electrolytic, OS-
CON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view. The maximum value of ESR is
calculated:
L(PP)
OUT(pp)
CI
V
ESR OUT ≤ (20)
where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
IL(PP) = peak-to-peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 21:
()
2
CL(PP)
2
SWOUT
L(PP)
OUT(pp) OUT
ESRI
8fC
I
V×+
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
××
=
(21)
where:
D = duty cycle
COUT = output capacitance value
fsw = switching frequency
As described in the “Theory of Operation” subsection in
Functional Description, the MIC2176 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly. Also,
the output voltage ripple should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 22:
12
I
IL(PP)
(RMS)COUT =
OUTOUTOUT C
2
(RMS)C)DISS(C ESRIP ×=
(22)
The power dissipated in the output capacitor is:
(23)
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
VIN = IL(pk) × ESRCIN (24)
Micrel, Inc. MIC2176
November 2010 20 M9999-111710-A
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
(25) D)(1DII OUT(max)CIN(RMS) −××≈
The power dissipated in the input capacitor is:
PDISS(CIN) = ICIN(RMS)
2 × ESRCIN (26)
Voltage Setting Components
The MIC2176 requires two resistors to set the output
voltage as shown in Figure 5:
Figure 5. Voltage-Divider Configuration
The output voltage is determined by the equation:
)
R2
R1
(1VV FBOUT +×= (27)
where, VFB = 0.8V. A typical value of R1 can be between
3k and 10k. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small in value, it will decrease the efficiency of the power
supply, especially at light loads. Once R1 is selected, R2
can be calculated using:
FBOUT
FB
VV
R1V
R2 −
×
= (28)
Ripple Injection
The VFB ripple required for proper operation of the
MIC2176 gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as a 1V, the output voltage ripple is
only 10mV to 20mV, and the feedback voltage ripple is
less than 20mV. If the feedback voltage ripple is so small
that the gm amplifier and error comparator can’t sense it,
then the MIC2176 will lose control and the output voltage
is not regulated. In order to have some amount of VFB
ripple, a ripple injection method is applied for low output
voltage ripple applications.
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1. Enough ripple at the feedback voltage due to the
large ESR of the output capacitors.
As shown in Figure 6a, the converter is stable
without any ripple injection. The feedback voltage
ripple is:
(pp)
LCFB(pp) IESR
R2R1
R2
VOUT ××
+
=
(pp)
LFB(pp) IESRV×≈
(29)
where IL(pp) is the peak-to-peak value of the
inductor current ripple.
2. Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin
through a feedforward capacitor Cff in this situation,
as shown in Figure 6b. The typical Cff value is
between 1nF and 100nF. With the feedforward
capacitor, the feedback voltage ripple is very close
to the output voltage ripple:
(30)
3. Virtually no ripple at the FB pin voltage due to the
very-low ESR of the output capacitors:
Figure 6a. Enough Ripple at FB
Micrel, Inc. MIC2176
November 2010 21 M9999-111710-A
If the voltage divider resistors R1 and R2 are in the k
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in k range.
Figure 6b. Inadequate Ripple at FB
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 24:
Figure 6c. Invisible Ripple at FB
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 6c. The injected ripple
is:
τ
×
××××=
SW
divINFB(pp) f
1
D)-(1DKVV (31)
R1//R2R
R1//R2
K
inj
div +
= (32)
where:
VIN = Power stage input voltage
D = Duty cycle
fSW = Switching frequency
τ = (R1//R2//Rinj) × Cff
In Equations 21 and 22, it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
T
f
1
SW
<<=
×
ττ
(33)
D)(1D
f
V
V
KSW
IN
FB(pp)
div −×
×
×=
τ
(34)
Then the value of Rinj is obtained as:
1)
K
1
((R1//R2)R
div
inj −×= (35)
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
Micrel, Inc. MIC2176
November 2010 22 M9999-111710-A
PCB Layout Guidelines
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC2176 converter.
IC
• The 1µF ceramic capacitor, which is connected to
the VDD pin, must be located right at the IC. The
VDD pin is very noise sensitive and placement of the
capacitor is very critical. Use wide traces to connect
to the VDD and PGND pins.
• The signal ground pin (GND) must be connected
directly to the ground planes. Do not route the GND
pin to the PGND pin on the top layer.
• Place the IC close to the point of load (POL).
• Use fat traces to route the input and output power
lines.
• Signal and power grounds should be kept separate
and connected at only one location.
Input Capacitor
• Place the input capacitor next.
• Place the input capacitors on the same side of the
board and as close to the MOSFETs as possible.
• Place several vias to the ground plane close to the
input capacitor ground terminal.
• Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
• Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
• If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
• In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over-
voltage spike seen on the input supply with power is
suddenly applied.
RC Snubber
• Place the RC snubber on the same side of the board
and as close to the SW pin as possible.
Inductor
• Keep the inductor connection to the switch node
(SW) short.
• Do not route any digital lines underneath or close to
the inductor.
• Keep the switch node (SW) away from the feedback
(FB) pin.
• The SW pin should be connected directly to the
drain of the low-side MOSFET to accurate sense the
voltage across the low-side MOSFET.
• To minimize noise, place a ground plane underneath
the inductor.
Output Capacitor
• Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
• Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
• The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high-current
load trace can degrade the DC load regulation.
MOSFETs
• Low-side MOSFET gate drive trace (DL pin to
MOSFET gate pin) must be short and routed over a
ground plane. The ground plane should be the
connection between the MOSFET source and PGND.
• Chose a low-side MOSFET with a high CGS/CGD ratio
and a low internal gate resistance to minimize the
effect of dv/dt inducted turn-on.
• Do not put a resistor between the Low-side
MOSFET gate drive output and the gate.
• Use a 4.5V VGS rated MOSFET. Its higher gate
threshold voltage is more immune to glitches than a
2.5V or 3.3V rated MOSFET. MOSFETs that are
rated for operation at less than 4.5V VGS should not
be used.
Micrel, Inc. MIC2176
November 2010 23 M9999-111710-A
Evaluation Board Schematic
Figure 7. Schematic of MIC2176 Evaluation Board
(J1, J9, J12, R12, and R13 are for testing purposes)
Micrel, Inc. MIC2176
November 2010 24 M9999-111710-A
Bill of Materials
Item Part Number Manufacturer Description Qty
C1 B41125A9336M EPCOS(1) 33µF Aluminum Capacitor, SMD, 100V 1
C2, C3 GRM32ER72A225K Murata(2) 2.2µF/100V Ceramic Capacitor, X7R, Size 1210 2
C4 6SEPC470M Sanyo(3) 470µF/6.3V OSCON Capacitor 1
C5, C15 GRM32ER60J104KA94D Murata(2) 100µF/6.3V Ceramic Capacitor, X7R, Size 1210 2
C6, C7, C16 GRM188R71H104KA94L Murata(2) 0.1µF/6.3V Ceramic Capacitor, X7R, Size 0603 3
C8 GRM188R70J105KA01D Murata(2) 1µF/6.3V Ceramic Capacitor, X7R, Size 0603 1
C9, C10 GRM188R72A104KA35D Murata(2) 0.1µF/100V Ceramic Capacitor, X7R, Size 0603 2
C11 GRM188R72A102KA01D Murata(2) 1nF/100V Cermiac Capacitor, X7R, Size 0603 1
C12 GRM188R71H103K Murata(2) 10nF/50V Ceramic Capacitor, X7R, Size 0603 1
C14 GRM31CR60J475KA01L Murata(2) 4.7µF/6.3V Ceramic Capacitor, X5R, Size 1206 1
D1 BAT46W Diodes, Inc.(4) 100V Small Signal Schottky Diode, SOD123 1
D2 CMDZ5L6 Central Semi(5) 5.6V Zener Diode, SOD323 1
L1 HCL1305-4R0-R Cooper Bussmann(6) 4.0µH Inductor, 10A RMS Current 1
Q1 SIR432DP Vishay(7) MOSFET, N-CH, Power SO-8 1
Q2 SIR804DP Vishay(7) MOSFET, N-CH, Power SO-8 1
Q3 FCX493 ZETEX(4) 100V NPN Transistor, SOT89 1
R1, R3 CRCW060310K0FKEA Vishay Dale(7) 10k Resistor, Size 0603, 1% 2
R2 CRCW08051R21FKEA Vishay Dale(7) 1.21 Resistor, Size 0805, 5% 1
R4 CRCW060380K6FKEA Vishay Dale(7) 80.6k Resistor, Size 0603, 1% 1
R5 CRCW060340K2FKEA Vishay Dale(7) 40.2k Resistor, Size 0603, 1% 1
R6 CRCW060320K0FKEA Vishay Dale(7) 20k Resistor, Size 0603, 1% 1
Notes:
1. EPCOS: www.epcos.com.
2. Murata: www.murata.com.
3. Sanyo: www.sanyo.com.
4. Diodes Inc.: www.diodes.com.
5. Central Semi: www.centralsemi.com.
6. Cooper Bussmann: www.cooperbussmann.com.
7. Vishay: www.vishay.com.
Micrel, Inc. MIC2176
November 2010 25 M9999-111710-A
Bill of Materials (Continued)
Item Part Number Manufacturer Description Qty
R7 CRCW060311K5FKEA Vishay Dale(7) 11.5k Resistor, Size 0603, 1% 1
R8 CRCW06038K06FKEA Vishay Dale(7) 8.06k Resistor, Size 0603, 1% 1
R9 CRCW06034K75FKEA Vishay Dale(7) 4.75k Resistor, Size 0603, 1% 1
R10 CRCW06033K24FKEA Vishay Dale(7) 3.24k Resistor, Size 0603, 1% 1
R11 CRCW06031K91FKEA Vishay Dale(7) 1.91k Resistor, Size 0603, 1% 1
R12 CRCW060349K24FKEA Vishay Dale(7) 49.9 Resistor, Size 0603, 1% 1
R13, R21 CRCW06030000FKEA Vishay Dale(7) 0 Resistor, Size 0603, 5% 2
R14 CRCW08059K7FKEA Vishay Dale(7) 9.7k Resistor, Size 0805, 5% 1
U1 MIC2176-2YMM Micrel. Inc.(8) 75V Synchronous Buck DC-DC Regulator 1
Notes:
8. Micrel, Inc.: www.micrel.com.
Micrel, Inc. MIC2176
November 2010 26 M9999-111710-A
PCB Layout
Figure 8. MIC2176 Evaluation Board To p Layer
Micrel, Inc. MIC2176
November 2010 27 M9999-111710-A
PCB Layout (Continued)
Figure 9. MIC2176 Evaluation Board Mid-Layer 1 (Ground Plane)
Micrel, Inc. MIC2176
November 2010 28 M9999-111710-A
PCB Layout (Continued)
Figure 10. MIC2176 Evaluation Board Mid-L ayer 2
Micrel, Inc. MIC2176
November 2010 29 M9999-111710-A
PCB Layout (Continued)
Figure 11. MIC2176 Evaluation Board Bottom Layer
Micrel, Inc. MIC2176
November 2010 30 M9999-111710-A
Recommended Land Pattern
Micrel, Inc. MIC2176
November 2010 31 M9999-111710-A
Package Information
10-Pin MSOP (MM)
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