ADE7761AARSZ-RL - Analog Devices Inc.

Energy Metering IC with On-Chip Fault and

Missing Neutral Detection

ADE7761A

Rev. 0

Information furnished by Analog Devices is believed to be accurate and reliable. However, no

responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other

rights of third parties that may result from its use. Specifications subject to change without notice. No

license is granted by implication or otherwise under any patent or patent rights of Analog Devices.

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One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.

Tel: 781.329.4700 www.analog.com

FEATURES

High accuracy, active energy measurement IC supports

IEC 62053-21

Less than 0.1% error over a dynamic range of 500 to 1

Supplies active power on the frequency outputs, F1 and F2

High frequency output CF is intended for calibration and

supplies instantaneous active power

Continuous monitoring of the phase and neutral current

allows fault detection in 2-wire distribution systems

Current channels input level best suited for shunt and

current transformer sensors

Uses the larger of the two currents (phase or neutral) to

bill—even during a fault condition

Continuous monitoring of the voltage and current inputs

allows missing neutral detection

Uses one current input (phase or neutral) to bill when

missing neutral is detected

Two logic outputs (FAULT and REVP) can be used to indicate

a potential miswiring, fault, or missing neutral condition

Direct drive for electromechanical counters and 2-phase

stepper motors (F1 and F2)

Proprietary ADCs and DSP provide high accuracy over large

variations in environmental conditions and time

Reference 2.5 V ± 8% (drift 30 ppm/°C typical) with external

overdrive capability

Single 5 V supply, low power

GENERAL DESCRIPTION

The ADE7761A is a high accuracy, fault-tolerant, electrical

energy measurement IC intended for use with 2-wire distribution

systems. The part specifications surpass the accuracy requirements

as quoted in the IEC 62053-21 standard. The only analog circuitry

used on the ADE7761A is in the ADCs and reference circuit.

All other signal processing (such as multiplication and filtering)

is carried out in the digital domain. This approach provides

superior stability and accuracy over extremes in environmental

conditions and over time. The ADE7761A incorporates a fault

detection scheme similar to the ADE7751 by continuously

monitoring both phase and neutral currents. A fault is indicated

when the currents differ by more than 6.25%.

The ADE7761A incorporates a missing neutral detection

scheme by continuously monitoring the input voltage. When a

missing neutral condition is detected—no voltage input—the

ADE7761A continues billing based on the active current signal

(see the Missing Neutral Mode section). The missing neutral

condition is indicated when the FAULT pin goes high. The

ADE7761A supplies average active power information on the

low frequency outputs, F1 and F2. The CF logic output gives

instantaneous active power information.

The ADE7761A includes a power-supply monitoring circuit on

the VDD supply pin. Internal phase matching circuitry ensures

that the voltage and current channels are matched. An internal

no-load threshold ensures that the ADE7761A does not exhibit

any creep when there is no load.

FUNCTIONAL BLOCK DIAGRAM

MISCAL

ADC

AGND FAULT

2.5V

REFERENCE

INTERNAL

OSCILLATOR

MISSING NEUTRAL

GAIN ADJUST

MISSING NEUTRAL

DETECTION

914 17 10 11 12

B>A

A<>B

A>B

ZERO CROSSING

DETECTION

ADE7761A

SIGNAL PROCESSING

BLOCK

15 18

POWER

SUPPLY MONITOR

DIGITAL-TO-FREQUENCY CONVERTER

16 18 19 20

F1F2CFREVPS0S1SCFDGNDRCLKINREF

IN/OUT

HPF

LPF

3kΩ

05040-0-001

PGA

Figure 1.

ADE7761A

Rev. 0 | Page 2 of 24

TABLE OF CONTENTS

Features .............................................................................................. 1

General Description......................................................................... 1

Functional Block Diagram .............................................................. 1

Revision History ............................................................................... 2

Specifications..................................................................................... 3

Timing Characteristics ................................................................ 4

Absolute Maximum Ratings............................................................ 5

Performance Issues That May Affect Billing Accuracy........... 5

ESD Caution.................................................................................. 5

Pin Configuration and Function Descriptions............................. 6

Terminology ...................................................................................... 8

Typical Performance Characteristics ............................................. 9

Test Circuit ...................................................................................... 10

Operation......................................................................................... 11

Power Supply Monitor............................................................... 11

Analog Inputs ............................................................................. 11

Internal Oscillator ...................................................................... 12

Analog-to-Digital Conversion.................................................. 13

Active Power Calculation.......................................................... 14

Digital-to-Frequency Conversion............................................ 16

Transfer Function ....................................................................... 16

Fault Detection ........................................................................... 17

Missing Neutral Mode............................................................... 18

Applications..................................................................................... 21

Interfacing to a Microcontroller for Energy Measurement.. 21

Selecting a Frequency for an Energy Meter Application ...... 21

Negative Power Information..................................................... 22

Outline Dimensions....................................................................... 23

Ordering Guide .......................................................................... 23

REVISION HISTORY

7/06—Revision 0: Initial Version

ADE7761A

Rev. 0 | Page 3 of 24

SPECIFICATIONS

VDD = 5 V ± 5%, AGND = DGND = 0 V, on-chip reference, on-chip oscillator, TMIN to TMAX = −40°C to +85°C.

Table 1.

Parameter Value Unit Test Conditions/Comments

ACCURACY1

Measurement Error20.1 % of reading, typ Over a dynamic range of 500 to 1

Phase Error Between Channels

PF = 0.8 Capacitive ±0.05 Degrees, max Phase lead 37°

PF = 0.5 Inductive ±0.05 Degrees, max Phase lag 60°

AC Power Supply Rejection2

Output Frequency Variation 0.01 %, typ V1A = V1B = V2P = ±100 mV rms

DC Power Supply Rejection2

Output Frequency Variation 0.01 %, typ V1A = V1B = V2P = ±100 mV rms

FAULT DETEC TION2, 3 See the Fault Detection section

Fault Detection Threshold

Inactive Input <> Active Input 6.25 %, typ V1A or V1B active

Input Swap Threshold

Inactive Input <> Active Input 6.25 % of larger, typ V1A or V1B active

Accuracy Fault Mode Operation

V1A Active, V1B = AGND 0.1 % of reading, typ Over a dynamic range of 500 to 1

V1B Active, V1A = AGND 0.1 % of reading, typ Over a dynamic range of 500 to 1

Fault Detection Delay 3 Seconds, typ

Swap Delay 3 Seconds, typ

MISSING NEUTRAL MODE2, 4 See the Missing Neutral Detection section

Missing Neutral Detection Threshold

V2P − V2N 59.4 mV peak, min

Accuracy Missing Neutral Mode

V1A Active, V1B = V2P = AGND 0.1 % of reading, typ Over a dynamic range of 500 to 1

V1B Active, V1A = V2P = AGND 0.1 % of reading, typ Over a dynamic range of 500 to 1

Missing Neutral Detection Delay 3 Seconds, typ

ANALOG INPUTS V1A − V1N, V1B − V1N, V2P − V2N

Maximum Signal Levels ±660 mV peak, max Differential input

660 mV peak, max Differential input MISCAL − V2N

Input Impedance (DC) 400 kΩ, min

Bandwidth (−3 dB) 7 kHz, typ

ADC Offset Error215 mV, typ Uncalibrated error, see the Terminology section for details

Gain Error ±4 %, typ External 2.5 V reference

Gain Error Match2±3 %, typ External 2.5 V reference

REFERENCE INPUT

REFIN/OUT Input Voltage Range 2.7 V, max 2.5 V + 8%

2.3 V, min 2.5 V − 8%

Input Impedance 3 kΩ, min

Input Capacitance 10 pF, max

ON-CHIP REFERENCE

Reference Error ±200 mV, max

Temperature Coefficient 30 ppm/°C, typ

Current Source 20 µA, min

ON-CHIP OSCILLATOR

Oscillator Frequency 450 kHz

Oscillator Frequency Tolerance ±12 % of reading, typ

Temperature Coefficient 30 ppm/°C, typ

ADE7761A

Rev. 0 | Page 4 of 24

Parameter Value Unit Test Conditions/Comments

LOGIC INPUTS5

PGA, SCF, S1, and S0

Input High Voltage, VINH 2.4 V, min VDD = 5 V ± 5%

Input Low Voltage, VINL 0.8 V, max VDD = 5 V ± 5%

Input Current, IIN ±3 µA, max Typical 10 nA, VIN = 0 V to VDD

Input Capacitance, CIN 10 pF, max

LOGIC OUTPUTS5

CF, REVP, and FAULT

Output High Voltage, VOH 4 V, min VDD = 5 V ± 5%

Output Low Voltage, VOH 1 V, max VDD = 5 V ± 5%

F1 and F2

Output High Voltage, VOH 4 V, min VDD = 5 V ± 5%, ISOURCE = 10 mA

Output Low Voltage, VOH 1 V, max VDD = 5 V ± 5%, ISINK = 10 mA

POWER SUPPLY For specified performance

VDD 4.75 V, min 5 V − 5%

5.25 V, max 5 V + 5%

VDD 3 mA, max

1 See plots in the Typical Performance Characteristics section.

2 See the Terminology section for explanation of specifications.

3 See the Fault Detection section for explanation of fault detection functionality.

4 See the Missing Neutral Detection section for explanation of missing neutral detection functionality.

5 Sample tested during initial release and after any redesign or process change that might affect this parameter.

TIMING CHARACTERISTICS

VDD = 5 V ± 5%, AGND = DGND = 0 V, on-chip reference, on-chip oscillator, TMIN to TMAX = −40°C to +85°C. Sample tested during

initial release and after any redesign or process change that might affect this parameter. See Figure 2.

Table 2.

Parameter Value Unit Test Conditions/Comments

t11120 ms F1 and F2 Pulse Width (Logic High).

t2See Table 7 s Output Pulse Period. See the Transfer Function section.

t31/2 t2s Time Between F1 Falling Edge and F2 Falling Edge.

t4190 ms CF Pulse Width (Logic High).

t5See Table 8 s CF Pulse Period. See the Transfer Function section.

t6CLKIN/4 s Minimum Time Between F1 and F2 Pulse.

1 The pulse widths of F1, F2, and CF are not fixed for higher output frequencies. See the Transfer Function section.

05040-002

Figure 2. Timing Diagram for Frequency Outputs

ADE7761A

Rev. 0 | Page 5 of 24

ABSOLUTE MAXIMUM RATINGS

TA = 25°C, unless otherwise noted.

Table 3.

Parameter Rating

VDD to AGND −0.3 V to +7 V

Analog Input Voltage to AGND

V1A, V1B, V1N, V2N, V2P, MISCAL

−6 V to +6 V

Reference Input Voltage to AGND −0.3 V to VDD + 0.3 V

Digital Input Voltage to DGND −0.3 V to VDD + 0.3 V

Digital Output Voltage to DGND −0.3 V to VDD + 0.3 V

Operating Temperature Range

Industrial −40°C to +85°C

Storage Temperature Range −65°C to +150°C

Junction Temperature 150°C

20-Lead SSOP, Power Dissipation 450 mW

θJA Thermal Impedance 112°C/W

Lead Temperature, Soldering

Vapor Phase (60 sec) 215°C

Infrared (15 sec) 220°C

Stresses above those listed under Absolute Maximum Ratings

may cause permanent damage to the device. This is a stress

rating only; functional operation of the device at these or any

other conditions above those listed in the operational sections

of this specification is not implied. Exposure to absolute

maximum rating conditions for extended periods may affect

device reliability.

PERFORMANCE ISSUES THAT MAY AFFECT

BILLING ACCURACY

The ADE7761A provides pulse outputs—CF, F1, and F2—

intended to be used for the billing of active energy. Pulses

are generated at these outputs in two different situations.

Case 1: When the analog input V2P – V2N complies with the

conditions described in Figure 34, CF, F1, and F2 frequencies

are proportional to active power and can be used to bill

active energy.

Case 2: When the analog input V2P – V2N does not comply with

the conditions described in Figure 34, the ADE7761A does not

measure active energy but a quantity proportional to kAh. This

quantity is used to generate pulses on the same CF, F1, and F2.

This situation is indicated when the FAULT pin is high.

Analog Devices, Inc. cautions users of the ADE7761A about the

following:

• Billing active energy in Case 1 is consistent with the

understanding of the quantity represented by pulses on CF,

F1, and F2 outputs (watt-hour).

• Billing active energy while the ADE7761A is in Case 2 must

be decided knowing that the entity measured by the

ADE7761A in this case is ampere-hour and not watt-hour.

Users should be aware of this limitation and decide if the

ADE7761A is appropriate for their application.

ESD CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on

the human body and test equipment and can discharge without detection. Although this product features

proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy

electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance

degradation or loss of functionality.

ADE7761A

Rev. 0 | Page 6 of 24

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

REF

IN/OUT

SCF S1

DGND

REVP

FAULT

MISCAL RCLKIN

AGND PGA

813

912

10 11

ADE7761A

TOP VIEW

(Not to Scale)

05040-003

Figure 3. Pin Configuration (SSOP)

Table 4. Pin Function Descriptions

Pin No. Mnemonic Description

1 VDD Power Supply. This pin provides the supply voltage for the digital circuitry in the ADE7761A. The supply voltage

should be maintained at 5 V ± 5% for specified operation. This pin should be decoupled with a 10 µF capacitor

in parallel with a ceramic 100 nF capacitor.

2, 3 V1A, V1B Analog Inputs for Channel 1 (Current Channel). These inputs are fully differential voltage inputs with maximum

differential input signal levels of ±660 mV with respect to V1N for specified operation. The maximum signal level

at these pins is ±1 V with respect to AGND. Both inputs have internal ESD protection circuitry, and an overvoltage

of ±6 V can also be sustained on these inputs without risk of permanent damage.

4 V1N Negative Input for Differential Voltage Inputs, V1A and V1B. The maximum signal level at this pin is ±1 V with

respect to AGND. The input has internal ESD protection circuitry, and an overvoltage of ±6 V can also be sustained

on this input without risk of permanent damage. The input should be directly connected to the burden resistor

and held at a fixed potential, that is, AGND. See the Analog Inputs section.

5 V2N Negative Input for Differential Voltage Inputs, V2P and MISCAL. The maximum signal level at this pin is ±1 V with

respect to AGND. The input has internal ESD protection circuitry, and an overvoltage of ±6 V can also be sustained

on this input without risk of permanent damage. The input should be held at a fixed potential, that is, AGND. See

the Analog Inputs section.

6 V2P Analog Input for Channel 2 (Voltage Channel). This input is a fully differential voltage input with maximum

differential input signal levels of ±660 mV with respect to V2N for specified operation. The maximum signal level at

this pin is ±1 V with respect to AGND. This input has internal ESD protection circuitry, and an overvoltage of ±6 V

can also be sustained on this input without risk of permanent damage.

7 MISCAL

Analog Input for Missing Neutral Calibration. This pin can be used to calibrate the CF-F1-F2 frequencies in the

missing neutral condition. This input is a fully differential voltage input with maximum differential input signal

levels of 660 mV with respect to V2N for specified operation. The maximum signal level at this pin is ±1 V with

respect to AGND. This input has internal ESD protection circuitry, and an overvoltage of ±6 V can also be

sustained on this input without risk of permanent damage.

8 AGND

This pin provides the ground reference for the analog circuitry in the ADE7761A, that is, ADCs and reference. This

pin should be tied to the analog ground plane of the PCB. The analog ground plane is the ground reference for all

analog circuitry, such as antialiasing filters, and current and voltage transducers. For good noise suppression, the

analog ground plane should be connected only to the digital ground plane at the DGND pin.

9 REFIN/OUT This pin provides access to the on-chip voltage reference. The on-chip reference has a nominal value of

2.5 V ± 8% and a typical temperature coefficient of 30 ppm/°C. An external reference source can also be

connected at this pin. In either case, this pin should be decoupled to AGND with a 1 F ceramic capacitor and

100 nF ceramic capacitor.

10 SCF Select Calibration Frequency. This logic input is used to select the frequency on the calibration output CF.

Table 7 shows how the calibration frequencies are selected.

11, 12 S1, S0 These logic inputs are used to select one of four possible frequencies for the digital-to-frequency conversion.

This offers the designer greater flexibility when designing the energy meter. See the Selecting a Frequency for an

Energy Meter Application section.

13 PGA This logic input is used to select the gain for the analog inputs, V1A and V1B. The possible gains are 1 and 16.

14 RCLKIN

To enable the internal oscillator as a clock source on the chip, a precise low temperature drift resistor at

nominal value of 6.2 kΩ must be connected from this pin to DGND.

ADE7761A

Rev. 0 | Page 7 of 24

Pin No. Mnemonic Description

15 FAULT This logic output goes active high when a fault or missing neutral condition occurs. A fault is defined as a

condition under which the signals on V1A and V1B differ by more than 6.25%. A missing neutral condition is

defined when the chip is powered up with no voltage at the input. The logic output is reset to zero when a fault

or missing neutral condition is no longer detected. See the Fault Detection section and the Missing Neutral Mode

section.

16 REVP This logic output goes logic high when negative power is detected, that is, when the phase angle between the

voltage and current signals is greater than 90°. This output is not latched and is reset when positive power is once

again detected. The output goes high or low at the same time as a pulse is issued on CF.

17 DGND This pin provides the ground reference for the digital circuitry in the ADE7761A, that is, multiplier, filters, and

digital-to-frequency converters. This pin should be tied to the digital ground plane of the PCB. The digital ground

plane is the ground reference for all digital circuitry, such as counters (mechanical and digital), MCUs, and

indicator LEDs. For good noise suppression, the analog ground plane should be connected only to the digital

ground plane at the DGND pin.

18 CF Calibration Frequency Logic Output. The CF logic output, active high, gives instantaneous active power

information. This output is used for operational and calibration purposes. See the Digital-to-Frequency Conversion

section.

19, 20 F2, F1 Low Frequency Logic Outputs. F1 and F2 supply average active power information. The logic outputs can be

used to directly drive electromechanical counters and 2-phase stepper motors.

ADE7761A

Rev. 0 | Page 8 of 24

TERMINOLOGY

Measurement Error

The error associated with the energy measurement made by the

ADE7761A is defined by

⎟

⎠

⎞

⎜

⎝

⎛×

−

%100

EnergyTrue

EnergyTrueADE7761AbyregisteredEnergy

Erro

Percentag

Phase Error Between Channels

The high-pass filter (HPF) in the current channel has a phase

lead response. To offset this phase response and equalize the

phase response among channels, a phase correction network is

also placed in the current channel. The phase correction network

ensures a phase match between the current channels and

voltage channels to within ±0.1° over a range of 45 Hz to

65 Hz and ±0.2° over a range of 40 Hz to 1 kHz.

Power Supply Rejection (PSR)

PSR quantifies the ADE7761A measurement error as a

percentage of reading when the power supplies are varied. For

the ac PSR measurement, a reading at nominal supplies (5 V) is

taken. A second reading is obtained with the same input signal

levels when an ac (175 mV rms/100 Hz) signal is introduced

onto the supplies. Any error introduced by this ac signal is

expressed as a percentage of reading (see the Measurement

Error definition).

For the dc PSR measurement, a reading at nominal supplies

(5 V) is taken. A second reading is obtained with the same input

signal levels when the power supplies are varied ±5%. Any error

introduced is again expressed as a percentage of reading.

ADC Offset Error

The dc offset associated with the analog inputs to the ADCs.

With the analog inputs connected to AGND, the ADCs still see

a dc analog input signal. The magnitude of the offset depends on

the input gain and range selection (see the Typical Performance

Characteristics section). However, when HPFs are switched on,

the offset is removed from the current channels and the power

calculation is not affected by this offset.

Gain Error

The gain error in the ADE7761A ADCs is defined as the

difference between the measured output frequency (minus the

offset) and the ideal output frequency. It is measured with a

gain of 1 in Channel V1A. The difference is expressed as a

percentage of the ideal frequency, which is obtained from the

transfer function (see the Transfer Function section).

Gain Error Match

The gain error match is defined as the gain error (minus the

offset) obtained when switching between a gain of 1 or 16. It is

expressed as a percentage of the output ADC code obtained

under a gain of 1.

ADE7761A

Rev. 0 | Page 9 of 24

TYPICAL PERFORMANCE CHARACTERISTICS

05040-004

CURRENT (% of Full Scale)

100.00.1 1.0 10.0

% ERROR

–1.0

1.0

0.8

0.6

0.4

0.2

–0.2

–0.4

–0.6

–0.8

PF = 1

ON-CHIP REFERENCE

–40°C

+25°C

+85°C

Figure 4. Active Power Error as a Percentage of Reading

with Gain = 1 and Internal Reference

05040-005

CURRENT (% of Full Scale)

1000.1 1.0 10

% ERROR

–1.0

1.5

1.0

0.5

–0.5

PF = DIFFERENT VALUES

ON-CHIP REFERENCE

–40°C; PF = 0.5

+25°C; PF = 1

+25°C; PF = 0.5

+85°C; PF = 0.5

Figure 5. Active Power Error as a Percentage of Reading

over Power Factor with Gain = 1 and Internal Reference

–1.0

–0.8

–0.6

–0.4

–0.2

0.2

0.4

0.10 1 10 100

CURRENT (% Full Scale)

ERRO

05040-034

+25°C

+85°C

–40°C

0.6

0.8

1.0

PF = 1. GAIN = 16

ON-CHIP REFERENCE

Figure 6. Active Power Error as a Percentage of Reading

with Gain = 16 and Internal Reference

–0.2

0.2

0.4

0.10 1 10 100

CURRENT (% Full Scale)

ERRO

0.6

GAIN = 16

ON-CHIP REFERENCE

05040-035

PF = –0.5

PF = +1

PF = +0.5

1.0

0.8

–0.8

–0.6

–0.4

–1.0

Figure 7. Active Power Error as a Percentage of Reading

over Power Factor with Gain = 16 and Internal Reference

05040-006

CURRENT (% of Full Scale)

100.00.1 1.0 10.0

% ERROR

–1.0

1.0

0.8

0.6

0.4

0.2

–0.2

–0.4

–0.6

–0.8

PF = 1

ON-CHIP REFERENCE

5.25V

5.00V

4.75V

Figure 8. Active Power Error as a Percentage of Reading

over Power Supply with Gain = 1 and Internal Reference

05040-007

CURRENT (% of Full Scale)

100.00.1 1.0 10.0

% ERROR

–2.0

2.0

1.5

1.0

0.5

–0.5

–1.0

–1.5

ON-CHIP REFERENCE

+85°C

+25°C

–40°C

Figure 9. Ampere Hour Error as a Percentage of Reading

in Missing Neutral Mode with Gain = 1 and Internal Reference

ADE7761A

Rev. 0 | Page 10 of 24

TEST CIRCUIT

VDD

REFIN/OUT

100nF

10µF

AGND DGNDPGA

FAULT

RCLKIN

SCF

PS2501-1

ADE7761A

TO FREQ.

COUNTER

V1A

V1B

V1N

100nF 10µF

V2N

33nF

1kΩ

6.2kΩ

10kΩ

40A TO 80mA

RB = 18Ω

1713 8

2kΩ

V2P

MISCAL

33nF1kΩ

1MΩ

220

100kΩ

560kΩ

33nF

1kΩ

RB 33nF

1kΩ

RB 33nF

1kΩ

05040-008

Figure 10. Test Circuit for Performance Curves

ADE7761A

Rev. 0 | Page 11 of 24

OPERATION

POWER SUPPLY MONITOR

The ADE7761A continuously monitors the power supply (VDD)

with its on-chip, power supply monitor. If the supply is less than

4 V ± 5%, the ADE7761A goes into an inactive state, that is, no

energy is accumulated and the CF, F1, and F2 outputs are

disabled. This is useful to ensure correct device operation at

power-up and during power-down. The power supply monitor

has built-in hysteresis and filtering, which provides a high

degree of immunity to false triggering due to noisy supplies.

The power supply and decoupling for the part should be such

that the ripple at VDD does not exceed 5 V ± 5% as specified for

normal operation.

ADE7761A

REVP - FAULT - CF -

F1 - F2 OUTPUTS

INACTIVE ACTIVE

TIME

INACTIVE

05040-009

Figure 11. On-Chip, Power Supply Monitoring

ANALOG INPUTS

Channel V1 (Current Channel)

The voltage outputs from the current transducers are connected

to the ADE7761A at Channel V1. It has two voltage inputs, V1A

and V1B. These inputs are fully differential with respect to V1N.

However, at any one time, only one is selected to perform the

power calculation (see the Fault Detection section).

The maximum peak differential signal on V1A − V1N and V1B − V1N

is ±660 mV. However, Channel 1 has a programmable gain

amplifier (PGA) with user-selectable gains of 1 or 16 (see

Table 5). This gain facilitates easy transducer interfacing.

Table 5. Channel 1 Dynamic Range

PGA Gain Maximum Differential Signal (mV)

0 1 660

1 16 41

Figure 12 shows the maximum signal levels on V1A, V1B, and

V1N. The maximum differential voltage is ±660 mV divided by

the gain selection. The differential voltage signal on the inputs

must be referenced to a common mode (usually AGND).

DIFFERENTIAL INPUT B

±660mV MAX PEAK V

COMMON MODE

±100mV MAX

DIFFERENTIAL INPUT A

±660mV MAX PEAK

AGND

, V

05040-010

+660mV

GAIN

–660mV

GAIN

+ V

Figure 12. Maximum Signal Levels, Channel 1

Channel V2 (Voltage Channel)

The output of the line voltage transducer is connected to the

ADE7761A at this analog input. Channel V2 is a single-ended,

voltage input. The maximum peak differential signal on Channel 2

is ±660 mV with respect to V2N. Figure 13 shows the maximum

signal levels that can be connected to Channel 2.

05040-011

+660mV + V

–660mV + V

COMMON MODE

±100mV MAX

DIFFERENTIAL INPUT

±660mV MAX PEAK

Figure 13. Maximum Signal Levels, Channel 2

The differential voltage V2P − V2N must be referenced to a

common mode (usually AGND). The analog inputs of the

ADE7761A can be driven with common-mode voltages of up

to 100 mV with respect to AGND. However, the best results

are achieved using a common mode equal to AGND.

MISCAL Input

The input for the power calibration in missing neutral mode

is connected to the ADE7761A at this analog input. MISCAL is

a single-ended, voltage input. It is recommended to use a dc

signal derived from the voltage reference to drive this pin. The

maximum peak differential signal on MISCAL is 660 mV with

respect to V2N. Figure 14 shows the maximum signal levels that

can be connected to the MISCAL pin.

05040-012

+660mV + V

COMMON MODE

±100mV MAX

MISCAL

DIFFERENTIAL INPUT

±660mV MAX PEAK

MISCAL

AGND

Figure 14. Maximum Signal Levels, MISCAL

ADE7761A

Rev. 0 | Page 12 of 24

The differential voltage MISCAL − V2N must be referenced

to a common mode (usually AGND). The analog inputs of the

ADE7761A can be driven with common-mode voltages of up

to 100 mV with respect to AGND. However, best results are

achieved using a common mode equal to AGND.

Typical Connection Diagrams

Figure 15 shows a typical connection diagram for Channel V1.

The analog inputs are used to monitor both the phase and

neutral currents. Because of the large potential difference

between the phase and neutral, two current transformers (CTs)

must be used to provide the isolation. Note that both CTs are

referenced to analog ground (AGND); therefore, the common-

mode voltage is 0 V. The CT turns ratio and burden resistor

(RB) are selected to give a peak differential voltage of

±660 mV/gain.

AGND

INIP

PHASE

NEUTRAL

05040-013

±660mV

GAIN

±660mV

GAIN

Figure 15. Typical Connection for Channel 1

Figure 16 shows two typical connections for Channel V2.

The first option uses a potential transformer (PT) to provide

complete isolation from the main voltage. In the second option,

the ADE7761A is biased around the neutral wire, and a resistor

divider is used to provide a voltage signal that is proportional to

the line voltage. Adjusting the ratio of RA and RB + VR is a

convenient way to carry out a gain calibration on the meter.

05040-014

RB + VR = RF.

NEUTRAL

PHASE

NEUTRAL

PHASE

±660mV

AGND

Figure 16. Typical Connection for Channel 2

Figure 17 shows a typical connection for the MISCAL input.

The voltage reference input (REFIN/OUT) is used as a dc reference

to set the MISCAL voltage.

Adjusting the level of MISCAL to calibrate the meter in missing

neutral mode can be done by changing the ratio of RC and RD

+ VR1. When the internal reference is used, the values of RC,

RD, and VR1 must be chosen to limit the current sourced by

the internal reference sourcing current to below the specified

20 μA. Therefore, because VREF internal = 2.5 V, RC + RD +

VR1 > 600 kΩ.

05040-015

VR1

MISCAL

REF

IN/OUT

Figure 17. Typical Connection for MISCAL

INTERNAL OSCILLATOR

The nominal internal oscillator frequency is 450 kHz when

used with the recommended ROSC resistor value of 6.2 kΩ

between RCLKIN and DGND (see Figure 18).

The internal oscillator frequency is inversely proportional to the

value of this resistor. Although the internal oscillator operates

when used with an ROSC resistor value between 5 kΩ and 12 kΩ,

it is recommended to choose a value within the range of the

nominal value.

The output frequencies on CF, F1, and F2 are directly propor-

tional to the internal oscillator frequency; therefore, the resistor

ROSC must have a low tolerance and low temperature drift. A low

tolerance resistor limits the variation of the internal oscillator

frequency. A small variation of the clock frequency and

consequently of the output frequencies from meter to meter

contributes to a smaller calibration range of the meter.

A low temperature drift resistor directly limits the variation of

the internal clock frequency over temperature. The stability of

the meter to external variation is then better ensured by design.

2.5V

REFERENCE

INTERNAL

OSCILLATOR

DE7761

DGNDRCLKINREF

IN/OUT

3kΩ

05040-016

OSC

14 17

Figure 18. Internal Oscillator Connection

ADE7761A

Rev. 0 | Page 13 of 24

ANALOG-TO-DIGITAL CONVERSION

The analog-to-digital conversion in the ADE7761A is carried

out using second-order, Σ-Δ ADCs. Figure 19 shows a first-

order, Σ-Δ ADC (for simplicity). The converter is made up of

two parts: the Σ-Δ modulator and the digital low-pass filter.

04407-017

....10100101....

1-BIT DAC

LATCHED

COMPAR-

ATOR

INTEGRATOR

REF

MCLK

ANALOG

LOW-PASS FILTER DIGITAL

LOW-PASS FILTER

124

Figure 19. First-Order, Σ-Δ ADC

A Σ-Δ modulator converts the input signal into a continuous

serial stream of 1s and 0s at a rate determined by the sampling

clock. In the ADE7761A, the sampling clock is equal to CLKIN.

The 1-bit DAC in the feedback loop is driven by the serial data

stream. The DAC output is subtracted from the input signal. If

the loop gain is high enough, the average value of the DAC

output (and, therefore, the bit stream) approaches that of the

input signal level. For any given input value in a single sampling

interval, the data from the 1-bit ADC is virtually meaningless.

Only when a large number of samples are averaged is a meaningful

result obtained. This averaging is carried out in the second part

of the ADC, the digital low-pass filter. By averaging a large

number of bits from the modulator, the low-pass filter can

produce 24-bit data-words that are proportional to the input

signal level.

The Σ-Δ converter uses two techniques to achieve high

resolution from what is essentially a 1-bit conversion technique.

The first is oversampling, which means that the signal is sampled at

a rate (frequency) that is many times higher than the bandwidth

of interest. For example, the sampling rate in the ADE7761A is

CLKIN (450 kHz) and the band of interest is 40 Hz to 1 kHz.

Oversampling has the effect of spreading the quantization noise

(noise due to sampling) over a wider bandwidth. With the noise

spread more thinly over a wider bandwidth, the quantization

noise in the band of interest is lowered (see Figure 20).

However, oversampling alone is not an efficient enough method

to improve the signal-to-noise ratio (SNR) in the band of interest.

For example, an oversampling ratio of 4 is required just to

increase the SNR by only 6 dB (1 bit). To keep the oversampling

ratio at a reasonable level, it is possible to shape the quantization

noise so that the majority of the noise lies at the higher frequencies.

This is what happens in the Σ-Δ modulator; the noise is shaped

by the integrator, which has a high-pass type response for the

quantization noise. The result is that most of the noise is at the

higher frequencies, where it can be removed by the digital low-

pass filter. This noise shaping is also shown in Figure 20.

SHAPED NOISE

HIGH RESOLUTION

OUTPUT FROM

DIGITAL LFP

NOISE

SIGN

NOISE

SIGN

0 1 225 450

FREQUENCY (kHz)

0 1 225 450

FREQUENCY (kHz)

DIGITAL F ILTER

NTIALIAS FILTER (RC)

SAMPLING FREQUENCY

05040-018

Figure 20. Noise Reduction due to Oversampling and

Noise Shaping in the Analog Modulator

Antialias Filter

Figure 20 also shows an analog low-pass filter (RC) on input to

the modulator. This filter is present to prevent aliasing. Aliasing

is an artifact of all sampled systems, which means that frequency

components in the input signal to the ADC that are higher than

half the sampling rate of the ADC appear in the sampled signal

frequency below half the sampling rate. Figure 21 illustrates

the effect.

In Figure 21, frequency components (arrows shown in black)

above half the sampling frequency (also known as the Nyquist

frequency), that is, 225 kHz, are imaged or folded back down

below 225 kHz (arrows shown in gray). This happens with all

ADCs no matter what the architecture. In the example shown,

only frequencies near the sampling frequency (450 kHz) move

into the band of interest for metering (40 Hz to 1 kHz). This

fact allows the use of a very simple low-pass filter to attenuate

these frequencies (near 250 kHz) and thereby prevent distortion

in the band of interest. A simple RC filter (single pole) with a

corner frequency of 10 kHz produces an attenuation of

approximately 33 dB at 450 kHz (see Figure 21). This is

sufficient to eliminate the effects of aliasing.

05040-019

0 1 225 450

FREQUENCY (kHz)

IMAGE

FREQUENCIES

SAMPLING

FREQUENCY

NTIALIASING EFFECTS

Figure 21. ADC and Signal Processing in Current Channel or Voltage Channel

ADE7761A

Rev. 0 | Page 14 of 24

ACTIVE POWER CALCULATION

The ADCs digitize the voltage signals from the current and

voltage transducers. A high-pass filter in the current channel

removes any dc component from the current signal. This

eliminates any inaccuracies in the active power calculation

due to offsets in the voltage or current signals (see the HPF and

Offset Effects section).

The active power calculation is derived from the instantaneous

power signal. The instantaneous power signal is generated by a

direct multiplication of the current and voltage signals. To

extract the active power component (dc component), the

instantaneous power signal is low-pass filtered. Figure 22

illustrates the instantaneous active power signal and shows how

the active power information can be extracted by low-pass

filtering the instantaneous power signal. This scheme correctly

calculates active power for nonsinusoidal current and voltage

waveforms at all power factors. All signal processing is carried

out in the digital domain for superior stability over temperature

and time.

05040-020

DIGITAL-TO-

FREQUENCY

DIGITAL-TO-

FREQUENCY

HPF

MULTIPLIER LPF

ADC

CH1

CH2

INSTANTANEOUS

POWER SIGNAL –p(t)

INSTANTANEOUS

ACTIVE POWER SIGNAL

V × I

TIME

p(t) = i(t).v(t)

WHERE:

v(t) = V × cos(ωt)

i(t) = I × cos(ωt)

p(t) = V × I {1 + cos (2ωt)}

PGA

Figure 22. Signal Processing Block Diagram

The low frequency output of the ADE7761A is generated by

accumulating this active power information. This low frequency

inherently means a long accumulation time between output

pulses. The output frequency is, therefore, proportional to the

average active power. This average active power information

can, in turn, be accumulated (for example, by a counter) to

generate active energy information. Because of its high output

frequency and, therefore, shorter integration time, the CF

output is proportional to the instantaneous active power. This is

useful for system calibration purposes that take place under

steady load conditions.

Power Factor Considerations

The method used to extract the active power information from

the instantaneous power signal (by low-pass filtering) is still valid

even when the voltage and current signals are not in phase.

Figure 23 displays the unity power factor condition and a

displacement power factor (DPF = 0.5), that is, current signal

lagging the voltage by 60°.

INST

NEOUS

POWER SIGNAL

INSTANT

NEOUS

ACTIVE POWER SIGNAL

INSTANTANEOUS

POWER SIGNAL

INSTANTANEOUS

ACTIVE POWER SIGNAL

60°

CURRENT

VOLTAGE

05040-021

V × I

2× cos(60°)

Figure 23. Active Power Calculation over PF

If one assumes that the voltage and current waveforms are

sinusoidal, the active power component of the instantaneous

power signal (dc term) is given by

(V × I/2) × cos(60°)

This is the correct active power calculation.

Nonsinusoidal Voltage and Current

The active power calculation method also holds true for

nonsinusoidal current and voltage waveforms. All voltage

and current waveforms in practical applications have some

harmonic content. Using the Fourier transform, instantaneous

voltage and current waveforms can be expressed in terms of

their harmonic content

)sin(2)(

OthVVtv α+ω××+= ∑

∞

≠

(1)

where:

v(t) is the instantaneous voltage.

VO is the average value.

Vh is the rms value of voltage harmonic h.

αh is the phase angle of the voltage harmonic.

ADE7761A

Rev. 0 | Page 15 of 24

)sin(2)(

OthIIti β+ω××+= ∑

∞

≠

(2)

where:

i(t) is the instantaneous current.

IO is the dc component.

Ih is the rms value of current harmonic h.

βh is the phase angle of the current harmonic.

Using Equation 1 and Equation 2, the active power P can be

expressed in terms of its fundamental active power (P1) and

harmonic active power (PH).

P = P1 + PH

where:

P1 = V1 × I1 cos(Φ1)

Φ1 = α1 − β1 (3)

and

)cos(

∑

∞

Φ××=

hhhh

HIVP

hhh

−α=Φ (4)

As can be seen in Equation 4, a harmonic active power component

is generated for every harmonic provided that the harmonic is

present in both the voltage and current waveforms. The power

factor calculation was previously shown to be accurate in the

case of a pure sinusoid; therefore, the harmonic active power

must also correctly account for the power factor because it is

made up of a series of pure sinusoids.

Note that the input bandwidth of the analog inputs is 7 kHz

with an internal oscillator frequency of 450 kHz.

HPF and Offset Effects

Equation 5 shows the effect of offsets on the active power

calculation. Figure 24 shows the effect of offsets on the active

power calculation in the frequency domain.

)cos()cos(

))cos(())cos((

)()(

tIVtIV

tIItVV

tItV

0110

1010

ω××+ω××+

+×

=ω×+×ω×+

=×

(5)

As can be seen in Equation 5 and Figure 24, an offset on Channel 1

and Channel 2 contributes a dc component after multiplication.

Because this dc component is extracted by the LPF and used to

generate the active power information, the offsets contribute a

constant error to the active power calculation. This problem is

easily avoided in the ADE7761A with the HPF in Channel 1. By

removing the offset from at least one channel, no error component

can be generated at dc by the multiplication. Error terms at cos(ωt)

are removed by the LPF and the digital-to-frequency conversion

(see the Digital-to-Frequency Conversion section).

The HPF in Channel 1 has an associated phase response that is

compensated for on-chip. Figure 25 and Figure 26 show the

phase error between channels with the compensation network

activated. The ADE7761A is phase compensated up to 1 kHz as

shown, which ensures a correct active harmonic power calculation

even at low power factors.

05040-022

DC COMPONENT (INCLUDING ERROR TERM)

IS EXTRACTED BY THE LPF FOR ACTIVE

POWER CALCULATION

2ω

FREQUENCY (RAD/S)

0ϖ

× I

Figure 24. Effect of Channel Offsets on the Active Power Calculation

FREQUENCY (Hz)

0 100

PHASE (Degrees)

–0.05

–0.10

0.05

0.10

0.15

0.20

0.25

0.30

200 300 400 500 600 700 800 900 1000

05040-023

Figure 25. Phase Error Between Channels (0 Hz to 1 kHz)

FREQUENCY (Hz)

PHASE (Degrees)

–0.05

–0.10

0.05

0.10

0.15

0.20

0.25

0.30

45 50 55 60 65 70

05040-024

Figure 26. Phase Error Between Channels (40 Hz to 70 Hz)

ADE7761A

Rev. 0 | Page 16 of 24

DIGITAL-TO-FREQUENCY CONVERSION

As previously described, the digital output of the low-pass filter

after multiplication contains the active power information.

However, because this LPF is not an ideal brick wall filter

implementation, the output signal also contains attenuated

components at the line frequency and its harmonics, that is,

cos(hωt), where h = 1, 2, 3, …, and so on. The magnitude

response of the filter is given by

)Hz5.4/(1

)(

fH =

= (6)

For a line frequency of 50 Hz, this gives an attenuation of the 2ω

(100 Hz) component of approximately −26.9 dB. The dominating

harmonic is at twice the line frequency, cos(2ωt), due to the

instantaneous power signal.

DIGITAL-TO-

FREQUENCY

DIGITAL-TO-

FREQUENCY

MULTIPLIER LPF

0ω2ω

FREQUENCY (Rad/s)

LPF TO EXTRACT

ACTIVE POWER

(DC TERM)

TIME

05040-025

FREQUENCY FREQUENCY

FOUT

INSTANTANEOUS ACTIVE POWER SIGNAL (FREQUENCY DOMAIN)

Figure 27. Active Power to Frequency Conversion

Figure 27 shows the instantaneous active power signal output of

the LPF, which still contains a significant amount of instantaneous

power information, cos(2ωt). This signal is then passed to the

digital-to-frequency converter, where it is integrated (accumulated)

over time to produce an output frequency. This accumulation of

the signal suppresses or averages out any non-dc components in

the instantaneous active power signal. The average value of a

sinusoidal signal is zero. Therefore, the frequency generated by

the ADE7761A is proportional to the average active power.

Figure 27 also shows the digital-to-frequency conversion for

steady load conditions: constant voltage and current. As can be

seen in Figure 27, the frequency output CF varies over time,

even under steady load conditions. This frequency variation is

primarily due to the cos(2ωt) component in the instantaneous

active power signal.

The output frequency on CF can be up to 2048 times higher

than the frequency on F1 and F2. This higher output frequency

is generated by accumulating the instantaneous active power

signal over a much shorter time while converting it to a frequency.

This shorter accumulation period means less averaging of the

cos(2ωt) component. As a consequence, some of this instantaneous

power signal passes through the digital-to-frequency conversion.

This is not a problem in the application.

Where CF is used for calibration purposes, the frequency

should be averaged by the frequency counter, which removes

any ripple. If CF is being used to measure energy, such as in a

microprocessor-based application, the CF output should also be

averaged to calculate power. Because the outputs, F1 and F2,

operate at a much lower frequency, a lot more averaging of the

instantaneous active power signal is carried out. The result is a

greatly attenuated sinusoidal content and a virtually ripple-free

frequency output.

TRANSFER FUNCTION

Frequency Outputs F1 and F2

The ADE7761A calculates the product of two voltage signals

(on Channel 1 and Channel 2) and then low-pass filters this

product to extract active power information. This active power

information is then converted to a frequency. The frequency

information is output on F1 and F2 in the form of active high

pulses. The pulse rate at these outputs is relatively low, for

example, 0.34 Hz maximum for ac signals with S0 = S1 = 0

(see Table 8). This means that the frequency at these outputs is

generated from active power information accumulated over a

relatively long period. The result is an output frequency that is

proportional to the average active power. The averaging of the

active power signal is implicit to the digital-to-frequency

conversion. The output frequency or pulse rate is related to

the input voltage signals by

70.5

REF

41rmsrms

FV2V1Gain

FrequencyFF −

××

=− (7)

where:

F1 − F2 Frequency is the output frequency on F1 and F2 (Hz).

V1rms is the differential rms voltage signal on Channel 1 (V).

V2rms is the differential rms voltage signal on Channel 2 (V).

Gain is 1 or 16, depending on the PGA gain selection made

using the logic input PGA.

VREF is the reference voltage (2.5 V ± 8%) (V).

F1–4 is one of four possible frequencies selected by using the

logic inputs S0 and S1 (see Table 6).

ADE7761A

Rev. 0 | Page 17 of 24

Table 6. F1–4 Frequency Selection

S1 S0 F1–4 (Hz)1F1−4 = OSC/2n 2

0 0 1.72 OSC/218

0 1 3.44 OSC/217

1 0 6.86 OSC/216

1 1 13.7 OSC/215

1 Values are generated using the nominal frequency of 450 kHz.

2 F1–4 are a binary fraction of the master clock and, therefore, vary with the

internal oscillator frequency (OSC).

Frequency Output CF

The pulse output calibration frequency (CF) is intended for use

during calibration. The output pulse rate on CF can be up to

2048 times the pulse rate on F1 and F2. The lower the F1–4

frequency selected, the higher the CF scaling. Table 7 shows

how the two frequencies are related, depending on the states of

the logic inputs S0, S1, and SCF. Because of its relatively high

pulse rate, the frequency at this logic output is proportional to

the instantaneous active power. As with F1 and F2, the

frequency is derived from the output of the low-pass filter after

multiplication. However, because the output frequency is high,

this active power information is accumulated over a much

shorter time. Therefore, less averaging is carried out in the

digital-to-frequency conversion. With much less averaging of

the active power signal, the CF output is much more responsive

to power fluctuations (see Figure 22).

Table 7. Relationship Between CF and F1, F2 Frequency

Outputs

SCF S1 S0 F1–4 (Hz) CF Frequency Output

1 0 0 1.72 128 × F1, F2

0 0 0 1.72 64 × F1, F2

1 0 1 3.44 64 × F1, F2

0 0 1 3.44 32 × F1, F2

1 1 0 6.86 32 × F1, F2

0 1 0 6.86 16 × F1, F2

1 1 1 13.7 16 × F1, F2

0 1 1 13.7 2048 × F1, F2

Example

In this example, if ac voltages of ±660 mV peak are applied to

V1 and V2, then the expected output frequency on CF, F1, and

F2 is calculated as

Gain = 1, PGA = 0

F1–4 = 1.7 Hz, SCF = S1 = S0 = 0

V1rms = rms of 660 mV peak ac = 0.66/√2 V

V2rms = rms of 660 mV peak ac = 0.66/√2 V

VREF = 2.5 V (nominal reference value)

Note that if the on-chip reference is used, actual output

frequencies can vary from device to device due to a reference

tolerance of ±8%.

Hz34.0

5.222

Hz72.166.066.070.5

21 =

××

×××

=− FrequencyFF

CF Frequency = F1 − F2 × 64 = 22.0 Hz

As can be seen from these two example calculations, the

maximum output frequency for ac inputs is always half of that

for dc input signals. Table 8 shows a complete listing of all

maximum output frequencies for ac signals.

Table 8. Maximum Output Frequencies on CF, F1, and F2 for

AC Inputs

SCF S1 S0

F1, F2 Maximum

Frequency (Hz)

CF Maximum

Frequency (Hz)

CF-to-

F1 Ratio

1 0 0 0.34 43.52 128

0 0 0 0.34 21.76 64

1 0 1 0.68 43.52 64

0 0 1 0.68 21.76 32

1 1 0 1.36 43.52 32

0 1 0 1.36 21.76 16

1 1 1 2.72 43.52 16

0 1 1 2.72 5570 2048

FAULT DETECTION

The ADE7761A incorporates a novel fault detection scheme

that warns of fault conditions and allows the ADE7761A to

continue accurate billing during a fault event. The ADE7761A

does this by continuously monitoring both the phase and

neutral (return) currents. A fault is indicated when these

currents differ by more than 6.25%. However, even during a

fault, the output pulse rate on F1 and F2 is generated using the

larger of the two currents. Because the ADE7761A looks for a

difference between the voltage signals on V1A and V1B, it is

important that both current transducers be closely matched.

On power-up, the output pulse rate of the ADE7761A is

proportional to the product of the voltage signals on V1A and

Channel 2. If the difference between V1A and V1B on power-up is

greater than 6.25%, the fault indicator (FAULT) becomes active

after about 1 sec. In addition, if V1B is greater than V1A, the

ADE7761A selects V1B as the input. The fault detection is

automatically disabled when the voltage signal on Channel 1 is

less than 0.3% of the full-scale input range. This eliminates false

detection of a fault due to noise at light loads.

ADE7761A

Rev. 0 | Page 18 of 24

Fault with Active Input Greater than Inactive Input

If V1A is the active current input (that is, being used for billing),

and the voltage signal on V1B (inactive input) falls below 93.75%

of V1A, the fault indicator becomes active. Both analog inputs

are filtered and averaged to prevent false triggering of this logic

output. As a consequence of the filtering, there is a time delay of

approximately 3 sec on the logic output FAULT after the fault

event. The FAULT logic output is independent of any activity on

outputs F1 or F2. Figure 28 shows one condition under which

FAULT becomes active. Because V1A is the active input and it is

still greater than V1B, billing is maintained on V1A, that is, no

swap to the V1B input occurs. V1A remains the active input.

AGND

FILTER

AND

COMPARE

MULTIPLIER

FAULT

< 93.75% OF V

<0 ACTIVE POINT – INACTIVE INPUT

6.25% OF ACTIVE INPUT

05040-026

FAULT

Figure 28. Fault Conditions for Active Input Greater than Inactive Input

Fault with Inactive Input Greater than Active Input

Figure 29 illustrates another fault condition. If the difference

between V1B, the inactive input, and V1A, the active input (that

is, being used for billing), becomes greater than 6.25% of V1B,

the FAULT indicator becomes active and a swap over to the V1B

input occurs. The analog input V1B becomes the active input.

Again, a time constant of about 3 sec is associated with this

swap. V1A does not swap back to the active channel until V1A is

greater than V1B and the difference between V1A and V1B—in

this order—becomes greater than 6.25% of V1A. However, the

FAULT indicator becomes inactive as soon as V1A is within

6.25% of V1B. This threshold eliminates potential chatter

between V1A and V1B.

AGND

FILTER

AND

COMPARE

MULTIPLIER

FAULT

< 93.75% OF V

<0 ACTIVE POINT – INACTIVE INPUT

6.25% OF INACTIVE INPUT

05040-027

FAULT + SWAP

Figure 29. Fault Conditions for Inactive Input Greater than Active Input

Calibration Concerns

Typically, when a meter is being calibrated, the voltage and

current circuits are separated, as shown in Figure 30. This

means that current passes through only the phase or neutral

circuit. Figure 30 shows current being passed through the phase

circuit. This is the preferred option because the ADE7761A

starts billing on the input V1A on power-up. The phase circuit

CT is connected to V1A in Figure 30. Because there is no current

in the neutral circuit, the FAULT indicator comes on under

these conditions. However, this does not affect the accuracy of

the calibration and can be used as a means to test the functionality

of the fault detection.

AGND

RB 0V

PHASE

NEUTRAL

RB + VR = RF.

TEST

CURRENT

240V rms

05040-028

Figure 30. Conditions for Calibration of Channel B

If the neutral circuit is chosen for the current circuit in the

arrangement shown in Figure 30, this may have implications for

the calibration accuracy. The ADE7761A powers up with the

V1A input active as normal. However, because there is no

current in the phase circuit, the signal on V1A is zero. This

causes a fault to be flagged and the active input to be swapped

to V1B (neutral). The meter can be calibrated in this mode, but

the phase and neutral CTs may differ slightly. Because under

no-fault conditions all billing is carried out using the phase CT,

the meter should be calibrated using the phase circuit. Of

course, both phase and neutral circuits can be calibrated.

MISSING NEUTRAL MODE

The ADE7761A integrates a novel fault detection that warns

and allows the ADE7761A to continue to bill in case a meter is

connected to only one wire (see Figure 31). For correct

operation of the ADE7761A in this mode, the VDD pin of the

ADE7761A must be maintained within the specified range (5 V

± 5%). The missing neutral detection algorithm is designed to

work over a line frequency of 45 Hz to 55 Hz.

ADE7761A

Rev. 0 | Page 19 of 24

RB + VR = RF.

05040-029

LOAD

244V rms

POWER

GENERATOR

Figure 31. Missing Neutral System Diagram

The ADE7761A detects a missing neutral condition by

continuously monitoring the voltage channel input (V2P − V2N).

The FAULT pin is held high when a missing neutral condition is

detected. In this mode, the ADE7761A continues to bill the

energy based on the signal level on the current channel (see

Figure 32). The billing rate or frequency outputs can be adjusted

by changing the dc level on the MISCAL pin.

MISSING NEUTRAL

GAIN ADJUSMTENT DIGITAL-TO-

FREQUENCY

CONVERTERS

CF F1 F2

05040-030

ZERO

CROSSING

DETECTION

A > B

B > A

B <> A

ADC

MISCAL

LPF

HPF

Figure 32. Energy Calculation in Missing Neutral Mode

Important Note for Billing of Active Energy

The ADE7761A provides pulse outputs—CF, F1, and F2—

intended to be used for the billing of active energy. Pulses

are generated at these outputs in two different situations.

Case 1: When the analog input V2P – V2N complies with the

conditions described in Figure 34, CF, F1, and F2 frequencies

are proportional to active power and can be used to bill active

energy.

Case 2: When the analog input V2P – V2N does not comply with

the conditions described in Figure 34, the ADE7761A does not

measure active energy but a quantity proportional to kAh. This

quantity is used to generate pulses on the same CF, F1, and F2.

This situation is indicated when the FAULT pin is high.

Analog Devices, Inc. cautions users of the ADE7761A about the

following:

• Billing active energy in Case 1 is consistent with the

understanding of the quantity represented by pulses on CF,

F1, and F2 outputs (watt-hour).

• Billing active energy while the ADE7761A is in Case 2 must

be decided knowing that the entity measured by the

ADE7761A in this case is ampere-hour and not watt-hour.

Users should be aware of this limitation and decide if the

ADE7761A is appropriate for their application.

Missing Neutral Detection

The ADE7761A continuously monitors the voltage input and

detects a missing neutral condition when the voltage input peak

value is smaller than 9% of the analog full scale or when no zero

crossings are detected on this input (see Figure 33).

FSFS

05040-031

9% OF FS

FILTER AND

THRESHOLD

AGND

ADC

MISSING

NEUTRAL

|V2|

PEAK

< 9% OF FULL SCALE

–V

– V

NO ZERO CROSSING ON V2OR

Figure 33. Missing Neutral Detection

The ADE7761A leaves the missing neutral mode for normal

operation when both conditions are no longer valid, that is, a

voltage peak value of greater than 9% of full scale and zero

crossing on the voltage channel is detected (see Figure 34).

05040-032

FILTER AND

THRESHOLD

GND

ADC

MISSING

NEUTRAL

|V2|

PEAK

> 9% OF FULL SCALE

AND

ZERO CROSSING ON V2

+9% OF FS

–9% OF FS

– V

Figure 34. Return to Normal Mode after Missing Neutral Detection

ADE7761A

Rev. 0 | Page 20 of 24

Missing Neutral Gain Calibration Example

When the ADE7761A is in missing neutral mode, the energy is

bill based on the active current input signal level. The frequency

outputs in this mode can be calibrated with the MISCAL analog

input pin. In this mode, applying a dc voltage of 330 mV on

MISCAL is equivalent to applying, in normal mode, a pure sine

wave on the voltage input with a peak value of 330 mV. The

MISCAL input can vary from 0 V to 660 mV (see the Analog

Inputs section). When set to 0 V, the frequency outputs are

close to zero. When set to 660 mV dc, the frequency outputs are

twice that when MISCAL is at 330 mV dc. In other words,

Equation 7 can be used in missing neutral mode by replacing

V2rms by MISCALrms /√2.

In normal mode, ac voltages of ±330 mV peak are applied to V1

and V2, and then the expected output frequency on F1 and F2 is

calculated as:

Gain =1 ; PGA =0

F1–4 = 1.7 Hz, SCF = S1 = S0 = 0

V1 = rms of 330 mV peak ac = 0.33/√2 V

V2 = rms of 330 mV peak ac = 0.33/√2 V

VREF = 2.5 V (nominal reference value)

Hz084.0

5.222

Hz7.133.033.070.5

21 =

××

=FrequencyFF

2/170.5

REF

rmsrms

FMISCALVGain

Frequenc

−

××××

(8)

CF Frequency = F1 − F2 Frequency × 64 = 5.4 Hz

In missing neutral mode, the ac voltage of ±330 mV peak is

applied to V1, no signal is connected on V2, and a 330 mV dc

input is applied to MISCAL. With the ADE7761A in the same

configuration as the previous example, the expected output

frequencies on CF, F1, and F2 are

where:

F1, F2 Frequency is the output frequency on F1 and F2 (Hz).

V1rms is the differential rms voltage signal on Channel 1 (V).

MISCALrms is the differential rms voltage signal on the MISCAL

pin (V). Hz084.0

5.22

Hz7.12/33.033.070.5

,2=

×××

=FrequencyFF 21

Gain is 1 or 16, depending on the PGA gain selection made

using logic input PGA. CF Frequency = F1, F2 Frequency × 64 = 5.4 Hz

VREF is the reference voltage (2.5 V ± 8%) (V).

F1-4 is one of four possible frequencies selected by using the

logic inputs S0 and S1 (see Table 6).

ADE7761A

Rev. 0 | Page 21 of 24

APPLICATIONS

INTERFACING TO A MICROCONTROLLER FOR

ENERGY MEASUREMENT

The easiest way to interface the ADE7761A to a microcontroller

is to use the CF high frequency output with the output frequency

scaling set to 2048 × F1, F2. This is done by setting SCF = 0

and S0 = S1 = 1 (see Table 8). With full-scale ac signals on the

analog inputs, the output frequency on CF is approximately

5.5 kHz. Figure 35 illustrates one scheme that could be used to

digitize the output frequency and carry out the necessary

averaging mentioned in the Frequency Output CF section.

TIME

±10%

AVERAGE

FREQUENCY

RIPPLE

MCU

UP/DOWN

COUNTER

LOGIC

REVP

ADE7761A

FAULT

05040-033

REVP MUST BE USED IF THE METER IS BIDIRECTIONAL OR

DIRECTION OF ENERGY FLOW IS NEEDED.

FAULT MUST BE USED TO RECORD ENERGY IN FAULT CONDITION.

Figure 35. Interfacing the ADE7761A to an MCU

As shown in Figure 35 the frequency output CF is connected to

an MCU counter or port, which counts the number of pulses in

a given integration time, determined by an MCU internal timer.

The average power, proportional to the average frequency, is

Time

Counter

PowerActiveAverageFrequencyAverage ==

The energy consumed during an integration period is

CounterTime

Time

Counter

TimePowerAverageEnergy =×=×=

For the purpose of calibration, this integration time could be

10 sec to 20 sec to accumulate enough pulses to ensure correct

averaging of the frequency. In normal operation, the integration

time could be reduced to 1 sec or 2 sec depending on, for

example, the required update rate of a display. With shorter

integration times on the MCU, the amount of energy in each

update may still have a small amount of ripple, even under

steady load conditions. However, over a minute or more, the

measured energy has no ripple.

SELECTING A FREQUENCY FOR AN ENERGY

METER APPLICATION

As shown in Table 6, the user can select one of four frequencies.

This frequency selection determines the maximum frequency

on F1 and F2. These outputs are intended to be used to drive

the energy register (electromechanical or other). Because only

four different output frequencies can be selected, the available

frequency selection was optimized for a meter constant of

100 impulses/kWh with a maximum current of between 10 A

and 120 A. Table 9 shows the output frequency for several

maximum currents (IMAX) with a line voltage of 240 V. In all

cases, the meter constant is 100 impulses/kWh.

Table 9. F1 and F2 Frequency at 100 Impulses/kWh

IMAX (A) F1 and F2 (Hz)

12.5 0.083

25 0.166

40 0.266

60 0.4

80 0.533

120 0.8

The F1–4 frequencies allow complete coverage of this range of

output frequencies on F1 and F2. When designing an energy

meter, the nominal design voltage on Channel 2 (voltage)

should be set to half-scale to allow for calibration of the meter

constant. The current channel should also be no more than half-

scale when the meter sees maximum load, which accommodates

overcurrent signals and signals with high crest factors. Table 10

shows the output frequency on F1 and F2 when both analog

inputs are half-scale. The frequencies listed in Table 10 align

well with those listed in Table 9 for maximum load.

Table 10. F1 and F2 Frequency with Half-Scale AC Inputs

S0 S1 F1–4 (Hz)

Frequency on F1 and F2, Ch 1 and Ch 2,

Half-Scale AC Inputs (Hz)

0 0 1.72 0.085

0 1 3.44 0.17

1 0 6.86 0.34

1 1 13.5 0.68

When selecting a suitable F1–4 frequency for a meter design, the

frequency output at IMAX (maximum load) with a meter constant

of 100 impulses/kWh should be compared with Column 4 of

Table 10. The frequency that is closest in Table 10 determines

the best choice of frequency (F1-4). For example, if a meter with

a maximum current of 40 A is being designed, the output

frequency on F1 and F2 with a meter constant of 100 impulses

per kWh is 0.266 Hz at 40 A and 240 V (see Table 9).

ADE7761A

Rev. 0 | Page 22 of 24

Looking at Table 10, the closest frequency to 0.266 Hz

in Column 4 is 0.17 Hz. Therefore, F2 (3.4 Hz; see Table 6) is

selected for this design.

Frequency Outputs

Figure 2 is a timing diagram for the various frequency outputs.

The high frequency CF output is intended for communication

and calibration purposes. CF produces a 90 ms wide, active

high pulse (t4) at a frequency that is proportional to active

power. The CF output frequencies are given in Table 8. As with

F1 and F2, if the period of CF (t5) falls below 180 ms, the CF

pulse width is set to half the period. For example, if the CF

frequency is 20 Hz, the CF pulse width is 25 ms.

No-Load Threshold

The ADE7761A includes a no-load threshold and start-up

current feature that eliminates creep effects in the meter. The

ADE7761A is designed to issue a minimum output frequency.

Any load generating a frequency lower than this minimum

frequency does not cause a pulse to be issued on F1, F2, or CF.

The minimum output frequency is given as 0.0045% of the full-

scale output frequency. (See Table 8 for maximum output

frequencies for ac signals).

For example, an energy meter with a meter constant of

100 impulses per kWh on F1, F2 using SCF = 1, S1 = 0, and

S0 = 1, the maximum output frequency at F1 or F2 is 0.68 Hz

and 43.52 Hz on CF. The minimum output frequency at F1

or F2 is 0.0045% of 0.68 Hz or 3.06 × 10–5 Hz. This is 1.96 ×

10–3 Hz at CF (64 × F1 Hz).

In this example, the no-load threshold is equivalent to 1.1 W of

load or a startup current of 4.6 mA at 240 V. Compare this value

to the IEC 32053-21 specification, which states that the meter

must start up with a load equal to or less than 0.4% of IB. For a

5 A (I

BB) meter, 0.4% of IB is equivalent to 20 mA. B

Note that the no-load threshold is not enabled when using the

high CF frequency mode: SCF = 0, S1 = S0 = 1.

NEGATIVE POWER INFORMATION

The ADE7761A detects when the current and voltage channels

have a phase shift greater than 90°. This mechanism can detect

a wrong connection of the meter or the generation of negative

power. The REVP pin output goes active high when negative

power is detected and active low when positive power is

detected. The REVP pin output changes state as a pulse is

issued on CF.

ADE7761A

Rev. 0 | Page 23 of 24

OUTLINE DIMENSIONS

COMPLIANT TO JEDEC STANDARDS MO-150-AE

060106-A

20 11

7.50

7.20

6.90

8.20

7.80

7.40

5.60

5.30

5.00

SEATING

PLANE

0.05 MIN

0.65 BSC

2.00 MAX

0.38

0.22

COPLANARITY

0.10

1.85

1.75

1.65

0.25

0.09

0.95

0.75

0.55

8°

4°

0°

Figure 36. 20-Lead Shrink Small Outline Package [SSOP]

(RS-20)

Dimensions shown in millimeters

ORDERING GUIDE

Model Temperature Range Package Description Package Option

ADE7761AARS –40°C to +85°C 20-Lead Shrink Small Outline Package (SSOP) RS-20

ADE7761AARS-RL –40°C to +85°C 20-Lead Shrink Small Outline Package (SSOP) RS-20

ADE7761AARSZ1–40°C to +85°C 20-Lead Shrink Small Outline Package (SSOP) RS-20

ADE7761AARSZ-RL1–40°C to +85°C 20-Lead Shrink Small Outline Package (SSOP) RS-20

ADE7761AARS-REF Reference Board

1 Z = Pb-free part.

ADE7761A

Rev. 0 | Page 24 of 24

NOTES

registered trademarks are the property of their respective owners.

D05040-0-7/06(0)