NCP1341 High-Voltage, Quasi-Resonant, Controller Featuring Valley Lock-Out Switching The NCP1341 is a highly integrated quasi-resonant flyback controller suitable for designing high-performance off-line power converters. With an integrated active X2 capacitor discharge feature, the NCP1341 can enable no-load power consumption below 30 mW. The NCP1341 features a proprietary valley-lockout circuitry, ensuring stable valley switching. This system works down to the 6th valley and transitions to frequency foldback mode to reduce switching losses. As the load decreases further, the NCP1341 enters quiet-skip mode to manage the power delivery while minimizing acoustic noise. The NCP1341 integrates power excursion mode (PEM) to minimize transformer size in designs requiring high transient load capability. If transient load capability is not desired, the NCP1340 offers the same performance and features without PEM. To help ensure converter ruggedness, the NCP1341 implements several key protective features such as internal brownout detection, a non-dissipative Over Power Protection (OPP) for constant maximum output power regardless of input voltage, a latched overvoltage and NTC-ready overtemperature protection through a dedicated pin, and line removal detection to safely discharge the X2 capacitors when the ac line is removed. Features * * * * * * * * * * * * * * * * Integrated High-Voltage Startup Circuit with Brownout Detection Integrated X2 Capacitor Discharge Capability Wide VCC Range from 9 V to 28 V 28 V VCC Overvoltage Protection Abnormal Overcurrent Fault Protection for Winding Short Circuit or Saturation Detection Internal Temperature Shutdown Valley Switching Operation with Valley-Lockout for Noise-Free Operation Frequency Foldback with 25 kHz Minimum Frequency Clamp for Increased Efficiency at Light Loads Skip Mode with Quiet-Skip Technology for Highest Performance During Light Loads Minimized Current Consumption for No Load Power Below 30 mW Frequency Jittering for Reduced EMI Signature Latching or Auto-Recovery Timer-Based Overload Protection Adjustable Overpower Protection (OPP) Fixed or Adjustable Maximum Frequency Clamp Fault Pin for Severe Fault Conditions, NTC Compatible for OTP (9-Pin Version Only) 4 ms Soft-Start Timer (c) Semiconductor Components Industries, LLC, 2017 October, 2017 - Rev. 4 1 www.onsemi.com 8 9 1 1 SOIC-9 NB D SUFFIX CASE 751BP SOIC-8 NB D SUFFIX CASE 751 MARKING DIAGRAM 9 1341xz ALYW G 1 1341xz x z A L Y W G = Specific Device Code = A or B = 1, 2, 3 or 4 = Assembly Location = Wafer Lot = Year = Work Week = Pb-Free Package PIN CONNECTIONS 1 HV Fault FMAX FB ZCD/OPP CS VCC DRV GND 1 HV FMAX FB VCC ZCD/OPP DRV GND CS (Top Views) ORDERING INFORMATION See detailed ordering and shipping information on page 2 of this data sheet. Publication Order Number: NCP1341/D NCP1341 TYPICAL APPLICATION SCHEMATIC + + + + Vout NCP1341xx HV FMAX + L N FB EMI Filter VCC ZCD/OPP DRV CS GND + Figure 1. NCP1341 8-Pin Typical Application Circuit Vout NCP1341xx Fault FMAX + L N FB EMI Filter HV VCC ZCD/OPP DRV CS GND -t + Figure 2. NCP1341 9-Pin Typical Application Circuit Table 1. ORDERABLE PART NUMBERS Ordering Code Device Marking Pins Fault Pin FMAX Pin PEM OTP Protection Overload Protection Frequency Clamp Jitter NCP1341A1D1R2G 1341A1 9 Yes Yes Yes Latched Latched None 1.3 kHz NCP1341B1DR2G 1341B1 8 No Yes Yes Auto-Restart Auto-Restart None 1.3 kHz NCP1341B1D1R2G 1341B1 9 Yes Yes Yes Auto-Restart Auto-Restart None 1.3 kHz NCP1341B4D1R2G 1341B4 9 Yes Yes Yes Auto-Restart None None None www.onsemi.com 2 NCP1341 FUNCTIONAL BLOCK DIAGRAM TSD IFMAX FMAX Control FMAX Abnormal OCP OVLD OVP OTP QR_FMAX Fault Management Dead-Time Control PEM_FMAX OPP FB Jitter Ramp KFB FB / PEM VDD PEM Clamp Quiet-Skip Control S DRV Q R PEM Control On-Time Control PEM Detect VCC ttout QR_FMAX RFB FB HV VCC OPP Control VFB(open) VCC(OVP) X2/BO Detect + VCC Management Fault Off-Time Control ZCD/OPP GND VDD Fault ICS PEM OVP ILIM1 Detect tLEB1 CS X2 Valley/VCO Control FB IFB BO PEM_FMAX tOVLD OVLD OTP OVP/OTP Detect IOTP VDD Fault RFault(clamp) OPP ILIM2 Detect tLEB2 8-Pin Count 4 VFault(clamp) Abnormal OCP 9-Pin Figure 3. NCP1341 Block Diagram Table 2. PIN FUNCTIONAL DESCRIPTION 8-Pin 9-Pin Pin Name - 1 Fault The controller enters fault mode if the voltage on this pin is pulled above or below the fault thresholds. A precise pull up current source allows direct interface with an NTC thermistor. Function 1 2 FMAX A resistor to ground sets the value for the maximum switching frequency in CCM mode. For versions x3, it also sets the maximum switching frequency in QR mode. For versions A/B, pulling this pin above 4 V switches the PEM control method to fixed frequency mode. 2 3 FB 3 4 ZCD/OPP 4 5 CS 5 6 GND Ground reference. 6 7 DRV This is the drive pin of the circuit. The DRV high-current capability (-0.5 /+0.8 A) makes it suitable to effectively drive high gate charge power MOSFETs. 7 8 VCC This pin is the positive supply of the IC. The circuit starts to operate when VCC exceeds 17 V and turns off when VCC goes below 9 V (typical values). After start-up, the operating range is 9 V up to 28 V. - 9 N/C Removed for creepage distance. 8 10 HV This pin is the input for the high voltage startup and brownout detection circuits. It also contains the line removal detection circuit to safely discharge the X2 capacitors when the line is removed. Feedback input for the QR Flyback controller. Allows direct connection to an optocoupler. A resistor divider from the auxiliary winding to this pin provides input to the demagnetization detection comparator and sets the OPP compensation level. Input to the cycle-by-cycle current limit comparator. www.onsemi.com 3 NCP1341 Table 3. MAXIMUM RATINGS Rating Symbol Value High Voltage Startup Circuit Input Voltage VHV(MAX) -0.3 to 700 V High Voltage Startup Circuit Input Current IHV(MAX) 20 mA Supply Input Voltage VCC(MAX) -0.3 to 30 V Supply Input Current ICC(MAX) 30 mA Supply Input Voltage Slew Rate dVCC/dt 1 V/ms Fault Input Voltage VFault(MAX) -0.3 to VCC + 0.7 V V Fault Input Current IFault(MAX) 10 mA Zero Current Detection and OPP Input Voltage VZCD(MAX) -0.3 to VCC + 0.7 V V Zero Current Detection and OPP Input Current IZCD(MAX) -2/+5 mA VMAX -0.3 to 5.5 V Maximum Input Voltage (Other Pins) Unit Maximum Input Current (Other Pins) IMAX 10 mA Driver Maximum Voltage (Note 1) VDRV -0.3 to VDRV(high) V IDRV(SRC) IDRV(SNK) 500 800 mA TJ -40 to 125 C TSTG -60 to 150 C Driver Maximum Current Operating Junction Temperature Storage Temperature Range Power Dissipation (TA = 25C, 1 oz. Cu, 42 DR2G Suffix, SOIC-8 D1R2G Suffix, SOIC-9 mm2 Copper Clad Printed Circuit) PD(MAX) mW 450 330 Thermal Resistance (TA = 25C, 1 oz. Cu, 42 mm2 Copper Clad Printed Circuit) DR2G Suffix, SOIC-8 D1R2G Suffix, SOIC-9 ESD Capability Human Body Model per JEDEC Standard JESD22-A114F (All pins except HV) Human Body Model per JEDEC Standard JESD22-A114F (HV Pin) Charge Device Model per JEDEC Standard JESD22-C101F Latch-Up Protection per JEDEC Standard JESD78E RqJA C/W 225 300 2000 800 1000 100 V V V mA Stresses exceeding those listed in the Maximum Ratings table may damage the device. If any of these limits are exceeded, device functionality should not be assumed, damage may occur and reliability may be affected. 1. Maximum driver voltage is limited by the driver clamp voltage, VDRV(high), when VCC exceeds the driver clamp voltage. Otherwise, the maximum driver voltage is VCC. www.onsemi.com 4 NCP1341 Table 4. ELECTRICAL CHARACTERISTICS: (VCC = 12 V, VHV = 120 V, VFault = open, VFB = 2 V, VCS = 0 V, VZCD = 0 V, VFMAX = 0 V, CVCC = 100 nF , CDRV = 100 pF, for typical values TJ = 25C, for min/max values, TJ is - 40C to 125C, unless otherwise noted) Characteristics Conditions Symbol Min Typ Max VCC(on) 17.0 18.0 9.0 - 6.5 0.70 18.0 19.0 9.5 - 7.5 1.05 Unit START-UP AND SUPPLY CIRCUITS Supply Voltage Startup Threshold Discharge Voltage During Line Removal Minimum Operating Voltage Operating Hysteresis Internal Latch / Logic Reset Level Transition from Istart1 to Istart2 dV/dt = 0.1 V/ms VCC increasing VCC decreasing VCC decreasing VCC(on) - VCC(off) VCC decreasing VCC increasing, IHV = 650 mA VCC(X2_reg) VCC(off) VCC(HYS) VCC(reset) VCC(inhibit) 16.0 17.0 8.5 7.5 4.5 0.40 VCC(off) Delay VCC decreasing tdelay(VCC_off) 25 32 40 ms Startup Delay Delay from VCC(on) to DRV Enable tdelay(start) - - 500 ms VHV(MIN) - - 40 V Minimum Voltage for Start-Up Current Source Inhibit Current Sourced from VCC Pin V Vcc = 0 V Istart1 0.2 0.5 0.65 mA Start-Up Current Sourced from VCC Pin Vcc = Vcc(on) - 0.5 V Istart2 2.4 3.75 5.0 mA Start-Up Circuit Off-State Leakage Current VHV = 162.5 V VHV = 325 V VHV = 700 V IHV(off1) IHV(off2) IHV(off3) - - - - - - 15 20 50 mA VCC = VCC(on) - 0.5 V VFB = 0 V fsw = 50 kHz, CDRV = open ICC1 ICC2 ICC3 - - - 0.115 0.230 1.0 0.150 0.315 1.5 Supply Current Fault or Latch Skip Mode (excluding FB current) Operating Current mA VCC Overvoltage Protection Threshold VCC Overvoltage Protection Delay VCC(OVP) 27 28 29 V tdelay(VCC_OVP) 25 32 40 ms tline(removal) 65 100 135 ms tline(discharge) 21 32 43 ms tline(detect) 21 32 43 ms X2 CAPACITOR DISCHARGE Line Voltage Removal Detection Timer Discharge Timer Duration Line Detection Timer Duration VCC Discharge Current VCC = 20 V HV Discharge Level ICC(discharge) 13 18 23 mA VHV(discharge) - - 30 V VBO(start) 107 112 116 V BROWNOUT DETECTION System Start-Up Threshold VHV increasing Brownout Threshold VHV decreasing VBO(stop) 93 98 102 V Hysteresis VHV increasing VBO(HYS) 9.0 14 - V Brownout Detection Blanking Time VHV decreasing tBO(stop) 40 70 100 ms Rise Time VDRV from 10% to 90% tDRV(rise) - 20 40 ns Fall Time VDRV from 90% to 10% tDRV(fall) - 5 30 ns IDRV(SRC) IDRV(SNK) - - 500 800 - - GATE DRIVE Current Capability Source Sink mA High State Voltage VCC = VCC(off) + 0.2 V, RDRV = 10 kW VCC = 30 V, RDRV = 10 kW VDRV(high1) VDRV(high2) 8.0 10 - 12 - 14 V Low Stage Voltage VFault = 0 V VDRV(low) - - 0.25 V VFB(open) 4.9 5.0 5.1 V KFB - 3 - - RFB 350 400 420 kW IFB 92 100 108 mA FEEDBACK Open Pin Voltage VFB to Internal Current Setpoint Division Ratio Internal Pull-Up Resistor VFB = 0.4 V Internal Pull-Up Current www.onsemi.com 5 NCP1341 Table 4. ELECTRICAL CHARACTERISTICS: (VCC = 12 V, VHV = 120 V, VFault = open, VFB = 2 V, VCS = 0 V, VZCD = 0 V, VFMAX = 0 V, CVCC = 100 nF , CDRV = 100 pF, for typical values TJ = 25C, for min/max values, TJ is - 40C to 125C, unless otherwise noted) Characteristics Conditions Symbol Min Typ Max Unit VFB decreasing VFB decreasing VFB decreasing VFB decreasing VFB decreasing VFB increasing VFB increasing VFB increasing VFB increasing VFB increasing V1to2 V2to3 V3to4 V4to5 V5to6 V6to5 V5to4 V4to3 V3to2 V2to1 0.987 0.846 0.776 0.705 0.635 1.199 1.269 1.340 1.410 1.551 1.050 0.900 0.825 0.750 0.675 1.275 1.350 1.425 1.500 1.650 1.113 0.954 0.874 0.795 0.715 1.352 1.431 1.511 1.590 1.749 VFMAX = 0.7 V VFMAX = 3.5 V fMAX1 fMAX2 fMAX3 100 300 60 110 360 75 120 420 85 VFMAX(mode) 3.85 4.00 4.15 V FEEDBACK Valley Thresholds Transition from 1st to 2nd valley Transition from 2nd to 3rd valley Transition from 3rd to 4th valley Transition from 4th to 5th valley Transition from 5th to 6th valley Transition from 6th to 5th valley Transition from 5th to 4th valley Transition from 4th to 3rd valley Transition from 3rd to 2nd valley Transition from 2nd to 1st valley Maximum Frequency Clamp Versions A2/B2/C2/D2/E2/F2 Versions A3/B3/C3/D3/E3/F3 Versions A3/B3/C3/D3/E3/F3 V kHz FMAX Secondary Mode Threshold IFMAX 9.0 10 11 mA ton(MAX) 28 32 40 ms VZCD decreasing VZCD(trig) 35 60 90 mV FMAX Pin Source Current Maximum On Time DEMAGNETIZATION INPUT ZCD threshold voltage ZCD hysteresis Demagnetization Propagation Delay ZCD Clamp Voltage Positive Clamp Negative Clamp VZCD increasing VZCD(HYS) 15 25 55 mV VZCD step from 4.0 V to -0.3 V tdemag - 80 250 ns IQZCD = 5.0 mA IQZCD = -2.0 mA VZCD(MAX) VZCD(MIN) 12.4 -0.9 12.7 -0.7 13 0 V tZCD(blank) 2.7 3.0 3.5 ms While in soft-start After soft-start complete t(tout1) t(tout2) 80 5.1 100 6.0 120 6.9 ms Current Limit Threshold Voltage Version C/D Version A/B/E/F VCS increasing VILIM1 0.76 0.95 0.80 1.00 0.84 1.05 Leading Edge Blanking Duration DRV minimum width minus tdelay(ILIM1) tLEB1 220 265 330 ns Step VCS 0 V to VILIM1 + 0.5 V, VFB = 4 V tdelay(ILIM1) - 95 175 ns Step VCS 0 V to 0.7 V, VFB = 2 V tdelay(PWM) - 125 175 ns Vfreeze 170 200 230 mV 1.125 1.400 1.200 1.500 1.275 1.600 Blanking Delay After Turn-Off Timeout After Last Demagnetization Detection CURRENT SENSE Current Limit Threshold Propagation Delay PWM Comparator Propagation Delay Minimum Peak Current Freeze Setpoint V Abnormal Overcurrent Fault Threshold Version C/D Version A/B/E/F VCS increasing, VFB = 4 V Abnormal Overcurrent Fault Blanking Duration DRV minimum width minus tdelay(ILIM2) tLEB2 80 110 140 ns Step VCS 0 V to VILIM2 + 0.5 V, VFB = 4 V tdelay(ILIM2) - 80 175 ns nILIM2 - 4 - tOPP(delay) - 95 175 ns tOPP(blank) 220 280 330 ns ICS 0.7 1.0 1.5 mA Abnormal Overcurrent Fault Propagation Delay Number of Consecutive Abnormal Overcurrent Faults to Enter Latch Mode Overpower Protection Delay VCS dv/dt = 1 V/ms, measured from VOPP(MAX) to DRV falling edge Overpower Signal Blanking Delay Pull-Up Current Source VCS = 1.5 V www.onsemi.com 6 VILIM2 V NCP1341 Table 4. ELECTRICAL CHARACTERISTICS: (VCC = 12 V, VHV = 120 V, VFault = open, VFB = 2 V, VCS = 0 V, VZCD = 0 V, VFMAX = 0 V, CVCC = 100 nF , CDRV = 100 pF, for typical values TJ = 25C, for min/max values, TJ is - 40C to 125C, unless otherwise noted) Characteristics Conditions Symbol Min Typ Max Unit Jitter Frequency fjitter 1.0 Peak Jitter Voltage Added to PWM Comparator Vjitter 90 1.3 1.6 kHz 100 115 mV 0.760 0.630 0.800 0.667 0.840 0.705 JITTERING POWER EXCURSION MODE PEM Activation Threshold Versions A/B/C/D Versions E/F VPEM Maximum Duty Ratio During PEM V DMAX - 75 - % Maximum FB Voltage for Off-Time Scaling VFB increasing VFB(MAX) 3.5 - - V Maximum Frequency Scaling During PEM VFB = 3.6 V Kscale(MAX) 2.2 - - - VPEM(arm) 1.0 1.5 2.0 V tSSTART 2.8 4.0 5.0 ms PEM Arming Threshold FAULT PROTECTION Soft-Start Period Flyback Overload Fault Timer Overvoltage Protection (OVP) Threshold Measured from 1st DRV pulse to VCS = VILIM1 VCS = VILIM1 tOVLD 120 160 200 ms VFault increasing VFault(OVP) 2.79 3.00 3.21 V OVP Detection Delay VFault increasing tdelay(OVP) 22.5 30 37.5 ms Overtemperature Protection (OTP) Threshold (Note 2) VFault decreasing VFault(OTP_in) 380 400 420 mV VFault increasing Versions B/D/F Only VFault(OTP_out) 874 910 966 mV VFault decreasing tdelay(OTP) 22.5 30 37.5 ms VFault = VFault(OTP_in) + 0.2 V IOTP 42.5 45.0 48.5 mA Fault Input Clamp Voltage VFault(clamp) 1.15 1.7 2.25 V Fault Input Clamp Series Resistor RFault(clamp) 1.32 1.55 1.78 kW trestart 1.8 2.0 2.2 s Overtemperature Protection (OTP) Exiting Threshold (Note 2) OTP Detection Delay OTP Pull-Up Current Source Autorecovery Timer LIGHT/NO LOAD MANAGEMENT Minimum Frequency Clamp Dead-Time Added During Frequency Foldback VFB = 300 mV Quiet-Skip Timer fMIN 21.5 25 27.0 kHz tDT(MAX) 34 - - ms tquiet 1.25 - - ms Skip Threshold VFB decreasing Vskip 263 300 337 mV Skip Hysteresis VFB increasing Vskip(HYS) 10.0 37.5 60.0 mV THERMAL PROTECTION Thermal Shutdown Temperature increasing TSHDN - 140 - C Thermal Shutdown Hysteresis Temperature decreasing TSHDN(HYS) - 40 - C 2. NTC with R110 = 8.8 kW www.onsemi.com 7 NCP1341 INTRODUCTION The NCP1341 implements a quasi-resonant flyback converter utilizing current-mode architecture where the switch-off event is dictated by the peak current. This IC is an ideal candidate where low parts count and cost effectiveness are the key parameters, particularly in ac-dc adapters, open-frame power supplies, etc. The NCP1341 incorporates all the necessary components normally needed in modern power supply designs, bringing several enhancements such as non-dissipative overpower protection (OPP), brownout protection, and frequency reduction management for optimized efficiency over the entire power range. Accounting for the needs of extremely low standby power requirements, the controller features minimized current consumption and includes an automatic X2 capacitor discharge circuit that eliminates the need to install power-consuming resistors across the X2 input capacitors. A novel power excursion mode (PEM) is also included to allow brief operation in CCM at up to 2x the maximum output power without the need for a larger transformer. * High-Voltage Start-Up Circuit: Low standby power consumption cannot be obtained with the classic resistive start-up circuit. The NCP1341 incorporates a high-voltage current source to provide the necessary current during start-up and then turns off during normal operation. * Internal Brownout Protection: The ac input voltage is sensed via the high-voltage pin. When this voltage is too low, the NCP1341 stops switching. No restart attempt is made until the ac input voltage is back within its normal range. * X2-Capacitor Discharge Circuitry: Per the IEC60950 standard, the time constant of the X2 input capacitors and their associated discharge resistors must be less than 1 s in order to avoid electrical shock when the user unplugs the power supply and inadvertently touches the ac input cord terminals. By providing an automatic means to discharge the X2 capacitors, the NCP1341 eliminates the need to install X2 discharge resistors, thus reducing power consumption. * Quasi-Resonant, Current-Mode Operation: Quasi-Resonant (QR) mode is a highly efficient mode of operation where the MOSFET turn-on is synchronized with the point where its drain-source voltage is at the minimum (valley). A drawback of this mode of operation is that the operating frequency is inversely proportional to the system load. The NCP1341 incorporates a valley lockout (VLO) and frequency foldback technique to eliminate this drawback, thus maximizing the efficiency over the entire power range. * Valley Lockout: In order to limit the maximum frequency while remaining in QR mode, one would * * * * * * * traditionally use a frequency clamp. Unfortunately, this can cause the controller to jump back and forth between two different valleys, which is often undesirable. The NCP1341 patented VLO circuitry solves this issue by determining the operating valley based on the system load, and locking out other valleys unless a significant change in load occurs. Frequency Foldback: As the load continues to decrease, it becomes beneficial to reduce the switching frequency. When the load is light enough, the NCP1341 enters frequency foldback mode. During this mode, the peak current is frozen and dead-time is added to the switching cycle, thus reducing the frequency and switching operation to discontinuous conduction mode (DCM). Dead-time continues to be added until skip mode is reached, or the switching frequency reaches its minimum level of 25 kHz. Skip Mode: To further improve light or no-load power consumption while avoiding audible noise, the NCP1341 enters skip mode when the operating frequency reaches its minimum value. foldback isavoid acoustic noise, the circuit prevents the switching frequency from decaying below 25 kHz. This allows regulation via burst of pulses at 25 kHz or greater instead of operating in the audible range. Quiet-Skip: To further reduce acoustic noise, the NCP1341 incorporates a novel circuit to prevent the skip mode burst period from entering the audible range as well. Internal OPP: In order to limit power delivery at high line, a scaled version of the negative voltage present on the auxiliary winding during the on-time is routed to the ZCD/OPP pin. This provides the designer with a simple and non-dissipative means to reduce the maximum power capability as the bulk voltage increases. Frequency Jittering: In order to reduce the EMI signature, a low frequency triangular voltage waveform is added to the iniput of the PWM comparator. This helps by spreading out the energy peaks during noise analysis. Internal Soft-Start: The NCP1341 includes a 4 ms soft-start to prevent the main power switch from being overly stressed during start-up. Soft-start is activated each time a new startup sequence occurs or during auto-recovery mode. Dedicated Fault Input: The NCP1341 includes a dedicated fault input. It can be used to sense an overvoltage condition and latch off the controller by pulling the pin above the overvoltage protection (OVP) threshold. The controller is also disabled if the Fault pin is pulled below the overtemperature protection (OTP) www.onsemi.com 8 NCP1341 * * * threshold. The OTP threshold is configured for use with a NTC thermistor. Overload/Short-Circuit Protection: The NCP1341 implements overload protection by limiting the maximum time duration for operation during overload conditions. The overload timer operates whenever the maximum peak current is reached. In addition to this, special circuitry is included to prevent operation in CCM during extreme overloads, such as an output short-circuit. Maximum Frequency Clamp: The NCP1341 includes a maximum frequency clamp. In all versions, the clamp is available disabled or fixed at 110 kHz. It can also be adjusted via an external resistor from the FMAX Pin to ground, or be disabled by pulling the FMAX pin above 4 V. Power Excursion Mode (PEM): When the power demand exceeds the power excursion threshold, the NCP1341 enters Power Excursion Mode (PEM) and forces the system into CCM to allow momentary power excursions of up to 2x for A and B versions or 1.5x for C and D versions, thus reducing or eliminating the need AC CON for a larger transformer. For versions E and F, the PEM control mode is set to fixed frequency, where the switching frequency is frozen and the peak current is increased to achieve 2x power. This allows for lower switching losses at the expense of a slightly larger transformer. This is also accomplished in versions A and B to achieve 1.5x power by pulling the FMAX pin above 4 V. HIGH VOLTAGE START-UP The NCP1341 contains a multi-functional high voltage (HV) pin. While the primary purpose of this pin is to reduce standby power while maintaining a fast start-up time, it also incorporates brownout detection and line removal detection. The HV pin must be connected directly to the ac line in order for the X2 discharge circuit to function correctly. Line and neutral should be diode "ORed" before connecting to the HV pin as shown in Figure 4. The diodes prevent the pin voltage from going below ground. A resistor in series with the pin should be used to protect the pin during EMC or surge testing. A low value resistor should be used (<5 kW) to reduce the voltage offset during start-up. EMI HV Controller Figure 4. High-Voltage Input Connection Start-up and VCC Management During start-up, the current source turns on and charges the VCC capacitor with Istart2 (typically 6 mA). When Vcc reaches VCC(on) (typically 16.0 V), the current source turns off. If the input voltage is not high enough to ensure a proper start-up (i.e. VHV has not reached VBO(start)), the controller will not start. VCC then begins to fall because the controller bias current is at ICC2 (typically 1 mA) and the auxiliary supply voltage is not present. When VCC falls to VCC(off) (typically 10.5 V), the current source turns back on and charges VCC. This cycle repeats indefinitely until VHV reaches VBO(start). Once this occurs, the current source immediately turns on and charges VCC to VCC(on), at which point the controller starts (see Figure 6). When VCC is brought below VCC(inhibit), the start-up current is reduced to Istart1 (typically 0.5 mA). This limits power dissipation on the device in the event that the VCC pin is shorted to ground. Once VCC rises back above VCC(inhibit), the start-up current returns to Istart2. Once VCC reaches VCC(on), the controller is enabled and the controller bias current increases to ICC3 (typically 2.0 mA). However, the total bias current is greater than this due to the gate charge of the external switching MOSFET. The increase in ICC due to the MOSFET is calculated using Equation 1. DI CC + f sw @ Q G @ 10 -3 (eq. 1) where DICC is the increase in milliamps, fsw is the switching frequency in kilohertz and QG is the gate charge of the external MOSFET in nanocoulombs. CVCC must be sized such that a VCC voltage greater than VCC(off) is maintained while the auxiliary supply voltage increases during start-up. If CVCC is too small, VCC will fall below VCC(off) and the controller will turn off before the auxiliary winding supplies the IC. The total ICC current after the controller is enabled (ICC3 plus DICC) must be considered to correctly size CVCC. www.onsemi.com 9 NCP1341 Figure 5. Start-up Circuitry Block Diagram VHV VBO(start) VHV(MIN) VCC VCC(on) VCC(off) Start-up Current = Istart2 Start-up Current = Istart1 VCC(inhibit ) DRV Figure 6. Start-up Timing www.onsemi.com 10 tdelay (start ) NCP1341 DRIVER The peak current level is clamped during the soft-start phase. The setpoint is actually limited by a clamp level ramping from 0 to 1.0 V within 4 ms. In addition to the PWM comparator, a dedicated comparator monitors the current sense voltage, and if it reaches the maximum value, VILIM (typically 1.00 V), the gate driver is turned off and the overload timer is enabled. This occurs even if the limit imposed by the feedback voltage is higher than VILIM1. Due to the parasitic capacitances of the MOSFET, a large voltage spike often appears on the CS Pin at turn-on. To prevent this spike from falsely triggering the current sense circuit, the current sense signal is blanked for a short period of time, tLEB1 (typically 275 ns), by a leading edge blanking (LEB) circuit. Figure 7 shows the schematic of the current sense circuit. The peak current is also limitied to a minimum level, Vfreeze (0.2 V, typically). This results in higher efficiency at light loads by increasing the minimum energy delivered per switching cycle, while reducing the overall number of switching cycles during light load. The NCP1341 maximum supply voltage, VCC(MAX), is 28 V. Typical high-voltage MOSFETs have a maximum gate voltage rating of 20 V. The DRV pin incorporates an active voltage clamp to limit the gate voltage on the external MOSFETs. The DRV voltage clamp, VDRV(high) is typically 12 V with a maximum limit of 14 V. REGULATION CONTROL Peak Current Control The NCP1341 is a peak current-mode controller, thus the FB voltage sets the peak current flowing in the transformer and the MOSFET. This is achieved by sensing the MOSFET current across a resistor and applying the resulting voltage ramp to the non-inverting input of the PWM comparator through the CS pin. The current limit threshold is set by applying the FB voltage divided by KFB (typically 3) to the inverting input of the PWM comparator. When the current sense voltage ramp exceeds this threshold, the output driver is turned off, however, the peak current is affected by several functions (see Figure 7): Figure 7. Current Sense Logic www.onsemi.com 11 NCP1341 Zero Current Detection As shown by Figure 13, a valley is detected once the ZCD pin voltage falls below the demagnetization threshold, VZCD(trig), typically 55 mV. The controller will either switch once the valley is detected or increment the valley counter, depending on the FB voltage. The NCP1341 is a quasi-resonant (QR) flyback controller. While the power switch turn-off is determined by the peak current set by the feedback loop, the switch turn-on is determined by the transformer demagnetization. The demagnetization is detected by monitoring the transformer auxiliary winding voltage. Turning on the power switch once the transformer is demagnetized has the benefit of reduced switching losses. Once the transformer is demagnetized, the drain voltage starts ringing at a frequency determined by the transformer magnetizing inductance and the drain lump capacitance, eventually settling at the input voltage. A QR flyback controller takes advantage of the drain voltage ringing and turns on the power switch at the drain voltage minimum or "valley" to reduce switching losses and electromagnetic interference (EMI). Overpower Protection The average bulk capacitor voltage of the QR flyback varies with the RMS line voltage. Thus, the maximum power capability at high line can be much higher than desired. An integrated overpower protection (OPP) circuit provides a relatively constant output power limit across the input voltage on the bulk capacitor, Vbulk. Since it is a high-voltage rail, directly measuring Vbulk will contribute losses in the sensing network that will greatly impact the standby power consumption. The NCP1341 OPP circuit achieves this without the need for a high-voltage sensing network, and is essentially lossless. Figure 8. OPP Circuit Schematic www.onsemi.com 12 VAUX (V) NCP1341 N - AUX . VBULK NP Figure 9. Auxiliary Winding Voltage Since the auxiliary winding voltage during the power switch on time is a reflection of the input voltage scaled by the primary to auxiliary winding turns ratio, NP:AUX (see Figure 9), OPP is achieved by scaling down reflected voltage during the on-time and applying it to the ZCD pin as a negative voltage, VOPP. The voltage is scaled down by a resistor divider comprised of ROPPU and ROPPL. The maximum internal current setpoint (VCS(OPP)) is simply the sum of VOPP and the peak current sense threshold, VILIM1. Figure 8 shows the schematic for the OPP circuit. As OPP is added, eventually VILIM1 will equal VPEM. At this point, any additional OPP will reduce both thresholds equally. The adjusted peak current limit is calculated using Equation 2. For example, a VOPP of -150 mV results in a peak current limit of 650 mV in NCP1341. V CS(OPP) + V OPP ) V ILIM1 Where VAUX is the voltage across the auxiliary winding and VF is the DOPP forward voltage drop. The ratio between RZCD and ROPPL is given by Equation 5. It is obtained by combining Equations 3 and 4. V R ZCD * V F * V ZCD + AUX R OPPL V ZCD A design example is shown below: System Parameters: V AUX + 18 V V F + 0.6 V N P:AUX + 0.18 The ratio between RZCD and ROPPL is calculated using Equation 5 for a minimum VZCD of 8 V. R ZCD 18 V * 0.6 V * 8 V + + 1.2 kW R OPPL 8V (eq. 2) To ensure optimal zero-crossing detection, a diode is needed to bypass ROPPU during the off-time. Equation 3 is used to calculate ROPPU and ROPPL. R ZCD ) R OPPU N @ V bulk * V OPP + * P:AUX R OPPL V OPP RZCD is arbitrarily set to 1 kW. ROPPL is also set to 1 kW because the ratio between the resistors is close to 1. The NCP1341 maximum overpower compensation or peak current setpoint reduction is 31.25% for a VOPP of -250 mV. We will use this value for the following example: Substituting values in Equation 3 and solving for ROPPU we obtain: (eq. 3) ROPPU is selected once a value is chosen for ROPPL. ROPPL is selected large enough such that enough voltage is available for the zero-crossing detection during the off-time. It is recommended to have at least 8 V applied on the ZCD pin for good detection. The maximum voltage is internally clamped to VCC. The off-time voltage on the ZCD Pin is given by Equation 4. V ZCD + R OPPL @ V AUX * V F R ZCD ) R OPPL (eq. 5) R ZCD ) R OPPU 0.18 @ 370 V * (-0.25 V) + + 271 R OPPL -0.25 V R OPPU + 271 @ R OPPL * R ZCD R OPPU + 271 @ 1 kW * 1 kW + 270 kW For optimum performance over temperature, it is recommended to keep ROPPL below 3 kW. (eq. 4) www.onsemi.com 13 NCP1341 Soft-Start in CCM for several cycles until the voltage on the ZCD pin is high enough to prevent the timer from running. Therefore, a longer timeout period, ttout1 (typically 100 ms), is used during soft-start to prevent CCM operation. Soft-start is achieved by ramping up an internal reference, VSSTART, and comparing it to the current sense signal. VSSTART ramps up from 0 V once the controller initially powers up. The peak current setpoint is then limited by the VSSTART ramp resulting in a gradual increase of the switch current during start-up. The soft-start duration, tSSTART, is typically 4 ms. During startup, demagnetization phases are long and difficult to detect since the auxiliary winding voltage is very small. In this condition, the 6 ms steady-state timeout is generally shorter than the inductor demagnetization period. If it is used to restart a switching cycle, it can cause operation Frequency Jittering In order to help meet stringent EMI requirements, the NCP1341 features frequency jittering to average the energy peaks over the EMI frequency range. As shown in Figure 10, the function consists of summing a 0 to 100 mV, 1.3 kHz triangular wave (Vjitter) with the CS signal immediately before the PWM comparator. This current acts to modulate the on-time and hence the operation frequency. Figure 10. Jitter Implementation 1000 Since the jittering function modulates the peak current level, the FB signal will attempt to compensate for this effect in order to limit the output voltage ripple. Therefore, the bandwidth of the feedback loop must be well below the jitter frequency, or the jitter function will be filtered by the loop. Due to the frozen peak current, the effect of the jittering circuit will not be seen during frequency foldback mode. 900 FSW(MAX) (kHz) 800 Maximum Frequency Clamp The NCP1341 includes a maximum frequency clamp. In all versions, the clamp is available disabled or fixed at 110 kHz. It can also be adjusted via an external resistor from the FMAX Pin to ground, or disabled by pulling the FMAX pin above 4 V. The maximum frequency can be programmed using Equation 6, and is shown in Figure 11. F SW(MAX) + 261 kHz * 1 V R FMAX * 10 mA 700 600 500 400 300 200 100 0 0 50 100 150 200 250 RFMAX (kW) 300 350 400 Figure 11. FSW(MAX) vs. RFMAX (eq. 6) www.onsemi.com 14 NCP1341 LIGHT LOAD MANAGEMENT a valley is selected, the controller stays locked in this valley until the output power changes significantly. This technique extends the QR mode operation over a wider output power range while maintaining good efficiency and limiting the maximum operating frequency. The operating valley (1st, 2nd, 3rd, 4th, 5th or 6th) is determined by the FB voltage. An internal counter increments each time a valley is detected by the ZCD/OPP Pin. Figure 12 shows a typical frequency characteristic obtainable at low line in a 65 W application. Valley Lockout Operation The operating frequency of a traditional QR flyback controller is inversely proportional to the system load. In other words, a load reduction increases the operating frequency. A maximum frequency clamp can be useful to limit the operating frequency range. However, when used by itself, such an approach often causes instabilities since when this clamp is active, the controller tends to jump (or hesitate) between two valleys, thus generating audible noise. Instead, the NCP1341 also incorporates a patented valley lockout (VLO) circuitry to eliminate valley jumping. Once 1x10 Fsw (Hz) 8x10 6x10 6th 5th 4th 5 2x10 2nd 1st VCO 4 mode 4 6th 4x10 3rd 5th 4th 3rd 2nd 1st 4 4 0 VCO mode 0 20 40 60 Pout (W) Figure 12. Valley Lockout Frequency vs. Output Power peak current to deliver the necessary output power. Each valley selection comparator features a 600 mV hysteresis that helps stabilize operation despite the FB voltage swing produced by the regulation loop. When an "n" valley is asserted by the valley selection circuitry, the controller is locked in this valley until the FB voltage decreases to the lower threshold ("n+1" valley activates) or increases to the "n valley threshold" + 600 mV ("n-1" valley activates). The regulation loop adjusts the Table 5. VALLEY FB THRESHOLDS (typical values) FB Falling 1st to 2nd valley FB Rising 1.050 V 2nd to 1st valley 1.650 V 0.900 V 3rd valley 0.825 V 4th 4th to 5th valley 0.750 V 0.675 V 6th 1.275 V 2nd 3rd 5th to 3rd to 4th to 6th valley valley Valley Timeout to 2nd valley 1.500 V to 3rd valley 1.425 V 5th to 4th valley 1.350 V to 5th valley signal acts as a substitute for the ZCD signal to the valley counter. Figure 13 shows the valley timeout circuit schematic. The steady state timeout period, ttout2, is set at 6 ms (typical) to limit the frequency step. During startup, the voltage offset added by the OPP diode, DOPP, prevents the ZCD Comparator from accurately detecting the valleys. In this condition, the steady state In case of extremely damped oscillations, the ZCD comparator may not be able to detect the valleys. In this condition, drive pulses will stop while the controller waits for the next valley or ZCD event. The NCP1341 ensures continued operation by incorporating a maximum timeout period after the last demagnetization detection. The timeout www.onsemi.com 15 NCP1341 the FB voltage sets VLO mode to turn on at the fifth valley, and the ZCD ringing is damped such that the ZCD circuit is only able to detect: * Valleys 1 to 4: the circuit generates a DRV pulse 6 ms (steady-state timeout delay) after the 4th valley detection. * Valleys 1 to 3: the timeout delay must run twice, and the circuit generates a DRV pulse 12 ms after the 3rd valley detection. timeout period will be shorter than the inductor demagnetization period causing CCM operation. CCM operation lasts for a few cycles until the voltage on the ZCD pin is high enough to detect the valleys. A longer timeout period, ttout1, (typically 100 ms) is set during soft-start to limit CCM operation. In VLO operation, the number of timeout periods are counted instead of valleys when the drain-source voltage oscillations are too damped to be detected. For example, if Figure 13. Valley Timeout Circuitry Frequency Foldback As the output load decreases (FB voltage decreases), the valleys are incremented from 1 to 6. When the sixth valley is reached, if the FB voltage further decreases to 0.6 V, the peak current setpoint becomes internally frozen to Vfreeze (0.2 V typically), and the controller enters frequency foldback mode (FF). During this mode, the controller regulates the power delivery by modulating the switching frequency. In frequency foldback mode, the controller reduces the switching frequency by adding dead-time after the 6th valley is detected. This dead-time increases as the FB voltage decreases. There is no discontinuity when the system transitions from VLO to FF and the frequency smoothly reduces as FB decreases. The dead-time circuit is designed to add 0 ms dead-time when VFB = 0.6 V and linearly increases the total dead-time to tDT(3) (32 ms minimum) as VFB falls down to 0.3 V. The minimum frequency clamp prevents the switching frequency from dropping below 25 kHz to eliminate the risk of audible noise. Figure 14 summarizes the VLO to FF operation with respect to the FB voltage. www.onsemi.com 16 NCP1341 Operating Mode IIIII IIIII IIIII IIIII IIIII IIIII IIIII IIIII IIIII IIIII IIIII V decreases FB FF V FB increases Valley 6 Valley 5 Valley 4 Fault ! Valley 3 Valley 2 Valley 1 0.60 0.67 0.75 0.82 0.90 1.05 1.28 1.35 1.43 1.50 1.65 Figure 14. Valley Lockout Thresholds www.onsemi.com 17 3.5 V (V) FB NCP1341 Minimum Frequency Clamp and Skip Mode as the current drive pulses ends - it does not stop immediately. Once switching stops, FB will rise. As soon as FB crosses the skip-exit threshold, drive pulses will resume, but the controller remains in burst mode. At this point, a 1250 ms (min) timer, tquiet, is started together with a count-to-3 counter. The next time the FB voltage drops below the skip-in threshold, drive pulses stop at the end of the current pulse as long as 3 drive pulses have been counted (if not, they do not stop until the end of the 3rd pulse). They are not allowed to start again until the timer expires, even if the skip-exit threshold is reached first. It is important to note that the timer will not force the next cycle to begin - i.e. if the natural skip frequency is such that skip-exit is reached after the timer expires, the drive pulses will wait for the skip-exit threshold. This means that during no-load, there will be a minimum of 3 drive pulses, and the burst-cycle period will likely be much longer than 1250 ms. This operation helps to improve efficiency at no-load conditions. In order to exit burst mode, the FB voltage must rise higher than 800 mV. If this occurs before tquiet expires, the drive pulses will resume immediately - i.e. the controller won't wait for the timer to expire. Figure 15 provides an example of how Quiet-Skip works. As mentioned previously, the circuit prevents the switching frequency from dropping below fMIN (25 kHz typical). When the switching cycle would be longer than 40 ms, the circuit forces a new switching cycle. However, the fMIN clamp cannot generate a DRV pulse until the demagnetization is completed. In other words, it will not cause operation in CCM. Since the NCP1341 forces a minimum peak current and a minimum frequency, the power delivery cannot be continuously controlled down to zero. Instead, the circuit starts skipping pulses when the FB voltage drops below the skip level, Vskip, and recovers operation when VFB exceeds Vskip + Vskip(HYS). This skip-mode method provides an efficient method of control during light loads. Quiet-Skip To further avoid acoustic noise, the circuit prevents the burst frequency during skip mode from entering the audible range by limiting it to a maximum of 800 Hz. This is achieved via a timer (tquiet) that is activated during Quiet-Skip. The start of the next burst cycle is prevented until this timer has expired. As the output power decreases, the switching frequency decreases. Once it hits 25 kHz, the skip-in threshold is reached and burst mode is entered - switching stops as soon www.onsemi.com 18 NCP1341 Figure 15. Quiet-Skip Timing Diagram POWER EXCURSION MODE (PEM) When the power demand exceeds the maximum power limit, the NCP1341 linearly increases the switching frequency forcing the power stage into CCM. Versions C and D accomplish this by linearly increasing the switching frequency up to 2.5x, thus eliminating the need for a larger transformer. Versions A and B achieve 2x power by also www.onsemi.com 19 NCP1341 output power. The frequency continues to be scaled until the maximum switching frequency (set by FMAX) or the maximum feedback voltage, VFB(MAX) (3.5 V typical), is reached. This operation continues as long as the controller remains in PEM, and the PEM comparator is tripped before each drive turn-off. Once a drive turn-off occurs without first tripping the PEM comparator, PEM is exited immediately (in the same cycle) and the controller immediately defaults back to QR mode with the next switching cycle starting at the ZCD transition. Since CCM operation is maintained via off-time modulation instead of fixed-frequency duty cycle modulation, the system is naturally immune to subharmonic oscillations and slope compensation is not required. In addition to operation in CCM, the NCP1341 contains a maximum CS setpoint, VILIM1 (typically 1.0 V), to allow a 25% increase in peak current. When this comparator triggers, the drive pulse is terminated. This corresponds to a FB voltage of 3 V (typical). The VILIM1 comparator shares the same LEB as the VPEM comparator. While FB voltages higher than 3 V will not cause any additional increase in peak current, the switching frequency continues to increase until the FB pin reaches VFB(MAX). At this point, the switching frequency will be scaled by a maximum value of Kfscale(MAX), 2.5 typical, provided FMAX has not been reached. Figure 16 shows the block schematic for PEM, while Figure 17 shows the timing for a fixed frequency. Figure 18 shows the timing with a frequency excursion. increasing the peak current by 25%, requiring a significantly smaller transformer than a converter that remained in QR mode. Versions E and F achieve 2x power by freezing the switching frequency and increasing the peak current by 50%. This allows for lower switching losses at the expense of a slightly larger transformer. This is also accomplished in versions A and B by pulling the FMAX pin above 4 V, however the power increase is limited to 1.5x. In all versions, the maximum switching frequency (and power) is set by the FMAX pin. The NCP1341 contains a register to store the off-time during QR mode. During each switching period, the off-time is measured and the register is updated. As long as the PEM comparator is not tripped, this operation will continue indefinitely. When the PEM comparator is tripped (due to an increase in power demand), the NCP1341 will enter PEM on the following cycle. During PEM, the stored value in the off-time register becomes a maximum off-time clamp, and when that clamp is reached, the next drive cycle will commence. Since the demagnetization time of a QR flyback is directly proportional to the load, as the load increases, the system will naturally enter CCM with a fixed off-time. The switching frequency is then determined by the on-time (which increases with load) and the fixed off-time. This operation alone provides a 1.5x power increase. In order to achieve 2x power, the off-time clamp is decreased linearly as the FB voltages increases. This has the effect of increasing the switching frequency to boost the Figure 16. PEM Block Diagram www.onsemi.com 20 NCP1341 Figure 17. PEM Timing for Fixed Frequency Figure 18. PEM Timing for Scaled Frequency www.onsemi.com 21 NCP1341 FAULT MANAGEMENT external latch input. When the NCP1341 detects a latching fault, the driver is immediately disabled. The operation during a latching fault is identical to that of a non-latching fault except the controller will not attempt to restart at the next VCC(on), even if the fault is removed. In order to clear the latch and resume normal operation, VCC must first be allowed to drop below VCC(reset) or a line removal event must be detected. This operation is shown in Figure 19. The NCP1341 contains three separate fault modes. Depending on the type of fault, the device will either latch off, restart when the fault is removed, or resume operation after the auto-recovery timer expires. Latching Faults Some faults will cause the NCP1341 to latch off. These include the abnormal OCP (AOCP), VCC OVP, and the Fault Fault Applied Fault Removed time V CC V CC (on) V CC (off) time FDRV time I HV Istart 2 I start (off) time Figure 19. Operation During Latching Fault www.onsemi.com 22 NCP1341 Non-Latching Faults re-enabled when VCC reaches VCC(on) according to the initial power-on sequence, provided VHV is above VBO(start). This operation is shown in Figure 20. When VHV is reaches VBO(start), VCC immediately charges to VCC(on). If VCC is already above VCC(on) when the fault is removed, the controller will start immediately as long as VHV is above VBO(start). When the NCP1341 detects a non-latching fault (brownout or thermal shutdown), the drivers are disabled, and VCC falls towards VCC(off) due to the IC internal current consumption. Once VCC reaches VCC(off), the HV current source turns on and CVCC begins to charge towards VCC(on). When VCC, reaches VCC(on), the cycle repeats until the fault is removed. Once the fault is removed, the NCP1341 is Fault Fault Applied Fault Removed Waits for next V CC(on) before starting VCC time V CC (on ) V CC (off ) time FDRV time IHV Istart 2 Istart (off) time Figure 20. Operation During Non-Latching Fault www.onsemi.com 23 NCP1341 Auto-recovery Timer Faults running, the HV current source turns on and off to maintain Vcc between Vcc(off) and Vcc(on). Once the auto-recovery timer expires, the controller will attempt to start normally at the next VCC(on) provided VHV is above VBO(start). This operation is shown in Figure 21. Some faults faults cause the NCP1341 auto-recovery timer to run. If an auto-recovery fault is detected, the gate drive is disabled and the auto-recovery timer, tautorec (typically 1.2 s), starts. While the auto-recovery timer is Fault Applied Fault Removed Fault time VCC VCC(on) VCC(off) Restarts At V CC (on ) ( new burst cycle if Fault still present ) DRV time Controller stops Autorecovery Timer time 1.2 s t restart Figure 21. Operation During Auto-Recovery Fault www.onsemi.com 24 NCP1341 PROTECTION FEATURES Brownout Protection Figure 22 shows the brownout detector waveforms during a brownout. When a brownout is detected, the controller stops switching and enters non-latching fault mode (see Figure 20). The HV current source alternatively turns on and off to maintain VCC between VCC(on) and VCC(off) until the input voltage is back above VBO(start). A timer is enabled once VHV drops below its disable threshold, VBO(stop) (typically 99 V). The controller is disabled if VHV doesn't exceed VBO(stop) before the brownout timer, tBO (typically 54 ms), expires. The timer is set long enough to ignore a two cycle dropout. The timer starts counting once VHV drops below VBO(stop). V HV V BO (start ) V BO (stop ) time Brownout Timer Brownout detected Starts Charging Immediately V CC V CC (on) Fault Cleared Restarts at next V CC(on) time V CC (off ) tdelay (start ) time DRV Figure 22. Operation During Brownout Line Removal Detection and X2 Capacitor Discharge time discharge circuitry. A novel approach is used to reconfigure the high voltage startup circuit to discharge the input filter capacitors upon removal of the ac line voltage. The line removal detection circuitry is always active to ensure safety compliance. The line removal is detected by digitally sampling the voltage present at the HV pin, and monitoring the slope. A timer, tline(removal) (typically 100 ms), is used to detect when the slope of the input signal is negative or below the resolution level. The timer is reset any time a positive slope Safety agency standards require the input filter capacitors to be discharged once the ac line voltage is removed. A resistor network is the most common method to meet this requirement. Unfortunately, the resistor network consumes power across all operating modes and it is a major contributor of input power losses during light-load and no-load conditions. The NCP1341 eliminates the need for external discharge resistors by integrating active input filter capacitor www.onsemi.com 25 NCP1341 is detected. Once the timer expires, a line removal condition is acknowledged initiating an X2 capacitor discharge cycle, and the controller is disabled. If VCC is above VCC(on), it is first discharged to VCC(on). A second timer, tline(discharge) (typically 32 ms), is used for the time limiting of the discharge phase to protect the device against overheating. Once the discharge phase is complete, tline(discharge) is reused while the device checks to see if the line voltage is reapplied. During the discharge phase, if VCC drops to VCC(on), it is quickly recharged to VCC(X2_reg). The discharging process is cyclic and continues until the ac line is detected again or the voltage across the X2 capacitor is lower than VHV(discharge) (30 V maximum). This feature allows the device to discharge large X2 capacitors in the input line filter to a safe level. It is important to note that the HV pin cannot be connected to any dc voltage due to this feature, i.e. directly to the bulk capacitor. X2 Capacitor Discharge VHV VBO(start ) VBO(stop) X2 Capacitor Discharge AC Line Unplug VHV(discharge ) time AC Timer Starts Timer tline(removal ) AC Timer Restarts AC Timer Expires No AC Detection tline(discharge /detect ) tline(removal ) DRV tline(discharge ) tline(detect ) X2 Discharge X2 Discharge Current Device is stopped Istart 2 ICC ICC(discharge ) 0 ICC3 Istart 2 VCC VCC(X2_reg) VCC(on) Figure 23. Line Removal Timing www.onsemi.com 26 tline(discharge ) X2 Discharge time NCP1341 X2 Capacitor Discharge VHV VBO(start ) VBO(stop ) AC Line Unplug VHV(discharge ) AC Timer Starts Timer tline (removal ) AC Timer Expires AC Timer Restarts time AC Detected tline(discharge /detect ) time X2 Discharge Device is stopped X2 Discharge Current time tline (discharge ) tline (removal ) DRV tdelay (start ) Istart 2 time ICC ICC(discharge ) 0 ICC3 Istart 2 time VCC VCC(X2_reg) VCC(on) Figure 24. Line Removal Timing with AC Reapplied the lower fault threshold, VFault(OTP_in) (typically 0.4 V). The lower threshold is normally used for detecting an overtemperature fault. The controller operates normally while the Fault pin voltage is maintained within the upper and lower fault thresholds. Figure 25 shows the architecture of the Fault input. The Fault input signal is filtered to prevent noise from triggering the fault detectors. Upper and lower fault detector blanking delays, tdelay(OVP) and tdelay(OTP),are both typically 30 ms. A fault is detected if the fault condition is asserted for a period longer than the blanking delay. An over temperature protection block monitors the junction temperature during the discharge process to avoid thermal runaway, in particular during open/short pins safety tests. Please note that the X2 discharge capability is also active at all times, including off-mode and before the controller actually starts to pulse (e.g. if the user unplugs the converter during the start-up sequence). Dedicated Fault Input The NCP1341 includes a dedicated fault input accessible via the Fault pin (8-pin and 9-pin versions only). The controller can be latched by pulling up the pin above the upper fault threshold, VFault(OVP) (typically 3.0 V). The controller is disabled if the Fault pin voltage is pulled below www.onsemi.com 27 NCP1341 OVP voltage drop across the thermistor. The resistance of the NTC thermistor decreases at higher temperatures resulting in a lower voltage across the thermistor. The controller detects a fault once the thermistor voltage drops below VFault(OTP_in). The controller bias current is reduced during power up by disabling most of the circuit blocks including IFault(OTP). This current source is enabled once VCC reaches VCC(on). A filter capacitor is typically connected between the Fault and GND pins. This will result in a delay before VFault reaches its steady state value once IFault(OTP) is enabled. Therefore, the lower fault comparator (i.e. overtemperature detection) is ignored during soft-start. Version A latches off the controller after an overtemperature fault is detected according to Figure 19. In Version B, the controller is re-enabled once the fault is removed such that VFault increases above VFault(OTP_out), the auto-recovery timer expires, and VCC reaches VCC(on) as shown in Figure 21. An active clamp prevents the Fault pin voltage from reaching the upper latch threshold if the pin is open. To reach the upper threshold, the external pull-up current has to be higher than the pull-down capability of the clamp (set by RFault(clamp) at VFault(clamp)), i.e., approximately 1 mA. The upper fault threshold is intended to be used for an overvoltage fault using a zener diode and a resistor in series from the auxiliary winding voltage. The controller is latched once VFault exceeds VFault(OVP). Once the controller is latched, it follows the behavior of a latching fault according to Figure 19 and is only reset if VCC is reduced to VCC(reset), or X2 discharge is activated. In the typical application these conditions occur only if the ac voltage is removed from the system. OTP The lower fault threshold is intended to be used to detect an overtemperature fault using an NTC thermistor. A pull up current source, IFault(OTP) (typically 45.5 mA), generates a Figure 25. Fault Pin Internal Schematic www.onsemi.com 28 NCP1341 * The controller latches off (versions A/C/E) or * Enters a safe, low duty-ratio auto-recovery mode Overload Protection The overload timer integrates the duration of the overload fault. That is, the timer count increases while the fault is present and reduces its count once it is removed. The overload timer duration, tOVLD, is typically 160 ms. When the overload timer expires, the controller detects an overload condition does one of the following: (versions B/D/F). Figure 26 shows the overload circuit schematic, while Figure 27 and Figure 28 show operating waveforms for latched and auto-recovery overload conditions. Count 4 Figure 26. Overload Circuitry www.onsemi.com 29 NCP1341 Fault Latch Event Latch time V CC V CC(on) V CC(off) time DRV time I HV Istart2 IHV(off) time Figure 27. Latched Overload Operation www.onsemi.com 30 NCP1341 Output Load Overcurrent applied Fault disappears Max Load time Fault Flag Fault timer starts time V CC V CC(on) V CC(off) Restarts At V CC ( on ( new burst cycle if Fault still present DRV ) time ) Controller stops time Fault timer 160 ms t OVLD t restart Figure 28. Auto-Recovery Overload Operation www.onsemi.com 31 t delay time ( start ) NCP1341 Abnormal Overcurrent Protection (AOCP) ZCD pin voltage when in PEM. At the turn-off of each drive cycle, the ZCD voltage swings high and triggers this comparator. Once the CCM timer expires, the next drive cycle will only start if the comparator has been triggered. During an output short-circuit, the aux winding voltage collapses, and the ZCD pin will not swing high enough to trip the comparator. Therefore, when the CCM timer expires the drive cycle will be delayed until demagnetization occurs, i.e. the controller will operate as if in QR mode. The short circuit protection block is shown in Figure 29. Under some severe fault conditions, like a winding short-circuit, the switch current can increase very rapidly during the on-time. The current sense signal significantly exceeds VILIM1, but because the current sense signal is blanked by the LEB circuit during the switch turn-on, the power switch current can become huge and cause severe system damage. The NCP1341 protects against this fault by adding an additional comparator for Abnormal Overcurrent Fault detection. The current sense signal is blanked with a shorter LEB duration, tLEB2, typically 125 ns, before applying it to the Abnormal Overcurrent Fault Comparator. The voltage threshold of the comparator, VILIM2, typically 1.5 V, is set 50% higher than VILIM1, to avoid interference with normal operation. Four consecutive Abnormal Overcurrent faults cause the controller to enter latch mode. The count to 4 provides noise immunity during surge testing. The counter is reset each time a DRV pulse occurs without activating the Fault Overcurrent Comparator. Current Sense Pin Failure Protection Figure 29. Short Circuit Protection A 1 mA (typically) pull-up current source, ICS, pulls up the CS pin to disable the controller if the pin is left open. Additionally, the maximum on-time, ton(MAX) (32 ms typically), prevents the MOSFET from staying on permanently if the CS Pin is shorted to GND. VCC Overvoltage Protection An additional comparator on the VCC pin monitors the VCC voltage. If VCC exceeds VCC(OVP), the gate drive is disabled and the NCP1341 follows the operation of a latching fault (see Figure 19). Output Short Circuit Protection During an output short-circuit, there is not enough voltage across the secondary winding to demagnetize the core. Due to the valley timeout feature of the controller, the flux level will quickly walk up until the core saturates. This can cause excessive stress on the primary MOSFET and secondary diode. This is not a problem for the NCP1341, however, because the valley timeout timer is disabled while the ZCD Pin voltage is above the arming threshold. Since the leakage energy is high enough to arm the ZCD trigger, the timeout timer is disabled and the next drive pulse is delayed until demagnetization occurs. In PEM, the next drive pulse is not triggered by demagnetization, but must also be delayed if there is a short-circuit on the output. To accomplish this, the PEM arming comparator, VPEM(arm) (1.5 V typical), monitors the Thermal Shutdown An internal thermal shutdown circuit monitors the junction temperature of the controller. The controller is disabled if the junction temperature exceeds the thermal shutdown threshold, TSHDN (typically 140C). When a thermal shutdown fault is detected, the controller enters a non-latching fault mode as depicted in Figure 20. The controller restarts at the next VCC(on) once the junction temperature drops below below TSHDN by the thermal shutdown hysteresis, TSHDN(HYS), typically 40C. The thermal shutdown is also cleared if VCC drops below VCC(reset), or a line removal fault is detected. A new power up sequence commences at the next VCC(on) once all the faults are removed. www.onsemi.com 32 NCP1341 TYPICAL CHARACTERISTICS 17.14 9 17.12 8.99 17.1 8.98 VCC(off) (V) VCC(on) (V) 17.08 17.06 17.04 17.02 17 8.97 8.96 8.95 16.98 8.94 16.96 16.94 -40 -20 0 20 40 60 80 100 120 8.93 -40 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 30. VCC(on) vs. Temperature Figure 31. VCC(off) vs. Temperature 0.6 120 5 4.5 0.5 4 3.5 Istart2 (mA) Istart1 (mA) 0.4 0.3 0.2 3 2.5 2 1.5 1 0.1 0.5 0 -40 -20 0 20 40 60 80 100 0 -40 120 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 32. Istart1 vs. Temperature Figure 33. Istart2 vs. Temperature 7 120 9 8 6 7 IHV(off2) (mA) IHV(off1) (mA) 5 4 3 2 6 5 4 3 2 1 0 -40 1 -20 0 20 40 60 80 100 120 0 -40 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 34. IHV(off1) vs. Temperature Figure 35. IHV(off2) vs. Temperature www.onsemi.com 33 120 NCP1341 TYPICAL CHARACTERISTICS 0.126 0.255 0.124 0.250 0.122 0.245 0.118 ICC2 (mA) ICC1 (mA) 0.120 0.116 0.114 0.112 0.240 0.235 0.230 0.110 0.225 0.108 0.106 -40 -20 0 20 40 60 80 100 0.220 -40 120 -20 0 20 40 60 80 100 120 100 120 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 36. ICC1 vs. Temperature Figure 37. ICC2 vs. Temperature 1.075 28.35 1.070 28.3 1.065 VCC(OVP) (V) ICC3 (mA) 1.060 1.055 1.050 1.045 1.040 28.25 28.2 28.15 1.035 1.030 -40 -20 0 20 40 60 80 100 28.1 -40 120 20 40 60 80 TJ, JUNCTION TEMPERATURE (C) Figure 38. ICC3 vs. Temperature Figure 39. VCC(OVP) vs. Temperature 19.8 112.6 112.4 19.4 112.2 19.2 19 VBO(start) (V) ICC(discharge) (mA) 0 TJ, JUNCTION TEMPERATURE (C) 19.6 18.8 18.6 18.4 18.2 112 111.8 111.6 111.4 111.2 18 110 17.8 17.6 -40 -20 -20 0 20 40 60 80 100 120 110.8 -40 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 40. ICC(discharge) vs. Temperature Figure 41. VBO(start) vs. Temperature www.onsemi.com 34 120 NCP1341 TYPICAL CHARACTERISTICS 98.2 90 80 98 CDRV = 1 nF 97.8 tDRV(rise) (ns) VBO(stop) (V) 70 97.6 97.4 60 50 40 30 CDRV = 100 pF 20 97.2 97 -40 10 -20 0 20 40 60 80 100 0 -40 120 60 80 100 TJ, JUNCTION TEMPERATURE (C) 120 111.8 111.6 CDRV = 1 nF 111.4 30 fMAX1 (kHz) tDRV(fall) (ns) 40 Figure 43. tDRV(rise) vs. Temperature 35 25 20 15 111.2 111 110.8 110.6 10 CDRV = 100 pF 110.4 5 -20 0 20 40 60 80 100 110.2 -40 120 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 44. tDRV(fall) vs. Temperature Figure 45. fMAX1 vs. Temperature 367 120 73.45 366.5 73.4 366 73.35 365.5 73.3 fMAX3 (kHz) fMAX2 (kHz) 20 TJ, JUNCTION TEMPERATURE (C) 40 365 364.5 364 73.25 73.2 73.15 363.5 73.1 363 73.05 362.5 -40 0 Figure 42. VBO(stop) vs. Temperature 45 0 -40 -20 -20 0 20 40 60 80 100 120 73 -40 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 46. fMAX2 vs. Temperature Figure 47. fMAX3 vs. Temperature www.onsemi.com 35 120 NCP1341 TYPICAL CHARACTERISTICS 32.5 63.6 32.4 63.5 VZCD(trig) (mV) ton(MAX) (ms) 32.3 32.2 32.1 32 63.4 63.3 63.2 31.9 63.1 31.8 31.7 -40 -20 0 20 40 60 80 100 63 -40 120 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 49. VZCD(trig) vs. Temperature 120 12.95 25.6 12.9 25.55 VZCD(MAX) (V) VZCD(HYS) (mV) 0 Figure 48. ton(MAX) vs. Temperature 25.65 25.5 25.45 12.85 12.8 12.75 25.4 25.35 -40 -20 -20 0 20 40 60 80 100 12.7 -40 120 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 50. VZCD(HYS) vs. Temperature Figure 51. VZCD(MAX) vs. Temperature 0 120 198.8 -0.1 198.6 -0.3 Vfreeze (mV) VZCD(MIN) (V) -0.2 -0.4 -0.5 -0.6 -0.7 198.4 198.2 198 197.8 -0.8 -0.9 -40 -20 0 20 40 60 80 100 120 197.6 -40 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 52. VZCD(MIN) vs. Temperature Figure 53. Vfreeze vs. Temperature www.onsemi.com 36 120 NCP1341 TYPICAL CHARACTERISTICS 1.31 104.2 1.308 104 103.8 1.304 Vjitter (mV) fjitter (kHz) 1.306 1.302 1.3 103.2 1.296 103 -40 -20 0 20 40 60 80 100 102.8 -40 120 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) Figure 55. Vjitter vs. Temperature 402.5 3.09 402 3.08 401.5 3.07 3.06 3.05 120 401 400.5 400 399.5 3.04 -20 0 20 40 60 80 100 399 -40 120 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 56. VFault(OVP) vs. Temperature Figure 57. VFault(OTP_in) vs. Temperature 920 45.1 918 45 120 44.9 IOTP (mA) 916 914 912 910 44.8 44.7 44.6 44.5 908 906 -40 0 TJ, JUNCTION TEMPERATURE (C) 3.1 3.03 -40 -20 Figure 54. fjitter vs. Temperature VFault(OTP_in) (mV) VFault(OVP) (V) 103.4 1.298 1.294 VFault(OTP_out) (mV) 103.6 44.4 -20 0 20 40 60 80 100 120 44.3 -40 -20 0 20 40 60 80 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 58. VFault(OTP_out) vs. Temperature Figure 59. IOTP vs. Temperature www.onsemi.com 37 100 120 NCP1341 TYPICAL CHARACTERISTICS 1.731 1.55 1.545 1.54 RFault(clamp) (kW) VFault(clamp) (V) 1.73 1.729 1.728 1.535 1.53 1.525 1.52 1.515 1.51 1.727 1.505 1.726 -40 1.495 -40 1.5 -20 0 20 40 60 80 100 120 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 60. VFault(clamp) vs. Temperature Figure 61. RFault(clamp) vs. Temperature 24.5 120 1.39 24.45 1.385 24.4 24.3 tquiet (ms) fMIN (kHz) 24.35 24.25 24.2 1.38 1.375 24.15 1.37 24.1 24.05 -20 0 20 40 60 80 100 1.365 -40 120 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) Figure 62. fMIN vs. Temperature Figure 63. tquiet vs. Temperature 3.2 1.001 3.18 1 3.16 0.999 120 0.998 3.14 3.12 3.1 0.997 0.996 0.995 3.08 0.994 3.06 3.04 -40 -20 TJ, JUNCTION TEMPERATURE (C) VILIM1 (V) tZCD(blank) (ms) 24 -40 0.993 -20 0 20 40 60 80 100 120 0.992 -40 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 64. tZCD(blank) vs. Temperature Figure 65. VILIM1 vs. Temperature www.onsemi.com 38 120 NCP1341 TYPICAL CHARACTERISTICS 1.503 40.2 1.502 40.1 1.501 40 1.499 tDT(MAX) (ms) VILIM2 (V) 1.5 1.498 1.497 1.496 39.9 39.8 39.7 39.6 1.495 39.5 1.494 39.4 1.493 1.492 -40 -20 0 20 40 60 80 100 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) 0.801 VPEM (Versions A/B/C/D) (V) 296.5 295.5 295 294.5 120 0.8 0.799 0.798 0.797 0.796 -20 0 20 40 60 80 100 0.795 -40 120 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 68. Vskip vs. Temperature Figure 69. VPEM (Versions A/B/C/D) vs. Temperature 0.6705 120 2.905 0.67 2.9 0.6695 2.895 0.669 2.89 Kscale(MAX) VPEM (Versions E/F) (V) 20 Figure 67. tDT(MAX) vs. Temperature 0.802 0.6685 0.668 0.6675 2.885 2.88 2.875 0.667 2.87 0.6665 2.865 0.666 2.86 0.6655 -40 0 TJ, JUNCTION TEMPERATURE (C) 297 294 -40 -20 Figure 66. VILIM2 vs. Temperature 296 Vskip (mV) 39.3 -40 120 -20 0 20 40 60 80 100 120 2.855 -40 -20 0 20 40 60 80 100 TJ, JUNCTION TEMPERATURE (C) TJ, JUNCTION TEMPERATURE (C) Figure 70. VPEM (Versions E/F) vs. Temperature Figure 71. Kscale(MAX) vs. Temperature www.onsemi.com 39 120 NCP1341 PACKAGE DIMENSIONS SOIC-8 NB CASE 751-07 ISSUE AK NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. DIMENSION A AND B DO NOT INCLUDE MOLD PROTRUSION. 4. MAXIMUM MOLD PROTRUSION 0.15 (0.006) PER SIDE. 5. DIMENSION D DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.127 (0.005) TOTAL IN EXCESS OF THE D DIMENSION AT MAXIMUM MATERIAL CONDITION. 6. 751-01 THRU 751-06 ARE OBSOLETE. NEW STANDARD IS 751-07. -X- A 8 5 S B 0.25 (0.010) M Y M 1 4 K -Y- G C N DIM A B C D G H J K M N S X 45 _ SEATING PLANE -Z- 0.10 (0.004) H M D 0.25 (0.010) M Z Y S X J S SOLDERING FOOTPRINT* 1.52 0.060 7.0 0.275 4.0 0.155 0.6 0.024 1.270 0.050 SCALE 6:1 mm inches *For additional information on our Pb-Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D. www.onsemi.com 40 MILLIMETERS MIN MAX 4.80 5.00 3.80 4.00 1.35 1.75 0.33 0.51 1.27 BSC 0.10 0.25 0.19 0.25 0.40 1.27 0_ 8_ 0.25 0.50 5.80 6.20 INCHES MIN MAX 0.189 0.197 0.150 0.157 0.053 0.069 0.013 0.020 0.050 BSC 0.004 0.010 0.007 0.010 0.016 0.050 0 _ 8 _ 0.010 0.020 0.228 0.244 NCP1341 PACKAGE DIMENSIONS SOIC-9 NB CASE 751BP ISSUE A 2X 0.10 C A-B D D A 0.20 C 2X 0.10 C A-B 4 TIPS 10 F 6 H E 1 5 0.20 C 9X B 5 TIPS L2 b 0.25 A3 L C SEATING PLANE DETAIL A NOTES: 1. DIMENSIONING AND TOLERANCING PER ASME Y14.5M, 1994. 2. CONTROLLING DIMENSION: MILLIMETERS. 3. DIMENSION b DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE PROTRUSION SHALL BE 0.10mm TOTAL IN EXCESS OF 'b' AT MAXIMUM MATERIAL CONDITION. 4. DIMENSIONS D AND E DO NOT INCLUDE MOLD FLASH, PROTRUSIONS, OR GATE BURRS. MOLD FLASH, PROTRUSIONS, OR GATE BURRS SHALL NOT EXCEED 0.15mm PER SIDE. DIMENSIONS D AND E ARE DETERMINED AT DATUM F. 5. DIMENSIONS A AND B ARE TO BE DETERMINED AT DATUM F. 6. A1 IS DEFINED AS THE VERTICAL DISTANCE FROM THE SEATING PLANE TO THE LOWEST POINT ON THE PACKAGE BODY. C A-B D M TOP VIEW 9X h X 45 _ 0.10 C 0.10 C M A A1 e C DETAIL A SEATING PLANE END VIEW SIDE VIEW DIM A A1 A3 b D E e H h L L2 M MILLIMETERS MIN MAX 1.25 1.75 0.10 0.25 0.17 0.25 0.31 0.51 4.80 5.00 3.80 4.00 1.00 BSC 5.80 6.20 0.37 REF 0.40 1.27 0.25 BSC 0_ 8_ RECOMMENDED SOLDERING FOOTPRINT* 9X 1.00 PITCH 0.58 6.50 9X 1.18 1 DIMENSION: MILLIMETERS *For additional information on our Pb-Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D. ON Semiconductor and are trademarks of Semiconductor Components Industries, LLC dba ON Semiconductor or its subsidiaries in the United States and/or other countries. ON Semiconductor owns the rights to a number of patents, trademarks, copyrights, trade secrets, and other intellectual property. A listing of ON Semiconductor's product/patent coverage may be accessed at www.onsemi.com/site/pdf/Patent-Marking.pdf. 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