MIC2164/-2/-3/C
Synchronous Buck Controllers
Featuring Adaptive On-Time Control
28V Input, Constant Frequency
Hyper Speed Control™ Family
Hype
408) 474-1000 • http://www.micrel.com
r Speed Control and Any Capacitor is a trademark of Micrel, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (
Septembe M9999-091310-D
r 2010
General Description
The Micrel MIC2164/-2/-3/C are constant-frequency,
synchronous buck controllers featuring adaptive on-time
control. The MIC2164/-2/-3/C are the first products in the new
Hyper Speed Control™ family of buck controllers introduced
by Micrel.
The MIC2164/-2/-3/C controllers operate over an input supply
range of 3V to 28V, and are independent of the IC supply
voltage. The devices are capable of supplying 25A output
current. While the MIC2164 operates at 300kHz, the
MIC2164-2 operates at 600kHz, and the MIC2164-3 operates
at 1MHz.
A unique Hyper Speed Control™ architecture allows for ultra-
fast transient response while reducing the output capacitance
and also makes High VIN/Low VOUT operation possible. The
MIC2164/-2/-3/C controllers utilizes an architecture which is
adaptive TON ripple controlled. A UVLO feature is provided to
ensure proper operation under power-sag conditions to
prevent the external power MOSFET from overheating. A soft
start feature is provided to reduce the inrush current.
Foldback current limit and “hiccup” mode short-circuit
protection ensure FET and load protection.
The MIC2164/-2/-3/C controllers are available in a 10-pin
MSOP (MAX1954A-compatible) package with a junction
operating range from –40°C to +125°C.
All support documentation can be found on Micrel’s web site
at: www.micrel.com.
Features
• Hyper Speed Control™ architecture enables
- High delta V operation (VHSD = 28V and VOUT = 0.8V)
- Smaller output capacitors than competitors
• 3V to 28V input voltage
• Any CapacitorTM stable
- Zero ESR to high ESR
• 25A output current capability
• 300kHz/600kHz/1MHz switching frequency
• Adaptive on-time mode control
• Adjustable output from 0.8V to 5.5V with ±1%
(MIC2164/-2/-3) or ±3% (MIC2164C) FB accuracy
• Up to 95% efficiency
• Foldback current-limit and “hiccup” mode short-circuit
protection
• 6ms Internal soft start
• Thermal shutdown
• Safe start-up into pre-biased loads
• –40°C to +125°C junction temperature range
• Available in 10-pin MSOP package
Applications
• Set-top box, gateways and routers
• Printers, scanners, graphic cards and video cards
• Telecommunication, PCs and servers
________________________________________________________________________________________________________________________
Typical Application
MIC2164/-2/-3/C Synchronous Controllers Featuring Adaptive On-Time Control
MIC2164
12V to 3. 3V Eff ici ency
50
55
60
65
70
75
80
85
90
95
100
0481216
OUTPUT CURRENT (A)
EFFICIENCY (%)
20
VIN=5V
Micrel, Inc. MIC2164/-2/-3/C
September 2010 2 M9999-091310-D
Ordering Information
Part Number Voltage Swit ching
Frequency Accuracy Junction Temperature Range Package Lead Finish
MIC2164YMM Adjustable 300kHz ±1% –40° to +125°C 10-pin MSOP Pb-Free
MIC2164-2YMM Adjustable 600kHz ±1% –40° to +125°C 10-pin MSOP Pb-Free
MIC2164-3YMM Adjustable 1MHz ±1% –40° to +125°C 10-pin MSOP Pb-Free
MIC2164CYMM Adjustable 270kHz ±3% –40° to +125°C 10-pin MSOP Pb-Free
Pin Configur ation
10-Pin MSOP (MM)
Pin Description
Pin Number Pin Name Pin Function
1 HSD
High-Side N-MOSFET Drain Connection (input): Power to the drain of the external high-side N-channel
MOSFET. The HSD operating voltage range is from 3V to 28V. Input capacitors between HSD and the
power ground (PGND) are required.
2 EN
Enable (input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or floating
= enable, logic low = shutdown. In the off state, supply current of the device is greatly reduced (typically
0.8mA).
3 FB
Feedback (input): Input to the transconductance amplifier of the control loop. The FB pin is regulated to
0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output
voltage.
4 GND
Signal ground. GND is the ground path for the device input voltage VIN and the control circuitry. The loop
for the signal ground should be separate from the power ground (PGND) loop.
5 IN
Input Voltage (input): Power to the internal reference and control sections of the MIC2164/-2/-3. The IN
operating voltage range is from 3V to 5.5V. A 1µF and 0.1µF ceramic capacitors from IN to GND are
recommended for clean operation.
6 DL
Low-Side Drive (output): High-current driver output for external low-side MOSFET. The DL driving
voltage swings from ground-to-IN.
7 PGND
Power Ground. PGND is the ground path for the MIC2164/-2/-3 buck converter power stage. The PGND
pin connects to the sources of low-side N-Channel MOSFETs, the negative terminals of input capacitors,
and the negative terminals of output capacitors. The loop for the power ground should be as small as
possible and separate from the Signal ground (GND) loop.
8 DH
High-Side Drive (output): High-current driver output for external high-side MOSFET. The DH driving
voltage is floating on the switch node voltage (LX). It swings from ground to VIN minus the diode drop.
Adding a small resistor between DH pin and the gate of the high-side N-channel MOSFETs can slow
down the turn-on and turn-off time of the MOSFETs.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 3 M9999-091310-D
Pin Description (Continued)
Pin Number Pin Name Pin Function
9 LX
Switch Node and Current Sense input: High current output driver return. The LX pin connects directly
to the switch node. Due to the high speed switching on this pin, the LX pin should be routed away from
sensitive nodes. LX pin also senses the current by monitoring the voltage across the low-side
MOSFET during OFF time. In order to sense the current accurately, connect the low-side MOSFET
drain to LX using a Kelvin connection.
10 BST
Boost (output): Bootstrapped voltage to the high-side N-channel MOSFET driver. A Schottky diode is
connected between the IN pin and the BST pin. A boost capacitor of 0.1F is connected between the
BST pin and the LX pin. Adding a small resistor in series with the boost capacitor can slow down the
turn-on time of high-side N-Channel MOSFETs.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 4 M9999-091310-D
Absolute Maximum Ratings(1)
IN, FB, EN to GND .......................................... −0.3V to +6V
BST to LX ....................................................... −-0.3V to +6V
BST to GND .................................................. −0.3V to +37V
DH to LX............................................−0.3V to (VBST + 0.3V)
DL, COMP to GND.............................. −0.3V to (VIN + 0.3V)
HSD to GND.................................................... −0.3V to 31V
PGND to GND.............................................. −0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10sec)........................ 260°C
Operating Ratings(2)
Input Voltage (VIN)............................................ 3.0V to 5.5V
Supply Voltage (VHSD) ....................................... 3.0V to 28V
Operating Temperature Range ................. −40°C to +125°C
Junction Temperature (TJ) ........................ −40°C to +125°C
Junction Thermal Resistance
MSOP (θJA) ..................................................130.5°C/W
Continuous Power Dissipation (TA = 70°C)..............421mW
(derate 5.6mW/°C above 70°C)
Electrical Characteristics(4)
VBST − VLX = 5V; TA = +25°C, unless noted. Bold values indicate −40°C TJ +125°C.
Parameter Condition Min. Typ. Max. Units
General
Operating Input Voltage (VIN) (5) 3.0 5.5 V
HSD Voltage Range (VHSD) 3.0 28 V
Quiescent Supply Current (VFB = 1.5V, output switching but excluding
external MOSFET gate current) 1.4
3.0 mA
Standby Supply Current (6) VIN = VBST = 5.5V, VHSD = 28, LX = unconnected,
EN = GND 0.8
2 mA
Under-Voltage Lockout Trip Level 2.4 2.7 3 V
UVLO Hysteresis 50 mV
DC-DC Controller
Output-Voltage Adjust Range
(VOUT) 0.8 5.5 V
Error Amplifier
0°C TJ 85°C -1 1
−40°C TJ 125°C -2 2
FB Regulation Voltage
TJ = 25°C (MIC2164C) -3 3
%
FB Input Leakage Current 5 500 nA
VFB = 0.8V 103 130 162
VFB = 0V 19 48 77
VFB = 0.8V (MIC2164C) 95 130 170
Current-Limit Threshold
VFB = 0V (MIC2164C) 15 48 80
mV
Soft-Start
Soft-Start Period 6 ms
Oscillator
Switching Frequency (8)
MIC2164C
MIC2164
MIC2164-2
MIC2164-3
0.202
0.225
0.45
0.75
0.27
0.3
0.6
1
0.338
0.375
0.75
1.25
MHz
Micrel, Inc. MIC2164/-2/-3/C
September 2010 5 M9999-091310-D
Electrical Characteristics(4) (Continued)
VBST − VLX = 5V; TA = +25°C, unless noted. Bold values indicate −40°C TJ +125°C.
Parameter Condition Min. Typ. Max. Units
Maximum Duty Cycle (9)
MIC2164/ MIC2164C
MIC2164-2
MIC2164-3
87
74
66
%
Minimum Duty Cycle Measured at DH, VFB = 1V 0 %
FET Drives
DH, DL Output Low Voltage ISINK = 10mA 0.1 V
DH, DL Output High Voltage ISOURCE = 10mA
VIN − 0.1V
or
VBST − 0.1V
V
DH On-Resistance, High State 2.1 3.3
DH On-Resistance, Low State 1.8 3.3
DL On-Resistance, High State 1.8 3.3
DL On-Resistance, Low State 1.2 2.3
LX Leakage Current VLX = 28V, VIN = 5.5V,VBST = 33.5V 50 µA
HSD Leakage Current VLX = 28V, VIN = 5.5V,VBST = 33.5V 20 µA
Thermal Protection
Over-Temperature Shutdown 155 °C
Over-Temperature Shutdown
Hysteresis 10 °C
Shutdown Control
EN Logic Level Low 3V < VIN <5.5V 0.4 0.8 V
EN Logic Level High 3V < VIN <5.5V 0.9 1.2 V
EN Pull-Up Current 50
µA
Note:
1. Exceeding the absolute maximum rating may damage the device.
2. The device is not guaranteed to function outside its operating rating.
3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
4. Specification for packaged product only.
5. The application is fully functional at low IN (supply of the control section) if the external MOSFETs have enough low voltage VTH.
6. The current will come only from the internal 100k pull-up resistor sitting on the EN Input and tied to IN.
8. Measured in test mode.
9. Measured at DH. The maximum duty cycle is limited by the fixed mandatory off time TOFF of typical 363ns.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 6 M9999-091310-D
Typical Characteristics
MIC2164
12V t o 1.5V E f f iciency
50
55
60
65
70
75
80
85
90
95
100
0481216
OUTPUT CURRENT (A)
EFFICIE NCY (%)
20
VIN=5V
MIC2164
12V t o 3.3V E f f iciency
50
55
60
65
70
75
80
85
90
95
100
0481216
OUTPUT CURRENT (A)
EFFICIE NCY (%)
20
VIN=5V
MIC2164-2
12V t o 1.5V E f f iciency
50
55
60
65
70
75
80
85
90
95
100
0 3 6 9 12 15
OUTPUT CURRENT (A)
EFFICIE NCY (%)
VIN=5V
MIC2164-2
12V t o 3.3V E f f iciency
50
55
60
65
70
75
80
85
90
95
100
0 3 6 9 12 15
OUTPUT CURRENT (A)
EFFICIENCY (%)
VIN=5V
MIC2164-3
12V t o 1.5V E f f iciency
50
55
60
65
70
75
80
85
90
95
100
0246 8
OUTPUT CURRENT (A)
EFFICIENCY (%)
10
VIN=5V
MIC2164-3
12V t o 3.3V E f f iciency
50
55
60
65
70
75
80
85
90
95
100
0246 8
OUTPUT CURRENT (A)
EFFICIENCY (%)
10
VIN=5V
Feedback Voltage vs. Load
0.75
0.76
0.77
0.78
0.79
0.80
0.81
0.82
0.83
0.84
0.85
0481216
OUTPUT CURRENT (A)
FEEDBACK VOLT AGE (V
20
)
VHSD =12V
VIN=5V
Feedback Voltage
vs. Input Voltage
0.75
0.76
0.77
0.78
0.79
0.80
0.81
0.82
0.83
0.84
0.85
3 3.5 4 4.5 5 5.5
INPUT V OLTAGE (V)
FEEDBACK VOL TAGE (V
)
VHSD=12V
Feedback Voltage
vs. HSD Voltage
0.75
0.76
0.77
0.78
0.79
0.80
0.81
0.82
0.83
0.84
0.85
3 8 13 18 23 28
HSD VOLTAGE (V)
FEEDBACK VOLT AGE (V
)
VIN=5V
Feedback Voltage
vs. Temperature
0.790
0.792
0.794
0.796
0.798
0.800
0.802
0.804
0.806
0.808
0.810
-40-20 0 20406080100120
TEMPERATURE (°C)
FEEDBACK VOLT AGE (V
)
VIN=5V
MIC2164
Switching Frequency vs . Lo a d
250
260
270
280
290
300
310
320
330
340
350
0481216
OUTPUT CURRENT (A)
SWI TCHING FREQUENCY ( kHz)
20
VHSD=12V
VIN=5V
MIC2164-2
Switching Frequency vs. Load
500
520
540
560
580
600
620
640
660
680
700
0 3 6 9 12 15
OUTPUT CURRENT (A)
SW ITC H ING FR E QU ENCY (k Hz)
VHSD=12V
VIN=5V
Micrel, Inc. MIC2164/-2/-3/C
September 2010 7 M9999-091310-D
Typical Characteristics (Continued)
MIC2164-3
Switching Frequency vs. Loa d
850
880
910
940
970
1000
1030
1060
1090
1120
1150
02468
OUTPUT CURRENT (A)
SWITC HING F RE QUENCY ( kHz)
10
VHSD=12V
VIN=5V
MIC2164
Switching Frequency vs. VIN
250
260
270
280
290
300
310
320
330
340
350
3 3.5 4 4.5 5 5.5
INPUT VOLTAGE (V)
SWITCH ING FR E QUENCY ( kHz)
VHSD=12V
MIC2164-2
Switching Frequency vs. VIN
500
520
540
560
580
600
620
640
660
680
700
3 3.5 4 4.5 5 5.5
INPUT VOLTAGE (V)
SWITCH ING FR E QUENCY ( kHz)
VHSD=12V
MIC2164-3
Switching Frequency vs. VIN
850
880
910
940
970
1000
1030
1060
1090
1120
1150
33.5 44.5 55
INPUT V OLTAGE (V )
SWITCHI NG FREQUENCY (kHz)
.5
VHSD=12V
MIC2164
Switching Frequency v s . VHSD
250
260
270
280
290
300
310
320
330
340
350
3 8 13 18 23 28
HSD VOLTAGE (V)
SWIT CH ING FR E QUENCY ( kHz)
VIN=5V
MIC2164-2
Switching Frequency v s . VHSD
500
520
540
560
580
600
620
640
660
680
700
3 8 13 18 23 28
HSD VOLTAGE (V)
SWITCHING FREQUENCY (kHz)
VIN=5V
VOUT=2.5V
MIC2164-3
Switching Frequency v s . VHSD
850
3 8 13 18 23 28
HSD VOLTAGE (V)
MIC2164 Switching Frequency
vs. Temperature
250
-40 -20 0 20 40 60 80 100 120
TEMPERATURE (°C)
MIC2164-2 Switching
Frequenc y vs. Te m pe rature
500
-40 -20 0 20 40 60 80 100 120
TEMPERATURE (°C)
260
270
280
290
300
310
320
330
340
350
SWITC HING F RE QUENCY ( kHz)
520
540
560
580
600
620
640
660
680
700
SWITC HING F RE QUENCY ( kHz)
880
910
940
970
1000
1030
1060
1090
1120
1150
SWIT CH ING FR E QUENCY ( kHz)
VIN=5V
VIN= 5V VIN=5V
MIC2164-3 Switching
Frequenc y vs. Te m pe rature
880
910
940
970
1000
1030
1060
1090
1120
1150
SWITC HING F RE QUENCY ( kHz)
850
-40 -20 0 20 40 60 80 100 120
TEMPERATURE (°C)
VIN=5V
Current Limit Threshold vs .
Feedback Voltage Percentage
15
30
45
60
75
90
105
120
135
150
CURRENT LIMIT THRESHOLD
(mV)
0
0 102030405060708090100
Feedback Voltage Percentage (%)
VIN=5V
Current Limit Threshold
vs. Temperature
15
30
45
60
75
90
105
120
135
150
CURRENT LIMIT THRES HOLD
(mV)
0
-40-20 0 20406080100120
TEMPERATURE (°C)
VFB=0.8V
VFB=0V
Micrel, Inc. MIC2164/-2/-3/C
September 2010 8 M9999-091310-D
Typical Characteristics (Continued)
Quiescent Supply Current
vs. Input Voltage
0
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
1.8
2
33.544.555
INPUT VOLTAGE (V )
QUIESCENT SUPPLY
CURRENT (mA)
.5
Micrel, Inc. MIC2164/-2/-3/C
September 2010 9 M9999-091310-D
Functional Characteristics
MIC2164-2 Load Transient
Vout
(200mV/div)
AC-coupled
Iout
(5A/div)
Vhsd = 12V
Vin = 5V
Vout = 3.3V
L = 1.0H
Cout = 1100F
Time 200s/div
MIC2164-3 Load Transient
Vout
(200mV/div)
AC-coupled
Iout
(5A/div)
Vhsd = 12V
Vin = 5V
Vout = 3.3V
L = 1.0H
Cout = 660F
Time 200s/div
Micrel, Inc. MIC2164/-2/-3/C
September 2010 10 M9999-091310-D
Functional Characteristics (Continued)
Micrel, Inc. MIC2164/-2/-3/C
September 2010 11 M9999-091310-D
Functional Characteristics (continue)
MIC2164-2 Output Volt age Ripple
IL
(5A/div)
Vhsd = 12V
Vin = 5V
Vout = 3.3V
Iout = 15A
Time 2μs/div
VOUT
(50mV/div)
AC-coupled
L = 1.0H
Cout = 1100F
Micrel, Inc. MIC2164/-2/-3/C
September 2010 12 M9999-091310-D
Functional Diagram
Figure 1. MIC2164/-2/-3 Block Diagram
Micrel, Inc. MIC2164/-2/-3/C
September 2010 13 M9999-091310-D
Functional Description
The MIC2164/-2/-3 is an adaptive on-time synchronous
buck controller family built for low cost and high
performance. They are designed for a wide input voltage
range from 3V to 28V and for high output power buck
converters. An estimated-ON-time method is applied in
MIC2164/-2/-3 to obtain a constant switching frequency
and to simplify the control compensation. The over-
current protection is implemented without the use of an
external sense resistor. It includes an internal soft-start
function which reduces the power supply input surge
current at start-up by controlling the output voltage rise
time.
Theory of Operation
The MIC2164/-2/-3 is a adaptive on-time buck controller
family. Figure 1 illustrates the block diagram for the
control loop. The output voltage variation will be sensed
by the MIC2164/-2/-3 feedback pin FB via the voltage
divider R1 and R2, and compared to a 0.8V reference
voltage VREF at the error comparator through a low gain
transconductance (gm) amplifier, the amplifier improves
the MIC2164/-2/-3 converter output voltage regulation. If
the FB voltage decreases and the output of the gm
amplifier is below 0.8V, The error comparator will trigger
the control logic and generate an ON-time period, in
which DH pin is logic high and DL pin is logic low. The
ON-time period length is predetermined by the “FIXED
TON ESTIMATION” circuitry:
swHSD
OUT
ed)ON(estimat fV
V
T×
= (1)
where VOUT is the output voltage, VHSD is the power
stage input voltage, and fSW is the switching frequency
(300kHz for MIC2164, 600kHz for MIC2164-2, and
1MHz for MIC2164-3).
After ON-time period, the MIC2164/-2/-3 goes into the
OFF-time period, in which DH pin is logic low and DL pin
is logic high. The OFF-time period length is depending
on the FB voltage in most cases. When the FB voltage
decreases and the output of the gm amplifier is below
0.8V, then the ON-time period is trigger and the OFF-
time period ends. If the OFF-time period decided by the
FB voltage is less than the minimum OFF time TOFF(min),
which is about 363ns typical, then the MIC2164/-2/-3
control logic will apply the TOFF(min) instead. TOFF(min) is
required by the BST charging.
The maximum duty cycle is obtained from the 363ns
TOFF(min):
SS
OFF(min)S
T
363ns
1
T
TT
Dmax −=
−
=
where Ts = 1/fSW.
It is not recommended to use MIC2164/-2/-3 with a OFF
time close to TOFF(min) at the steady state. Also, as VOUT
increases, the internal ripple injection will increase and
reduce the line regulation performance. Therefore, the
maximum output voltage of the MIC2164/-2/-3 should be
limited to 5.5V. If a higher output voltage is required, use
the MIC2176 instead. Please refer to “Setting Output
Voltage” subsection in “Application Information” for more
details.
The estimated-ON-time method results in a constant
switching frequency in MIC2164/-2/-3. The actual ON
time is varied with the different rising and falling time of
the external MOSFETs. Therefore, the type of the
external MOSFETs, the output load current, and the
control circuitry power supply VIN will modify the actual
ON time and the switching frequency. Also, the minimum
TON results in a lower switching frequency in the high
VHSD and low VOUT applications, such as 24V to 1.0V
MIC2164-3 application. The minimum TON measured on
the MIC2164 evaluation board is about 138ns. During
the load transient, the switching frequency is changed
due to the varying OFF time.
To illustrate the control loop, the steady-state scenario
and the load transient scenario are analyzed. For easy
analysis, the gain of the gm amplifier is assumed to be 1.
With this assumption, the inverting input of the error
comparator is the same as the FB voltage. Figure 2
shows the MIC2164/-2/-3 control loop timing during the
steady-state. During the steady-state, the gm amplifier
senses the FB voltage ripple, which is proportional to the
output voltage ripple and the inductor current ripple, to
trigger the ON-time period. The ON time is
predetermined by the estimation. The ending of OFF
time is controlled by the FB voltage. At the valley of the
FB voltage ripple, which is below than VREF, OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 14 M9999-091310-D
Figure 2. MIC2164/-2/-3 Control Loop Timing
Figure 3 shows the load transient scenario of the
MIC2164/-2/-3 converter. The output voltage drops due
to the sudden load increasing, which would cause the
FB voltage to be less than VREF. This will cause the error
comparator to trigger ON-time period. At the end of the
ON-time period, a minimum OFF time TOFF(min) is
generated to charge BST since the FB voltage is still
below the VREF. Then, the next ON-time period is
triggered due to the low FB voltage. Therefore, the
switching frequency changes during the load transient.
With the varying duty cycle and switching frequency, the
output recovery time is fast and the output voltage
deviation is small in MIC2164/-2/-3 converter.
Figure 3. MIC2164/-2/-3 Load-Transient Response
Unlike the current-mode control, MIC2164/-2/-3 uses the
output voltage ripple, which is proportional to the
inductor current ripple if the ESR of the output capacitor
is large enough, to trigger an ON-time period. The
predetermined ON time makes MIC2164/-2/-3 control
loop has the advantage as the adaptive on-time mode
control. Therefore, the slope compensation, which is
necessary for the current-mode control, is not required in
the MIC2164/-2/-3 family.
The MIC2164/-2/-3 family has its own stability concern:
the FB voltage ripple should be in phase with the
inductor current ripple and large enough to be sensed by
the gm amplifier and the error comparator. The
recommended minimum FB voltage ripple is 20mV. If a
low ESR output capacitor is selected, then the FB
voltage ripple may be too small to be sensed by the gm
amplifier and the error comparator. Also, the output
voltage ripple and the FB voltage ripple are not in phase
with the inductor current ripple if the ESR of the output
capacitor is very low. Therefore, the ripple injection is
required for a low ESR output capacitor. Please refer to
“Ripple Injection” subsection in “Application Information”
for more details about the ripple injection.
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
MIC2164/-2/-3 implements an internal digital soft-start by
making the 0.8V reference voltage VREF ramp from 0 to
100% in about 6ms with a 9.7mV step. Therefore, the
output voltage is controlled to increase slowly by a stair-
case VREF ramp. Once the soft-start ends, the related
circuitry is disabled to reduce the current consumption.
VIN should be powered up no earlier than VHSD to make
the soft-start function behavior correctly.
Current Limit
The MIC2164/-2/-3 uses the RDS(ON) of the low-side
power MOSFET to sense over-current conditions. The
lower-side MOSFET is used because it displays much
lower parasitic oscillations during switching then the
high-side MOSFET. Using the low-side MOSFET RDS(ON)
as a current sense is an excellent method for circuit
protection. This method will avoid adding cost, board
space and power losses taken by discrete current sense
resistors.
In each switching cycle of the MIC2164/-2/-3 converter,
the inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. The sensed voltage is
compared with a current-limit threshold voltage VCL after
a blanking time of 150ns. If the sensed voltage is over
VCL, which is 130mV typical at 0.8V feedback voltage,
the MIC2164/-2/-3 turns off the high-side MOSFET and a
soft-start sequence is trigged. This mode of operation is
called the “hiccup mode” and its purpose is to protect the
down stream load in case of a hard short. The current
limit threshold VCL has a fold back characteristics related
to the FB voltage. Please refer to the “Typical
Characteristics” for the curve of VCL vs. FB voltage. The
circuit in Figure 4 illustrates the MIC2164/-2/-3 current
limiting circuit.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 15 M9999-091310-D
MOSFET Gate Drive
The MIC2164/-2/-3 high-side drive circuit is designed to
switch an N-Channel MOSFET. The block diagram of
Figure 1 shows a bootstrap circuit, consisting of D1 (a
Schottky diode is recommended) and CBST. This circuit
supplies energy to the high-side drive circuit. Capacitor
CBST is charged while the low-side MOSFET is on and
the voltage on the LX pin is approximately 0V. When the
high-side MOSFET driver is turned on, energy from CBST
is used to turn the MOSFET on. As the high-side
MOSFET turns on, the voltage on the LX pin increases
to approximately VHSD. Diode D1 is reversed biased and
CBST floats high while continuing to keep the high-side
MOSFET on. The bias current of the high-side driver is
less than 10mA so a 0.1F to 1F is sufficient to hold
the gate voltage with minimal droop for the power stroke
(high-side switching) cycle, i.e. BST = 10mA x
3.33s/0.1F = 333mV for MIC2164. When the low-side
MOSFET is turned back on, CBST is recharged through
D1. A small resistor RG, which is in series with CBST, can
slow down the turn-on time of the high-side N-channel
MOSFET.
Figure 4. MIC2164/-2/-3 Current Limiting Circuit
Using the typical VCL value of 130mV, the current limit
value is roughly estimated as:
DS(ON)
CL R
130mV
I≈
For designs where the current ripple is significant
compared to the load current IOUT, or for low duty cycle
operation, calculating the current limit ICL should take
into account that one is sensing the peak inductor
current and that there is a blanking delay of
approximately 150ns.
The drive voltage is derived from the supply voltage VIN.
The nominal low-side gate drive voltage is VIN and the
nominal high-side gate drive voltage is approximately VIN
– VDIODE, where VDIODE is the voltage drop across D1. An
approximate 30ns delay between the high-side and low-
side driver transitions is used to prevent current from
simultaneously flowing unimpeded through both
MOSFETs.
2
I
L
T*V
R
130mV
IL(pp)
DLYOUT
DS(ON)
CL −+= (2)
Lf
D)(1V
I
SW
OUT
L(pp) ⋅
−⋅
= (3)
where
VOUT = The output voltage
TDLY = Current limit blanking time, 150ns typical
IL(pp) = Inductor current ripple peak-to-peak value
D = Duty Cycle
fSW = Switching frequency
The MOSFET RDS(ON) varies 30 to 40% with
temperature; therefore, it is recommended to add a 50%
margin to ICL in the above equation to avoid false current
limiting due to increased MOSFET junction temperature
rise. It is also recommended to connect LX pin directly to
the drain of the low-side MOSFET to accurately sense
the MOSFETs RDS(ON).
Micrel, Inc. MIC2164/-2/-3/C
September 2010 16 M9999-091310-D
Application Information
MOSFET Selection
The MIC2164/-2/-3 controller works from power stage
input voltages of 3V to 28V and has an external 3V to
5.5V VIN to provide power to turn the external N-Channel
power MOSFETs for the high- and low-side switches.
For applications where VIN < 5V, it is necessary that the
power MOSFETs used are sub-logic level and are in full
conduction mode for VGS of 2.5V. For applications when
VIN > 5V; logic-level MOSFETs, whose operation is
specified at VGS = 4.5V must be used.
There are different criteria for choosing the high-side and
low-side MOSFETs. These differences are more
significant at lower duty cycles such as 12V to 1.8V
conversion. In such an application, the high-side
MOSFET is required to switch as quickly as possible to
minimize transition losses, whereas the low-side
MOSFET can switch slower, but must handle larger
RMS currents. When the duty cycle approaches 50%,
the current carrying capability of the high-side MOSFET
starts to become critical.
It is important to note that the on-resistance of a
MOSFET increases with increasing temperature. A 75°C
rise in junction temperature will increase the channel
resistance of the MOSFET by 50% to 75% of the
resistance specified at 25°C. This change in resistance
must be accounted for when calculating MOSFET power
dissipation and in calculating the value of current limit.
Total gate charge is the charge required to turn the
MOSFET on and off under specified operating conditions
(VDS and VGS). The gate charge is supplied by the
MIC2164/-2/-3 gate-drive circuit. At 300kHz switching
frequency and above, the gate charge can be a
significant source of power dissipation in the MIC2164/-
2/-3. At low output load, this power dissipation is
noticeable as a reduction in efficiency. The average
current required to drive the high-side MOSFET is:
SWGside]-G[high f Q(avg)I ×= (4)
where:
IG[high-side](avg) = Average high-side MOSFET gate
current
QG = Total gate charge for the high-side MOSFET taken
from the manufacturer’s data sheet for VGS = VIN.
fSW = Switching Frequency
The low-side MOSFET is turned on and off at VDS = 0
because an internal body diode or external freewheeling
diode is conducting during this time. The switching loss
for the low-side MOSFET is usually negligible. Also, the
gate-drive current for the low-side MOSFET is more
accurately calculated using CISS at VDS = 0 instead of
gate charge.
For the low-side MOSFET:
SWGSISSside]-G[low f V C (avg)I
×
×= (5)
Since the current from the gate drive comes from the
VIN, the power dissipated in the MIC2164/-2/-3 due to
gate drive is:
(avg))I (avg).(I V P side]-G[lowside]-G[highINGATEDRIVE +
=
(6)
A convenient figure of merit for switching MOSFETs is
the on resistance times the total gate charge RDS(ON) ×
QG. Lower numbers translate into higher efficiency. Low
gate-charge logic-level MOSFETs are a good choice for
use with the MIC2164/-2/-3. Also, the RDS(ON) of the low-
side MOSFET will determine the current limit value.
Please refer to “Current Limit” subsection is “Functional
Description” for more details.
Parameters that are important to MOSFET switch
selection are:
• Voltage rating
• On-resistance
• Total gate charge
The voltage ratings for the high-side and low-side
MOSFETs are essentially equal to the power stage input
voltage VHSD. A safety factor of 20% should be added to
the VDS(max) of the MOSFETs to account for voltage
spikes due to circuit parasitic elements.
The power dissipated in the MOSFETs is the sum of the
conduction losses during the on-time (PCONDUCTION) and
the switching losses during the period of time when the
MOSFETs turn on and off (PAC).
ACCONDUCTIONSW P P P +
=
(7)
DS(ON)
2
SW(RMS)CONDUCTION R * I P = (8)
AC(on)) AC(off
AC P P P += (9)
where:
RDS(ON) = on-resistance of the MOSFET switch
D = Duty Cycle = VOUT / VHSD
Micrel, Inc. MIC2164/-2/-3/C
September 2010 17 M9999-091310-D
Making the assumption that the turn-on and turn-off
transition times are equal; the transition times can be
approximated by:
G
HSDOSSINISS
TI
VCVC
t×+×
= (10)
where:
CISS and COSS are measured at VDS = 0
IG = gate-drive current
The total high-side MOSFET switching loss is:
SWTPKDHSD AC f t I) V(V P ×××+= (11)
where:
tT = Switching transition time
VD = Body diode drop (0.5v)
fSW = Switching Frequency
The high-side MOSFET switching losses increase with
the switching frequency and the input voltage VHSD. The
low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by the equation below:
(
)
(max)OUTSW(max)HSD
OUT(max)HSDOUT
I%20fV
VVV
L×××
−×
= (12)
where:
fSW = switching frequency
20% = ratio of AC ripple current to DC output current
VHSD(max) = maximum power stage input voltage
The peak-to-peak inductor current ripple is:
LfV
)VV(V
I
SW(max)HSD
OUT(max)HSDOUT
)PP(L ××
−×
=Δ (13)
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
L(PP)OUT(max)L(PK) I0.5 I I Δ×+= (14)
The RMS inductor current is used to calculate the I2R
losses in the inductor.
12
I
II
2
L(PP)
2
OUT(max)L(RMS) += (15)
Maximizing efficiency requires both the proper selection
of core material and the minimizing of the winding
resistance. The high frequency operation of the
MIC2164/-2/-3 requires the use of ferrite materials for all
but the most cost sensitive applications.
Lower cost iron powder cores may be used but the
increase in core loss will reduce the efficiency of the
power supply. This is especially noticeable at low output
power. The winding resistance decreases efficiency at
the higher output current levels. The winding resistance
must be minimized although this usually comes at the
expense of a larger inductor. The power dissipated in the
inductor is equal to the sum of the core and copper
losses. At higher output loads, the core losses are
usually insignificant and can be ignored. At lower output
currents, the core losses can be a significant contributor.
Core loss information is usually available from the
magnetics vendor. Copper loss in the inductor is
calculated by the equation below:
P
INDUCTORCu=IL(RMS)
2 × RWINDING (16)
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 18 M9999-091310-D
RWINDING = RWINDING(20°c) × (1 + 0.0042 × (TH – T20°C)) (17)
where:
TH = temperature of wire under full load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
Output Capacitor Selection
The type of the output capacitor is usually determined by
its ESR (equivalent series resistance). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitors
are tantalum, low-ESR aluminum electrolytic, OS-CON
and POSCAPS. The output capacitor’s ESR is usually
the main cause of the output ripple. The output capacitor
ESR also affects the control loop from a stability point of
view. The maximum value of ESR is calculated:
L(PP)
OUT(pp)
CI
V
ESR OUT ≤ (18)
where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
IL(PP) = peak-to-peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated below:
(
)
2
CL(PP)
2
SWOUT
L(PP)
OUT(pp) OUT
ESRI
8fC
I
V⋅+
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
⋅⋅
=
(19)
where:
D = duty cycle
COUT = output capacitance value
fSW = switching frequency
As described in the “Theory of Operation” subsection in
“Functional Description”, MIC2164/-2/-3 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator to behavior properly.
Also, the output voltage ripple should be in phase with
the inductor current. Therefore, the output voltage ripple
caused by the output capacitor COUT should be much
smaller than the ripple caused by the output capacitor
ESR. If low ESR capacitors are selected as the output
capacitors, such as ceramic capacitors, a ripple injection
method is applied to provide the enough FB voltage
ripples. Please refer to the “Ripple Injection” subsection
for more details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated below:
12
I
IL(PP)
(RMS)COUT = (20)
The power dissipated in the output capacitor is:
OUTOUTOUT C
2
(RMS)C)DISS(C ESRIP ⋅= (21)
Input Capacitor Selection
The input capacitor for the power stage input VHSD
should be selected for ripple current rating and voltage
rating. Tantalum input capacitors may fail when
subjected to high inrush currents, caused by turning the
input supply on. A tantalum input capacitor’s voltage
rating should be at least two times the maximum input
voltage to maximize reliability. Aluminum electrolytic,
OS-CON, and multilayer polymer film capacitors can
handle the higher inrush currents without voltage de-
rating. The input voltage ripple will primarily depend
upon the input capacitor’s ESR. The peak input current
is equal to the peak inductor current, so:
CIN)PK(LIN ESRIV ×=Δ (22)
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
()
D1DII (max)OUT)RMS(CIN −××≈ (23)
The power dissipated in the input capacitor is:
PDISS(CIN) = ICIN(RMS)
2×ESRCIN (24)
Micrel, Inc. MIC2164/-2/-3/C
September 2010 19 M9999-091310-D
External Schottky Diode (Optional)
An external freewheeling diode, which is generally not
necessary, can be used to keep the inductor current flow
continuous while both MOSFETs are turned off. This
dead time prevents current from flowing unimpeded
through both MOSFETs and is typically 30ns. The diode
conducts twice during each switching cycle. Although the
average current through this diode is small, the diode
must be able to handle the peak current.
SWOUTD(avg) f30ns2II ⋅⋅⋅= (25)
The reverse voltage requirement of the diode is:
HSDDIODE(rrm) V V =
The power dissipated by the Schottky diode is:
F D(avg)DIODE VI P ×= (26)
where, VF = forward voltage at the peak diode current.
The external Schottky diode is not necessary for the
circuit operation since the low-side MOSFET contains a
parasitic body diode. The external diode will improve
efficiency and decrease the high frequency noise. If the
MOSFET body diode is then used, it must be rated to
handle the peak and average current. The body diode
has a relatively slow reverse recovery time and a
relatively high forward voltage drop. The power lost in
the diode is proportional to the forward voltage drop of
the diode. As the high-side MOSFET starts to turn on,
the body diode becomes a short circuit for the reverse
recovery period, dissipating additional power. The diode
recovery and the circuit inductance will cause ringing
during the high-side MOSFET turn-on.
An external Schottky diode conducts at a lower forward
voltage preventing the body diode in the MOSFET from
turning on. The lower forward voltage drop dissipates
less power than the body diode. The lack of a reverse
recovery mechanism in a Schottky diode causes less
ringing and less power loss. Depending upon the circuit
components and operating conditions, an external
Schottky diode will give a ½ to 1% improvement in
efficiency.
Ripple Injection
The minimum FB voltage ripple requested by the
MIC2164/-2/-3 gm amplifier and error comparator is
20mV (100mV maximum). However, the output voltage
ripple is generally designed as 1% to 2% of the output
voltage. For a low output voltage, such as 1V output, the
output voltage ripple is only 10mV to 20mV, and the FB
voltage ripple is less than 20mV. If the FB voltage ripple
is so small that the gm amplifier and error comparator
could not sense it, then the MIC2164/-2/-3 will lose
control and the output voltage will not be regulated. In
order to have some amount of FB voltage ripple, the
ripple injection method is applied for low output voltage
ripple applications.
The applications are divided into three situations
according to the amount of the FB voltage ripple:
1) Enough ripple at the FB voltage due to the large ESR
of the output capacitors.
As shown in Figure 5a, the converter is stable without
any adding in this situation. The FB voltage ripple is:
(pp)
LCFB(pp) IESR
R2R1
R2
VOUT ⋅⋅
+
= (27)
where ΔIL(pp) is the peak-to-peak value of the inductor
current ripple.
2) Inadequate ripple at the FB voltage due to the small
ESR of the output capacitors.
The output voltage ripple is fed into the FB pin through a
feedforward capacitor Cff in this situation, as shown in
Figure 5b. The typical Cff value is between 1nF to
100nF. With the feedforward capacitor, the FB voltage
ripple is very close to the output voltage ripple:
(pp)
LFB(pp) IESRV⋅≈ (28)
3) Invisible ripple at the FB voltage is due to the very low
ESR of the output capacitors.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 20 M9999-091310-D
Figure 5a. Enough Ripple at FB
Figure 5b. Inadequate Ripple at FB
Figure 5c. Invisible Ripple at FB
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node LX via a resistor Rinj and a
capacitor Cinj, as shown in Figure 5c. The injected ripple
is:
τ×
××××=
SW
divHSDFB(pp) f
1
D)-(1DKVV (29)
R1//R2Rinj
R1//R2
Kdiv +
= (30)
where
VHSD = Power stage input voltage at HSD pin
D = Duty Cycle
fSW = switching frequency
Cff)Rinj//2R//1R( ⋅=τ
In the formula (29) and (30), it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1<<
τ
T
=
τ×fsw
1
If the voltage divider resistors R1 and R2 are in the k
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant consumption. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in k range.
Step 2. Select Rinj according to the expected feedback
voltage ripple. According to the equation (30):
)D1(D
f
V
V
KSW
HSD
)pp(FB
div −⋅
τ⋅
⋅
Δ
= (31)
Then the value of Rinj is obtained as:
)1
K
1
()2R//1R(R
div
inj −⋅= (32)
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 21 M9999-091310-D
Setting Output Voltage In addition to the external ripple injection added at the
FB pin, internal ripple injection is added at the inverting
input of the comparator inside the MIC2164/-2/-3, as
shown in Figure 7. The inverting input voltage VINJ is
clamped to 1.2V. As VOUT is increased, the swing of VINJ
will be clamped. The clamped VINJ reduces the line
regulation because it is reflected back as a DC error on
the FB terminal. Therefore, the maximum output voltage
of the MIC2164/-2/-3 should be limited to 5.5V to avoid
this problem. If a higher output voltage is required, use
the MIC2176 instead.
The MIC2164/-2/-3 requires two resistors to set the
output voltage, as shown in Figure 6:
Figure 6. Voltage-Divider Configuration
The output voltage is determined by the equation:
)
R2
R1
(1VV REFOUT +×= (33)
Figure 7. Internal Ripple Injection
where VREF = 0.8V. A typical value of R1 can be between
3k and 10k. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small, it will decrease the efficiency of the power supply,
especially at light loads. Once R1 is selected, R2 can be
calculated using:
REFOUT
REF
VV
R1V
R2 −
×
= (34)
Micrel, Inc. MIC2164/-2/-3/C
September 2010 22 M9999-091310-D
PCB Layout Guideline
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC2164/-2/-3 converter.
IC
• Place the IC and MOSFETs close to the point of
load (POL).
• Use fat traces to route the input and output power
lines.
• Signal and power grounds should be kept separate
and connected at only one location.
Input Capacitor
• Place the HSD input capacitor next.
• Place the HSD input capacitors on the same side of
the board and as close to the MOSFETs as
possible.
• Keep both the HSD and PGND connections short.
• Place several vias to the ground plane close to the
HSD input capacitor ground terminal.
• Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
• Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
• If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
• In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over-
voltage spike seen on the input supply with power is
suddenly applied.
• An additional Tantalum or Electrolytic bypass input
capacitor of 22uF or higher is required at the input
power connection.
• The 1µF and 0.1µF capacitors, which connect to the
VIN terminal, must be located right at the IC. The VIN
terminal is very noise sensitive and placement of the
capacitor is very critical. Connections must be made
with wide trace.
Inductor
• Keep the inductor connection to the switch node
(LX) short.
• Do not route any digital lines underneath or close to
the inductor.
• Keep the switch node (LX) away from the feedback
(FB) pin.
• The LX pin should be connected directly to the drain
of the low-side MOSFET to accurate sense the
voltage across the low-side MOSFET.
• To minimize noise, place a ground plane underneath
the inductor.
Output Capacitor
• Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
• Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
• The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high current
load trace can degrade the DC load regulation.
Schottky Diode (Optional)
• Place the Schottky diode on the same side of the
board as the MOSFETs and HSD input capacitor.
• The connection from the Schottky diode’s Anode to
the input capacitors ground terminal must be as
short as possible.
• The diode’s cathode connection to the switch node
(LX) must be keep as short as possible.
RC Snubber
• Place the RC snubber on the same side of the board
and as close to the MOSFETs as possible.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 23 M9999-091310-D
Evaluation Board Schematics
Figure 8. Schematic of MIC2164 20A Evaluation Board
Micrel, Inc. MIC2164/-2/-3/C
September 2010 24 M9999-091310-D
Bill of Materials
Item Part Number Manufacturer Description Qty.
B41125A7227M EPCOS(1)
C1
222215095001E3 Vishay(2)
220µF Aluminum Capacitor, SMD, 35V 1
1210YD226KAT2A AVX(3)
GRM32ER61C226KE20L Murata(4)
C2,C3
C3225X5R1C226K TDK(5)
22µF Ceramic Capacitor, X5R, Size 1210, 16V 2
06035C104KAT2A AVX(3)
GRM188R71H104KA93D Murata(4)
C6,C9,C10,
C14,C16
C1608X7R1H104K TDK(5)
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 5
0805ZD105KAT2A AVX(3)
GRM219R61A105KC01D Murata(4)
C7
C2012X5R1A105K TDK(5)
1µF Ceramic Capacitor, X5R, Size 0805, 10V 1
0805ZC225MAT2A AVX(3)
GRM21BR71A225KA01L Murata(4)
C8
C2012X7R1A225K TDK(5)
2.2µF Ceramic Capacitor, X7R, Size 0805, 10V 1
06035C102KAT2A AVX(3)
GRM188R71H102KA01D Murata(4)
C11
C1608X7R1H102K TDK(5)
1nF Ceramic Capacitor, X7R, Size 0603, 50V 1
06035C223KAZ2A AVX(3)
GRM188R71H223K Murata(4)
C12
C1608X7R1H223K TDK(5)
22nF Ceramic Capacitor, X7R, Size 0603, 50V 1
12106D107MAT2A AVX(3)
C13,C15
GRM32ER60J107ME20L Murata(4)
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 2
C17 16ME1000WG SANYO(6) 1000µF Aluminum Capacitor, 16V 1
SD103BWS-7 Diodes Inc(7)
D1
SD103BWS Vishay(2)
Small Signal Schottky Diode 1
L1 CDEP147NP-1R5M Sumida(8) 1.5µH Inductor, 27.2A Saturation Current 1
Q1,Q4 FDMS7672 Fairchild(9) 30V N-Channel MOSFET 6.9m RDS(ON) @ 4.5V 2
Q2,Q3 FDS8672S Fairchild(9) 30V N-Channel MOSFET 7m RDS(ON) @ 4.5V 2
Notes:
1. EPCOS: www.epcos.com.
2. Vishay: www.vishay.com.
3. AVX: www.avx.com.
4. Murata: www.murata.com.
5. TDK: www.tdk.com.
6. Sanyo: www.sanyo.com.
7. Diodes Inc: www.diodes.com.
8. Sumida: www.sumida.com.
9. Fairchild: www.fairchildsemi.com.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 25 M9999-091310-D
Bill of Materials (Continued)
Item Part Number Manufacturer Description Qty.
R1 CRCW06032R21FKEA Vishay/Dale(2) 2.21 Resistor, Size 0603, 1% 1
R2 CRCW06031R21FKEA Vishay/Dale(2) 1.21 Resistor, Size 0603, 1% 1
R3,R4 CRCW060310K0FKEA Vishay/Dale(2) 10k Resistor, Size 0603, 1% 2
R5 CRCW060320R0FKEA Vishay/Dale(2) 20 Resistor, Size 0603, 1% 1
R6 CRCW06033K24FKEA Vishay/Dale(2) 3.24k Resistor, Size 0603, 1% 1
U1 MIC2164YMM MICREL INC(10) 300kHz Buck Controller 1
U2(11) MIC5233-5.0YM5 MICREL INC(10) LDO 1
Notes:
10. Micrel, Inc.: www.micrel.com.
11. Optional: Required if 5V supply is not available in the system.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 26 M9999-091310-D
PCB Layout Recommendations
Figure 9. MIC2164 20A Evaluation Board Top Layer
Figure 10. MIC2164 20A Evaluation Board Bottom Layer
Micrel, Inc. MIC2164/-2/-3/C
September 2010 27 M9999-091310-D
PCB Layout Recommendations (Continued)
Figure 11. MIC2164 20A Evaluation Board Mid-Layer 1
Figure 12. MIC2164 20A Evaluation Board Mid-Layer 2
Micrel, Inc. MIC2164/-2/-3/C
September 2010 28 M9999-091310-D
Application Schematics and Bill of Materials
Figure 13. MIC2164 12V to 3.3V @ 20A Buck Converter
Micrel, Inc. MIC2164/-2/-3/C
September 2010 29 M9999-091310-D
Bill of Materials (MIC2164 12V to 3.3V @ 20A)
Item Part Number Manufacturer Description Qty.
C1, C8, C17, C19 06035C104KAT AVX(1) 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 4
C2 0805ZD225MAT AVX(1) 2.2µF Ceramic Capacitor, X5R, Size 0805, 10V 1
C3 222215095001 Vishay(2) 220µF Aluminum Capacitor, SMD, 35V 1
C4, C5, C6 1210YD226MAT AVX(1) 22µF Ceramic Capacitor, X5R, Size 1210, 16V 3
C9 0805ZD105KAT AVX(1) 1µF Ceramic Capacitor, X5R, Size 0805, 10V 1
C10 06035C223KAT AVX(1) 22nF Ceramic Capacitor, X7R, Size 0603, 50V 1
C11 16ME1000WGL Sanyo(3) 1000µF Aluminum Capacitor, 16V 1
C12 12106D107MAT AVX(1) 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1
C15 06035C102KAT AVX(1) 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1
D1 SD103BWS Vishay(2) Small Signal Schottky Diode 1
L1 CDEP147NP-1R5M Sumida(4) 1.5µH Inductor, 27.2A Saturation Current 1
Q1, Q4 FDMS7672 Fairchild(5) 30V N-Channel MOSFET 6.9m RDS(ON) @ 4.5V 2
Q2, Q3 FDS8672S Fairchild(5) 30V N-Channel MOSFET 7m RDS(ON) @ 4.5V 2
R1 CRCW06032R21FKEY3 Vishay Dale(2) 2.21 Resistor, Size 0603, 1% 1
R5 CRCW06031R21FKEY3 Vishay Dale(2) 1.21 Resistor, Size 0603, 1% 1
R6, R9 CRCW06031002FKEY3 Vishay Dale(2) 10k Resistor, Size 0603, 1% 2
R15 CRCW06033241FKEY3 Vishay Dale(2) 3.24k Resistor, Size 0603 1% 1
U1 MIC2164YMM Micrel. Inc.(6) 300kHz Buck Controller 1
U2 MIC5233-5.0YM5 Micrel. Inc.(6) LDO 1
Notes:
1. AVX: www.avx.com.
2. Vishay: www.vishay.com.
3. Sanyo: www.sanyo.com.
4. Sumida: www.sumida.com.
5. Fairchild: www.fairchildsemi.com.
6. Micrel, Inc: www.micrel.com.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 30 M9999-091310-D
Figure 14. MIC2164 12V to 1.8V @ 10A Buck Converter
Micrel, Inc. MIC2164/-2/-3/C
September 2010 31 M9999-091310-D
Bill of Materials (MIC2164 12V to 1.8V @ 10A)
Item Part Number Manufacturer Description Qty.
C1, C8, C17, C19 06035C104KAT AVX(1) 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 4
C2 0805ZD225MAT AVX(1) 2.2µF Ceramic Capacitor, X5R, Size 0805, 10V 1
C3 222215095001 Vishay(2) 220µF Aluminum Capacitor, SMD, 35V 1
C4, C5 1210YD106MAT AVX(1) 10µF Ceramic Capacitor, X5R, Size 1210, 16V 2
C9 0805ZD105KAT AVX(1) 1µF Ceramic Capacitor, X5R, Size 0805, 10V 1
C10 06035C223KAT AVX(1) 22nF Ceramic Capacitor, X7R, Size 0603, 50V 1
C11 6SEPC560MX Sanyo(3) 560µF OSCON Capacitor, 6.3V 1
C12 12106D107MAT AVX(1) 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1
C15 06035C102KAT AVX(1) 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1
D1 SD103BWS Vishay(2) Small Signal Schottky Diode 1
L1 CDEP105-2R0MC-32 Sumida(4) 2.0µH Inductor, 15.8A Saturation Current 1
Q1, Q2 FDS7764A Fairchild(5) 30V N-Channel MOSFET 7.5m RDS(ON) @ 4.5V 2
R1 CRCW06032R21FKEY3 Vishay Dale(2) 2.21 Resistor, Size 0603, 1% 1
R5 CRCW06031R21FKEY3 Vishay Dale(2) 1.21 Resistor, Size 0603, 1% 1
R6, R9 CRCW06031002FKEY3 Vishay Dale(2) 10k Resistor, Size 0603, 1% 2
R15 CRCW06038061FKEY3 Vishay Dale(2) 8.06k Resistor, Size 0603, 1% 1
U1 MIC2164YMM Micrel. Inc.(6) 300kHz Buck Controller 1
U2 MIC5233-5.0YM5 Micrel. Inc.(6) LDO 1
Notes:
1. AVX: www.avx.com.
2. Vishay: www.vishay.com.
3. Sanyo: www.sanyo.com.
4. Sumida: www.sumida.com.
5. Fairchild: www.fairchildsemi.com.
6. Micrel, Inc: www.micrel.com.
Micrel, Inc. MIC2164/-2/-3/C
Figure 15. MIC2164 12V to 1.0V @ 5A Buck Converter
September 2010 32 M9999-091310-D
Micrel, Inc. MIC2164/-2/-3/C
September 2010 33 M9999-091310-D
Bill of Materials (MIC2164 12V to 1.0V @ 5A)
Item Part Number Manufacturer Description Qty.
C1, C8, C17, C19 06035C104KAT AVX(1) 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 4
C2 0805ZD225MAT AVX 2.2µF Ceramic Capacitor, X5R, Size 0805, 10V 1
C3 222215095001 Vishay(2) 220µF Aluminum Capacitor, SMD, 35V 1
C4 1210YD106MAT AVX 10µF Ceramic Capacitor, X5R, Size 1210, 16V 1
C9 0805ZD105KAT AVX 1µF Ceramic Capacitor, X5R, Size 0805, 10V 1
C10 06035C223KAT AVX 22nF Ceramic Capacitor, X7R, Size 0603, 50V 1
C11, C12, C13 12106D107MAT AVX 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 3
C15 06035C102KAT AVX 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1
D1 SD103BWS Vishay Small Signal Schottky Diode 1
L1 CDRH104RNP-3R8 Sumida(4) 3.8µH Inductor, 6A Saturation Current 1
Q1 FDS6910 Fairchild(5) Dual 30V N-Channel MOSFET 17m RDS(ON) @ 4.5V 1
R1 CRCW06032R21FKEY3 Vishay Dale(2) 2.21 Resistor, Size 0603, 1% 1
R5 CRCW06031R21FKEY3 Vishay Dale 1.21 Resistor, Size 0603, 1% 1
R6, R9 CRCW06031002FKEY3 Vishay Dale 10k Resistor, Size 0603, 1% 2
R15 CRCW06034022FKEY3 Vishay Dale 40.2k Resistor, Size 0603, 1% 1
U1 MIC2164YMM Micrel. Inc.(6) 300kHz Buck Con t roller 1
U2 MIC5233-5.0YM5 Micrel. Inc. LDO 1
Notes:
1. AVX: www.avx.com.
2. Vishay: www.vishay.com.
3. Sanyo: www.sanyo.com.
4. Sumida: www.sumida.com.
5. Fairchild: www.fairchildsemi.com.
6. Micrel, Inc: www.micrel.com.
Micrel, Inc. MIC2164/-2/-3/C
Figure 16. MIC2164-2 12V to 3.3V @ 15A Buck Converter
September 2010 34 M9999-091310-D
Micrel, Inc. MIC2164/-2/-3/C
September 2010 35 M9999-091310-D
Bill of Materials (MIC2164-2 12V to 3.3V @ 15A)
Item Part Number Manufacturer Description Qty.
C1, C8, C17, C19 06035C104KAT AVX(1) 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 4
C2 0805ZD225MAT AVX(1) 2.2µF Ceramic Capacitor, X5R, Size 0805, 10V 1
C3 222215095001 Vishay(2) 220µF Aluminum Capacitor, SMD, 35V 1
C4, C5 1210YD226MAT AVX(1) 22µF Ceramic Capacitor, X5R, Size 1210, 16V 2
C9 0805ZD105KAT AVX(1) 1µF Ceramic Capacitor, X5R, Size 0805, 10V 1
C10 06035C472KAT AVX(1) 4.7nF Ceramic Capacitor, X7R, Size 0603, 50V 1
C11 16ME1000WGL Sanyo(3) 1000µF Aluminum Capacitor, 16V 1
C12 12106D107MAT AVX(1) 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1
C15 06035C102KAT AVX(1) 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1
D1 SD103BWS Vishay(2) Small Signal Schottky Diode 1
L1 HCP1305-1R0 Cooper Bussmann(4) 1.0µH Inductor, 29A DC Current 1
Q1 FDMS7672 Fairchild(5) 30V N-Channel MOSFET 6.9m RDS(ON) @ 4.5V 1
Q2, Q3 FDS8874 Fairchild(5) 30V N-Channel MOSFET 7.0m RDS(ON) @ 4.5V 2
R1 CRCW06032R21FKEY3 Vishay Dale(2) 2.21 Resistor, Size 0603, 1% 1
R5 CRCW06031R21FKEY3 Vishay Dale(2) 1.21 Resistor, Size 0603, 1% 1
R6 CRCW06031002FKEY3 Vishay Dale(2) 10k Resistor, Size 0603, 1% 1
R9 CRCW06034021FKEY3 Vishay Dale(2) 4.02k Resistor, Size 0603, 1% 1
R15 CRCW06033241FKEY3 Vishay Dale(2) 3.24k Resistor, Size 0603 1% 1
U1 MIC2164-2YMM Micrel. Inc.(6) 600 kHz Buck Controller 1
U2 MIC5233-5.0YM5 Micrel. Inc.(6) LDO 1
Notes:
1. AVX: www.avx.com.
2. Vishay: www.vishay.com.
3. Sanyo: www.sanyo.com.
4. Sumida: www.sumida.com.
5. Fairchild: www.fairchildsemi.com.
6. Micrel, Inc: www.micrel.com.
Micrel, Inc. MIC2164/-2/-3/C
Figure 17. MIC2164-3 12V to 1.8V @ 10A Buck Converter
September 2010 36 M9999-091310-D
Micrel, Inc. MIC2164/-2/-3/C
September 2010 37 M9999-091310-D
Bill of Materials (MIC2164-3 12V to 1.8V @ 10A)
Item Part Number Manufacturer Description Qty.
C1, C8, C17, C19 06035C104KAT AVX(1) 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 4
C2 0805ZD225MAT AVX(1) 2.2µF Ceramic Capacitor, X5R, Size 0805, 10V 1
C3 222215095001 Vishay(2) 220µF Aluminum Capacitor, SMD, 35V 1
C4 1210YD106MAT AVX(1) 10µF Ceramic Capacitor, X5R, Size 1210, 16V 1
C9 0805ZD105KAT AVX(1) 1µF Ceramic Capacitor, X5R, Size 0805, 10V 1
C10 06035C222KAT AVX(1) 2.2nF Ceramic Capacitor, X7R, Size 0603, 50V 1
C11 6SEPC560MX Sanyo(3) 560µF OSCON Capacitor, 6.3V 1
C12 12106D107MAT AVX(1) 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1
C15 06035C102KAT AVX(1) 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1
D1 SD103BWS Vishay(2) Small Signal Schottky Diode 1
L1 HCF1305-1R0 Cooper Bussmann(4) 1.0µH Inductor, 20A Saturation Current 1
Q1, Q2 FDS7764A Fairchild(5) 30V N-Channel MOSFET 7.5m Rds(on) @ 4.5V 2
R1 CRCW06032R21FKEY3 Vishay Dale(2) 2.21 Resistor, Size 0603, 1% 1
R5 CRCW06031R21FKEY3 Vishay Dale(2) 1.21 Resistor, Size 0603, 1% 1
R6 CRCW06031002FKEY3 Vishay Dale(2) 10k Resistor, Size 0603, 1% 1
R9 CRCW06032001FKEY3 Vishay Dale(2) 2k Resistor, Size 0603, 1% 1
R15 CRCW06038061FKEY3 Vishay Dale(2) 8.06k Resistor, Size 0603, 1% 1
U1 MIC2164-3YMM Micrel. Inc.(6) 1MHz Bu ck Controller 1
U2 MIC5233-5.0YM5 Micrel. Inc.(6) LDO 1
Notes:
1. AVX: www.avx.com.
2. Vishay: www.vishay.com.
3. Sanyo: www.sanyo.com.
4. Sumida: www.sumida.com.
5. Fairchild: www.fairchildsemi.com.
6. Micrel, Inc: www.micrel.com.
Micrel, Inc. MIC2164/-2/-3/C
September 2010 38 M9999-091310-D
Recommended Land Pattern
10-Pin MSOP (MM)
Micrel, Inc. MIC2164/-2/-3/C
September 2010 39 M9999-091310-D
Package Information
10-Pin MSOP (MM)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical impla
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
can nt
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