LM3224 RFB1
L
CC
D
RC
CIN
COUT
RFB2
SW
5
FB 2
GND
4
VC
VOUT
VIN
Battery or
Power Source
SS
8
CSS
Optional
FSLCT
VIN
6 7
SHDN
3
1
LM3224
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SNVS277C –DECEMBER 2004–REVISED MARCH 2013
LM3224 615kHz/1.25MHz Step-up PWM DC/DC Converter
Check for Samples: LM3224
1FEATURES DESCRIPTION
The LM3224 is a step-up DC/DC converter with a
2• Operating Voltage Range of 2.7V to 7V 0.15Ω(typ.), 2.45A (typ.) internal switch and pin
• 615kHz/1.25MHz Pin Selectable Frequency selectable operating frequency. With the ability to
Operation convert 3.3V to multiple outputs of 8V, -8V, and 23V,
• Over Temperature Protection the LM3224 is an ideal part for biasing TFT displays.
With the high current switch it is also ideal for driving
• Optional Soft-Start Function high current white LEDs for flash applications. The
• 8-Lead VSSOP Package LM3224 can be operated at switching frequencies of
615kHz and 1.25MHz allowing for easy filtering and
APPLICATIONS low noise. An external compensation pin gives the
user flexibility in setting frequency compensation,
• TFT Bias Supplies which makes possible the use of small, low ESR
• Handheld Devices ceramic capacitors at the output. An external soft-
• Portable Applications start pin allows the user to control the amount of
inrush current during start up. The LM3224 is
• GSM/CDMA Phones available in a low profile 8-lead VSSOP package.
• Digital Cameras
• White LED Flash/Torch Applications
Typical Application Circuit
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 2004–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
VC
1
FB
2
SHDN
3
GND
4
SS
8
FSLCT
7
SW 5
VIN 6
LM3224
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
www.ti.com
Connection Diagram
Figure 1. 8-Lead Plastic VSSOP
Top View
Package Number DGK0008A
PIN DESCRIPTIONS
Pin Name Function
1 VCCompensation network connection. Connected to the output of the voltage error amplifier.
2 FB Output voltage feedback input.
3 SHDN Shutdown control input, active low. This pin has an internal pulldown resistor so the default condition
is off. The pin must be pulled high to turn on the device.
4 GND Analog and power ground.
5 SW Power switch input. Switch connected between SW pin and GND pin.
6 VIN Analog power input.
7 FSLCT Switching frequency select input. VIN = 1.25MHz. Ground = 615kHz.
8 SS Soft-start Pin.
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PWM
COMP
OVP
COMP
ERROR
AMP
UVP
COMP
LOGIC
Drive
UVP
Reset
OVP
Set Reset
Thermal SD
Oscillator Duty
Cycle Limit Load Current
Measurement
Driver
Shutdown
Comparator
Thermal
Shutdown
Bandgap Voltage
Reference
+
-
-
+
BG
-
+
BG
+BG
-
SHDN VIN GND
VC
FSLCT
¦
SW
FB
SS
LM3224
www.ti.com
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
Block Diagram
General Description
The LM3224 utilizes a PWM control scheme to regulate the output voltage over all load conditions. The operation
can best be understood referring to the block diagram and Figure 21 of the Operation section. At the start of
each cycle, the oscillator sets the driver logic and turns on the NMOS power device conducting current through
the inductor, cycle 1 of Figure 21 (a). During this cycle, the voltage at the VCpin controls the peak inductor
current. The VCvoltage will increase with larger loads and decrease with smaller. This voltage is compared with
the summation of the SW voltage and the ramp compensation. The ramp compensation is used in PWM
architectures to eliminate the sub-harmonic oscillations that occur during duty cycles greater than 50%. Once the
summation of the ramp compensation and switch voltage equals the VCvoltage, the PWM comparator resets the
driver logic turning off the NMOS power device. The inductor current then flows through the schottky diode to the
load and output capacitor, cycle 2 of Figure 21 (b). The NMOS power device is then set by the oscillator at the
end of the period and current flows through the NMOS power device once again.
The LM3224 has dedicated protection circuitry running during normal operation to protect the IC. The Thermal
Shutdown circuitry turns off the NMOS power device when the die temperature reaches excessive levels. The
UVP comparator protects the NMOS power device during supply power startup and shutdown to prevent
operation at voltages less than the minimum input voltage. The OVP comparator is used to prevent the output
voltage from rising at no loads allowing full PWM operation over all load conditions. The LM3224 also features a
shutdown mode decreasing the supply current to 0.1µA (typ.).
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
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LM3224
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
www.ti.com
Absolute Maximum Ratings(1) (2)(2)
VIN 7.5V
SW Voltage 21V
FB Voltage (3) 7V
VCVoltage (4) 1.26V ± 0.3V
SHDN Voltage 7.5V
FSLCT 7.5V
Maximum Junction Temperature 150°C
Power Dissipation(5) Internally Limited
Lead Temperature 300°C
Vapor Phase (60 sec.) 215°C
Infrared (15 sec.) 220°C
ESD Susceptibility (6) Human Body Model 2kV
Machine Model 200V
(1) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications
(2) Absolute maximum ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions for which the
device is intended to be functional, but device parameter specifications may not be ensured. For ensured specifications and test
conditions, see the Electrical Characteristics().
(3) The FB pin should never exceed VIN.
(4) Under normal operation the VCpin may go to voltages above this value. This maximum rating is for the possibility of a voltage being
applied to the pin, however the VCpin should never have a voltage directly applied to it.
(5) The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(MAX), the junction-to-ambient thermal
resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation at any ambient temperature is calculated
using: PD(MAX) = (TJ(MAX) −TA)/θJA. Exceeding the maximum allowable power dissipation will cause excessive die temperature, and
the regulator will go into thermal shutdown.
(6) The human body model is a 100 pF capacitor discharged through a 1.5kΩresistor into each pin. The machine model is a 200pF
capacitor discharged directly into each pin.
Operating Conditions
Operating Junction Temperature Range −40°C to +125°C
Storage Temperature −65°C to +150°C
Supply Voltage 2.7V to 7V
Maximum Output Voltage 20V
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LM3224
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SNVS277C –DECEMBER 2004–REVISED MARCH 2013
Electrical Characteristics(1)
Specifications in standard type face are for TJ= 25°C and those with boldface type apply over the full Operating
Temperature Range ( TJ=−40°C to +125°C). VIN = 2.7V, FSLCT = SHDN = VIN, and IL= 0A, unless otherwise specified.
Min Typ Max
Symbol Parameter Conditions Units
(1) (2) (1)
IQQuiescent Current FB = 2V (Not Switching) 1.3 2.0 mA
VSHDN = 0V 0.1 2.0 µA
VFB Feedback Voltage 1.2285 1.26 1.2915 V
ICL(3) Switch Current Limit VIN = 2.7V (4) 1.9 2.45 2.8
VIN = 3V, VOUT = 8V 2.1 A
VIN = 3V, VOUT = 5V 2.2
%VFB/ΔVIN Feedback Voltage Line 2.7V ≤VIN ≤7V 0.085 0.15 %/V
Regulation
IBFB Pin Bias Current (5)(6) 35 250 nA
ISS SS Pin Current 7.5 11 13 µA
VSS SS Pin Voltage 1.2090 1.2430 1.2622
VIN Input Voltage Range 2.7 7 V
gmError Amp Transconductance ΔI = 5µA 40 87 135 µmho
AVError Amp Voltage Gain 78 V/V
DMAX Maximum Duty Cycle 85 92.5 %
fSSwitching Frequency FSLCT = Ground 450 615 750 kHz
FSLCT = VIN 0.9 1.25 1.5 MHz
ISHDN Shutdown Pin Current VSHDN = 2.7V 2.4 5.0 µA
VSHDN = 0.3V 0.3 1.2
ILSwitch Leakage Current VSW = 20V 0.2 8.0 µA
RDSON Switch RDSON VIN = 2.7V, ISW = 1A 0.15 0.4 Ω
ThSHDN Shutdown Threshold Output High 1.2 0.8 V
Output Low 0.8 0.3 V
UVP On Threshold 2.3 2.5 V
Off Threshold 2.6 2.7 V
(1) All limits ensured at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are
100% production tested. All limits at temperature extremes are ensured via correlation using standard Statistical Quality Control (SQC)
methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Typical numbers are at 25°C and represent the most likely norm.
(3) Duty cycle affects current limit due to ramp generator.
(4) Current limit at 0% duty cycle. See Typical Performance Characteristics for Switch Current Limit vs. VIN
(5) Bias current flows into FB pin.
(6) The FB pin should never exceed VIN.
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2.6 3.4 3.8 5.4 7.0
0.05
0.07
0.11
0.13
0.15
0.17
0.19
0.21
0.23
NMOS RDSON (:)
INPUT VOLTAGE (V)
0.09
3.0 4.2 4.6 5.0 5.8 6.2 6.6
TA = 85oC
ISW = 1.5A
TA = -40oC
TA = 25oC
2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0
INPUT VOLTAGE (V)
1.0
1.1
1.2
1.3
1.4
1.5
1.6
1.7
1.8
NON-SWITCHING IQ(mA)
2.0
TJ= -40oC
TJ= 25oC
TJ= 125oC
0
1
2
3
4
5
6
7
8
2.6 3.0 3.4 3.8 4.2 4.6 5.0 5.4 5.8 6.2 6.6 7.0
FSLCT VOLTAGE (V)
FSLCT PIN CURRENT (PA)
TJ = 125oC
FSLCT = VIN
TJ = -40oC
TJ = 25oC
-40 -25 -10 5 20 35 50 65 80 95 110
0
10
20
30
40
50
60
70
FB PIN CURRENT (nA)
TEMPERATURE (oC)
125
VIN = 2.7V
VIN = 7.0V
FB = 1.265V
10.2
10.4
10.6
10.8
11.0
11.2
11.4
11.6
11.8
-40 -25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (oC)
SS PIN CURRENT (PA)
VIN = 7.0V
VIN = 2.7V
LM3224
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
www.ti.com
Typical Performance Characteristics
SHDN Pin Current vs. SHDN Pin Voltage SS Pin Current vs. Temperature
Figure 2. Figure 3.
FSLCT Pin Current vs. FSLCT Pin Voltage FB Pin Current vs. Temperature
Figure 4. Figure 5.
NMOS RDSON vs. Input Voltage 615kHz Non-switching IQvs. Input Voltage
Figure 6. Figure 7.
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-40 -10 5 65 125
3.0
3.5
4.5
5.0
5.5
6.0
6.5
7.0
7.5
SWITCHING IQ (mA)
TEMPERATURE (oC)
4.0
-25 20 35 50 80 95 110
VIN = 2.7V
VIN = 7.0V
2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0
INPUT VOLTAGE (V)
0.0
1.0
2.0
3.0
4.0
5.0
6.0
7.0
8.0
SWITCHING IQ(mA)
2.0
TJ= -40oC
TJ= 25oC
TJ= 125oC
-25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (oC)
2.00
4.50
SWITCHING IQ(mA)
-40
4.00
3.50
3.00
2.50
2.25
4.25
3.75
3.25
2.75
VIN = 2.7V
VIN = 7.0V
2.0 3.0 4.0 5.0 6.0 7.0
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
SWITCHING IQ (mA)
INPUT VOLTAGE (V)
2.5 3.5 4.5 5.5 6.5
TJ = 125oC
TJ = -40oC
TJ = 25oC
2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0
1.0
1.1
1.2
1.3
1.4
1.5
1.6
1.7
1.8
1.9
2.0
NON-SWITCHING IQ(mA)
INPUT VOLTAGE (V)
TJ= 125oC
TJ= -40oC
TJ= 25oC
LM3224
www.ti.com
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
Typical Performance Characteristics (continued)
1.25MHz Non-switching IQvs. Input Voltage 615kHz Switching IQvs. Input Voltage
Figure 8. Figure 9.
1.25MHz Switching IQvs. Input Voltage 615kHz Switching IQvs. Temperature
Figure 10. Figure 11.
1.25MHz Switching IQvs. Temperature 615kHz Switching Frequency vs. Temperature
Figure 12. Figure 13.
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-30
-20
-10
0
10
20
30
40
50
60
70
80
90
TEMPERATURE (oC)
1.75
1.80
1.85
1.90
1.95
2.00
2.05
2.10
2.15
2.20
CURRENT LIMIT (A)
-40
VOUT = 8V
VIN = 4.2V
VIN = 5.5V
VIN = 3.0V
-40 -20 0 20 40 60 80
TEMPERATURE (oC)
1.40
1.50
1.60
1.70
1.80
1.90
2.00
2.10
CURRENT LIMIT (A)
-30 -10 10 30 50 70 90
VIN = 4.2V
VIN = 5.5V
VIN = 3.0V
VOUT = 15V
90.0
90.5
91.0
91.5
92.0
92.5
93.0
93.5
94.0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (oC)
MAXIMUM DUTY CYCLE (%)
VIN = 7.0V
VIN = 2.7V
2.6 5.8
INPUT VOLTAGE (V)
1.5
1.6
1.7
1.8
1.9
2.0
2.1
2.2
2.3
CURRENT LIMIT (A)
3.0 3.4 3.8 4.2 4.6 5.0 5.4
VOUT = 15V
VOUT = 12V
VOUT = 8V
-40 -10 5 65 125
1.16
1.18
1.22
1.24
1.26
1.28
1.30
1.32
1.34
SWITCHING FREQUENCY (MHz)
TEMPERATURE (oC)
1.20
-25 20 35 50 80 95 110
VIN = 2.7V
VIN = 7.0V
-25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (oC)
91.5
94.0
MAXIMUM DUTY CYCLE (%)
-40
93.5
93.0
92.5
92.0
VIN = 7.0V
VIN = 2.7V
LM3224
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
www.ti.com
Typical Performance Characteristics (continued)
1.25MHz Switching Frequency vs. Temperature 615kHz Maximum Duty Cycle vs. Temperature
Figure 14. Figure 15.
1.25MHz Maximum Duty Cycle vs. Temperature Switch Current Limit vs. VIN
Figure 16. Figure 17.
Switch Current Limit vs. Temperature Switch Current Limit vs. Temperature
Figure 18. Figure 19.
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0
10
20
30
40
50
100
EFFICIENCY (%)
0. 1 1 10 100 1000
LOAD CURRENT (mA)
60
70
80
90
10000
VOUT =8V
VIN = 4. 2V
VIN =5.5V
VIN = 2.7V
VIN = 3.3V
LM3224
www.ti.com
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
Typical Performance Characteristics (continued)
1.25MHz Efficiency vs. Load Current
Figure 20.
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Product Folder Links: LM3224
:
RFB1 = RFB2 xVOUT - 1.26
1.26
VOUT = VIN
1-D , D' = (1-D) = VIN
VOUT
VIN COUT
PWM
L
RLOAD
VIN
L
COUT RLOAD
X
VIN
L
C
OUT RLOAD
Cycle 1 Cycle 2
(a) (b)
+
VOUT
-
+
VOUT
-
D
LM3224
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
www.ti.com
OPERATION
Figure 21. Simplified Boost Converter Diagram
(a) First Cycle of Operation (b) Second Cycle Of Operation
CONTINUOUS CONDUCTION MODE
The LM3224 is a current-mode, PWM boost regulator. A boost regulator steps the input voltage up to a higher
output voltage. In continuous conduction mode (when the inductor current never reaches zero at steady state),
the boost regulator operates in two cycles.
In the first cycle of operation, shown in Figure 21 (a), the transistor is closed and the diode is reverse biased.
Energy is collected in the inductor and the load current is supplied by COUT.
The second cycle is shown in Figure 21 (b). During this cycle, the transistor is open and the diode is forward
biased. The energy stored in the inductor is transferred to the load and output capacitor.
The ratio of these two cycles determines the output voltage. The output voltage is defined approximately as:
where
• D is the duty cycle of the switch (1)
D and D′will be required for design calculations.
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the feedback pin and a resistor divider connected to the output as shown in the
typical operating circuit. The feedback pin voltage is 1.26V, so the ratio of the feedback resistors sets the output
voltage according to the following equation:
(2)
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Product Folder Links: LM3224
t (s)
t (s)
L
i'
IL_AVG
ID_AVG
=IOUT_AVG
D*Ts Ts
D*Ts Ts
IL (A)
ID (A)
L
VIN
LVV OUTIN
LVV OUTIN
(a)
(b)
LM3224
www.ti.com
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
SOFT-START CAPACITOR
The LM3224 has a soft-start pin that can be used to limit the inductor inrush current on start-up. The external SS
pin is used to tailor the soft-start for a specific application but is not required for all applications and can be left
open when not needed. When used, a current source charges the external soft-start capacitor, Css. The soft-
start time can be estimated as:
Tss = Css*1.24V/Iss (3)
THERMAL SHUTDOWN
The LM3224 includes thermal shutdown protection. If the die temperature exceeds 140°C the regulator will shut
off the power switch, significantly reducing power dissipation in the device. The switch will remain off until the die
temperature is reduced to approximately 120°C. If the cause of the excess heating is not removed (excessive
ambient temperature, excessive power dissipation, or both) the device will continue to cycle on and off in this
manner to protect from damage.
INTRODUCTION TO COMPENSATION
Figure 22. (a) Inductor current. (b) Diode current.
The LM3224 is a current mode PWM boost converter. The signal flow of this control scheme has two feedback
loops, one that senses switch current and one that senses output voltage.
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'iL = (in Amps)
VIND
2Lfs
VINRDSON
0.144 fs
L > (in H)
)(D
D' -1
Hz
1
2S(RC + RO)CC
fPC =
Hz
1
fZC =2SRCCC
LM3224
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
www.ti.com
To keep a current programmed control converter stable above duty cycles of 50%, the inductor must meet
certain criteria. The inductor, along with input and output voltage, will determine the slope of the current through
the inductor (see Figure 22 (a)). If the slope of the inductor current is too great, the circuit will be unstable above
duty cycles of 50%. A 10µH to 15µH inductor is recommended for most 615 kHz applications, while a 4.7µH to
10µH inductor may be used for most 1.25 MHz applications. If the duty cycle is approaching the maximum of
85%, it may be necessary to increase the inductance by as much as 2X. See Inductor and Diode Selection for
more detailed inductor sizing.
The LM3224 provides a compensation pin (VC) to customize the voltage loop feedback. It is recommended that a
series combination of RCand CCbe used for the compensation network, as shown in the typical application
circuit. For any given application, there exists a unique combination of RCand CCthat will optimize the
performance of the LM3224 circuit in terms of its transient response. The series combination of RCand CC
introduces a pole-zero pair according to the following equations:
(4)
where
• ROis the output impedance of the error amplifier (approximately 900kΩ) (5)
For most applications, performance can be optimized by choosing values within the range 5kΩ≤RC≤100kΩ(RC
can be up to 200kΩif CC2 is used, see High Output Capacitor ESR Compensation) and 680pF ≤CC≤10nF.
Refer to the Applications Information section for recommended values for specific circuits and conditions. Refer
to the Compensation section for other design requirement.
COMPENSATION
This section will present a general design procedure to help insure a stable and operational circuit. The designs
in this datasheet are optimized for particular requirements. If different conversions are required, some of the
components may need to be changed to ensure stability. Below is a set of general guidelines in designing a
stable circuit for continuous conduction operation, in most all cases this will provide for stability during
discontinuous operation as well. The power components and their effects will be determined first, then the
compensation components will be chosen to produce stability.
INDUCTOR AND DIODE SELECTION
Although the inductor sizes mentioned earlier are fine for most applications, a more exact value can be
calculated. To ensure stability at duty cycles above 50%, the inductor must have some minimum value
determined by the minimum input voltage and the maximum output voltage. This equation is:
where
• fs is the switching frequency
• D is the duty cycl
• RDSON is the ON resistance of the internal switch taken from the graph "NMOS RDSON vs. Input Voltage" in the
Typical Performance Characteristics section. (6)
This equation is only good for duty cycles greater than 50% (D>0.5), for duty cycles less than 50% the
recommended values may be used. The corresponding inductor current ripple as shown in Figure 22 (a) is given
by:
(7)
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m1
#
VINRDSON
L(in V/s)
n = 1+ 2mc
m1 (no unit)
Leff = L
(D')2
Zc(in rad/s)
2fs
nD'
#
ADC(DB) = 20log10 {[(ZcLeff)// RL]//RL}(in dB)
RFB1 + RFB2
RFB2
()gmROD'
RDSON
LM3224
www.ti.com
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
The inductor ripple current is important for a few reasons. One reason is because the peak switch current will be
the average inductor current (input current or ILOAD/D') plus ΔiL. As a side note, discontinuous operation occurs
when the inductor current falls to zero during a switching cycle, or ΔiLis greater than the average inductor
current. Therefore, continuous conduction mode occurs when ΔiLis less than the average inductor current. Care
must be taken to make sure that the switch will not reach its current limit during normal operation. The inductor
must also be sized accordingly. It should have a saturation current rating higher than the peak inductor current
expected. The output voltage ripple is also affected by the total ripple current.
The output diode for a boost regulator must be chosen correctly depending on the output voltage and the output
current. The typical current waveform for the diode in continuous conduction mode is shown in Figure 22 (b). The
diode must be rated for a reverse voltage equal to or greater than the output voltage used. The average current
rating must be greater than the maximum load current expected, and the peak current rating must be greater
than the peak inductor current. During short circuit testing, or if short circuit conditions are possible in the
application, the diode current rating must exceed the switch current limit. Using Schottky diodes with lower
forward voltage drop will decrease power dissipation and increase efficiency.
DC GAIN AND OPEN-LOOP GAIN
Since the control stage of the converter forms a complete feedback loop with the power components, it forms a
closed-loop system that must be stabilized to avoid positive feedback and instability. A value for open-loop DC
gain will be required, from which you can calculate, or place, poles and zeros to determine the crossover
frequency and the phase margin. A high phase margin (greater than 45°) is desired for the best stability and
transient response. For the purpose of stabilizing the LM3224, choosing a crossover point well below where the
right half plane zero is located will ensure sufficient phase margin.
To ensure a bandwidth of ½ or less of the frequency of the RHP zero, calculate the open-loop DC gain, ADC.
After this value is known, you can calculate the crossover visually by placing a −20dB/decade slope at each pole,
and a +20dB/decade slope for each zero. The point at which the gain plot crosses unity gain, or 0dB, is the
crossover frequency. If the crossover frequency is less than ½ the RHP zero, the phase margin should be high
enough for stability. The phase margin can also be improved by adding CC2 as discussed later in this section.
The equation for ADC is given below with additional equations required for the calculation:
(8)
(9)
(10)
(11)
mc ≊0.072fs (in V/s) (12)
where
• RLis the minimum load resistance
• VIN is the minimum input voltage
• gmis the error amplifier transconductance found in the Electrical Characteristics table
• RDSON is the value chosen from the graph "NMOS RDSON vs. Input Voltage" in the Typical Performance
Characteristics section (13)
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Product Folder Links: LM3224
fPC = 1
2S(RC + RO)CC(in Hz)
(in Hz)
RHPzero = VOUT(D')2
2S,LOADL
fZ1 = 1
2SRESRCOUT (in Hz)
fP1 = 1
2S(RESR + RL)COUT (in Hz)
LM3224
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
www.ti.com
INPUT AND OUTPUT CAPACITOR SELECTION
The switching action of a boost regulator causes a triangular voltage waveform at the input. A capacitor is
required to reduce the input ripple and noise for proper operation of the regulator. The size used is dependant on
the application and board layout. If the regulator will be loaded uniformly, with very little load changes, and at
lower current outputs, the input capacitor size can often be reduced. The size can also be reduced if the input of
the regulator is very close to the source output. The size will generally need to be larger for applications where
the regulator is supplying nearly the maximum rated output or if large load steps are expected. A minimum value
of 10µF should be used for the less stressful condtions while a 22µF to 47µF capacitor may be required for
higher power and dynamic loads. Larger values and/or lower ESR may be needed if the application requires very
low ripple on the input source voltage.
The choice of output capacitors is also somewhat arbitrary and depends on the design requirements for output
voltage ripple. It is recommended that low ESR (Equivalent Series Resistance, denoted RESR) capacitors be used
such as ceramic, polymer electrolytic, or low ESR tantalum. Higher ESR capacitors may be used but will require
more compensation which will be explained later on in the section. The ESR is also important because it
determines the peak to peak output voltage ripple according to the approximate equation:
ΔVOUT ≊2ΔiLRESR (in Volts) (14)
A minimum value of 10µF is recommended and may be increased to a larger value. After choosing the output
capacitor you can determine a pole-zero pair introduced into the control loop by the following equations:
(15)
where
• RLis the minimum load resistance corresponding to the maximum load current (16)
The zero created by the ESR of the output capacitor is generally very high frequency if the ESR is small. If low
ESR capacitors are used it can be neglected. If higher ESR capacitors are used see the High Output Capacitor
ESR Compensation section. Some suitable capacitor vendors include Vishay, Taiyo-Yuden, and TDK.
RIGHT HALF PLANE ZERO
A current mode control boost regulator has an inherent right half plane zero (RHP zero). This zero has the effect
of a zero in the gain plot, causing an imposed +20dB/decade on the rolloff, but has the effect of a pole in the
phase, subtracting another 90° in the phase plot. This can cause undesirable effects if the control loop is
influenced by this zero. To ensure the RHP zero does not cause instability issues, the control loop should be
designed to have a bandwidth of less than ½ the frequency of the RHP zero. This zero occurs at a frequency of:
where
• ILOAD is the maximum load current. (17)
SELECTING THE COMPENSATION COMPONENTS
The first step in selecting the compensation components RCand CCis to set a dominant low frequency pole in
the control loop. Simply choose values for RCand CCwithin the ranges given in the Introduction to
Compensation section to set this pole in the area of 10Hz to 500Hz. The frequency of the pole created is
determined by the equation:
where
• ROis the output impedance of the error amplifier, approximately 900kΩ(18)
14 Submit Documentation Feedback Copyright © 2004–2013, Texas Instruments Incorporated
Product Folder Links: LM3224
fPC2 = 1
2SCC2(RC //RO)(in Hz)
fZC = 1
2SCCRC(in Hz)
LM3224
www.ti.com
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
Since RCis generally much less than RO, it does not have much effect on the above equation and can be
neglected until a value is chosen to set the zero fZC. fZC is created to cancel out the pole created by the output
capacitor, fP1. The output capacitor pole will shift with different load currents as shown by the equation, so setting
the zero is not exact. Determine the range of fP1 over the expected loads and then set the zero fZC to a point
approximately in the middle. The frequency of this zero is determined by:
(19)
Now RCcan be chosen with the selected value for CC. Check to make sure that the pole fPC is still in the 10Hz to
500Hz range, change each value slightly if needed to ensure both component values are in the recommended
range.
HIGH OUTPUT CAPACITOR ESR COMPENSATION
When using an output capacitor with a high ESR value, or just to improve the overall phase margin of the control
loop, another pole may be introduced to cancel the zero created by the ESR. This is accomplished by adding
another capacitor, CC2, directly from the compensation pin VCto ground, in parallel with the series combination of
RCand CC. The pole should be placed at the same frequency as fZ1, the ESR zero. The equation for this pole
follows:
(20)
To ensure this equation is valid, and that CC2 can be used without negatively impacting the effects of RCand CC,
fPC2 must be greater than 10fZC.
CHECKING THE DESIGN
With all the poles and zeros calculated the crossover frequency can be checked as described in the section DC
Gain and Open-loop Gain. The compensation values can be changed a little more to optimize performance if
desired. This is best done in the lab on a bench, checking the load step response with different values until the
ringing and overshoot on the output voltage at the edge of the load steps is minimal. This should produce a
stable, high performance circuit. For improved transient response, higher values of RCshould be chosen. This
will improve the overall bandwidth which makes the regulator respond more quickly to transients. If more detail is
required, or the most optimum performance is desired, refer to a more in depth discussion of compensating
current mode DC/DC switching regulators.
POWER DISSIPATION
The output power of the LM3224 is limited by its maximum power dissipation. The maximum power dissipation is
determined by the formula
PD= (Tjmax - TA)/θJA
where
• Tjmax is the maximum specidfied junction temperature (125°C)
• TAis the ambient temperature
•θJA is the thermal resistance of the package (21)
LAYOUT CONSIDERATIONS
The input bypass capacitor CIN, as shown in the typical operating circuit, must be placed close to the IC. This will
reduce copper trace resistance which effects input voltage ripple of the IC. For additional input voltage filtering, a
100nF bypass capacitor can be placed in parallel with CIN, close to the VIN pin, to shunt any high frequency noise
to ground. The output capacitor, COUT, should also be placed close to the IC. Any copper trace connections for
the COUT capacitor can increase the series resistance, which directly effects output voltage ripple. The feedback
network, resistors RFB1 and RFB2, should be kept close to the FB pin, and away from the inductor, to minimize
copper trace connections that can inject noise into the system. Trace connections made to the inductor and
schottky diode should be minimized to reduce power dissipation and increase overall efficiency. For more detail
on switching power supply layout considerations see Application Note Layout Guidelines for Switching Power
Supplies (SNVA021).
Copyright © 2004–2013, Texas Instruments Incorporated Submit Documentation Feedback 15
Product Folder Links: LM3224
:
RFB2
=
RFB1 1.26
VOUT - 1.26
LM3224 RFB1
160k
L
10 PH
CC
1 nF
D1
RC
30k
COUT1
10 PF
RFB2
30k
SHDN
3
VIN
6
SW
5
FSLCT
7
GND
4
VC
1
C1
4.7 PF
D3
C2
4.7 PF
D2
-8V
8V
VIN = 2.7V - 5.5V
CIN
22 PFCOUT2
10 PF
D6
C6
1 PF
D5
C4
1 PFC5
1 PF
D4 D7
C7
1 PF
23V
SS
8
CSS CC2
68 pF
LM3224
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
www.ti.com
APPLICATION INFORMATION
Figure 23. Triple Output TFT Bias (615 kHz operation)
TRIPLE OUTPUT TFT BIAS
The circuit in Figure 23 shows how the LM3224 can be configured to provide outputs of 8V, −8V, and 23V,
convenient for biasing TFT displays. The 8V output is regulated, while the −8V and 23V outputs are unregulated.
The 8V output is generated by a typical boost topology. The basic operation of the boost converter is described
in the OPERATION section. The output voltage is set with RFB1 and RFB2 by:
(22)
The compensation network of RCand CCare chosen to optimally stabilize the converter. The inductor also
affects the stability. When operating at 615 kHz, a 10uH inductor is recommended to insure the converter is
stable at duty cycles greater than 50%. Refer to the COMPENSATION section for more information.
The -8V output is derived from a diode inverter. During the second cycle, when the transistor is open, D2
conducts and C1 charges to 8V minus a diode drop (≊0.4V if using a Schottky). When the transistor opens in the
first cycle, D3 conducts and C1's polarity is reversed with respect to the output at C2, producing -8V.
16 Submit Documentation Feedback Copyright © 2004–2013, Texas Instruments Incorporated
Product Folder Links: LM3224
2.9V - 4.2V
LM3224
L
4.7 PH
CC
2.2 nF
D
RC
2k
CIN
22 PF
COUT
10 PF
ceramic
RTORCH
6.2:
(200 mA)
SHDN
3
VIN
6SW
5
FSLCT 7
FB 2
GND
4
VC
1
Battery or
Power Source
SS
8
High Current
White LED
Pull high for TORCH.
Pull Flash Enable high
for FLASH.
RFLASH
2.55:
(500 mA)
Torch
Enable
Flash
Enable
2.9V - 4.2V
LM3224
L
4.7 PH
CC
2.2 nF
D
RC
2k
CIN
22 PF
COUT
10 PF
ceramic
RSET
1.8:
(700 mA)
SHDN
3
VIN
6SW
5
FSLCT 7
FB 2
GND
4
VC
1
Battery or
Power Source
SS
8
High Current
White LED
Pull high for FLASH or
constant full current,
PWM for TORCH or
partial current.
Optional
Disconnect
FET
LM3224
www.ti.com
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
The 23V output is realized with a series of capacitor charge pumps. It consists of four stages: the first stage
includes C4, D4, and the LM3224 switch; the second stage uses C5, D5, and D1; the third stage includes C6,
D6, and the LM3224 switch; the final stage is C7 and D7. In the first stage, C4 charges to 8V when the LM3224
switch is closed, which causes D5 to conduct when the switch is open. In the second stage, the voltage across
C5 is VC4 + VD1 - VD5 = VC4 ≊8V when the switch is open. However, because C5 is referenced to the 8V
output, the voltage at C5 is 16V when referenced to ground. In the third stage, the 16V at C5 appears across C6
when the switch is closed. When the switch opens, C6 is referenced to the 8V output minus a diode drop, which
raises the voltage at C6 with respect to ground to about 24V. Hence, in the fourth stage, C7 is charged to 24V
when the switch is open. From the first stage to the last, there are three diode drops that make the output voltage
closer to 24 - 3xVDIODE (about 22.8V if a 0.4V forward drop is assumed).
Figure 24. PWM White LED Flash/Torch Driver
Figure 25. Continuously Operating White LED Flash/Torch Driver
Copyright © 2004–2013, Texas Instruments Incorporated Submit Documentation Feedback 17
Product Folder Links: LM3224
LM3224
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
www.ti.com
The LM3224 can be configured to drive high current white LEDs for the flash and torch functions of a digital
camera, camera phone, or any other similar light source. The flash/torch can be set up with the circuit in
Figure 24 by using the resistor RSET to determine the amount of current that will flow through the LED using the
equation:
ILED = VFB/RSET (23)
If the flash and torch modes will both be used the resistor RSET can be chosen for the higher current flash value.
To flash the circuit pull the SHDN high for the time duration needed for the flash. To enable a lower current torch
mode a PWM signal can be applied to the SHDN pin. The torch current would then be approximately the percent
ON time of the PWM signal multiplied by the flash (or maximum) current. The optional disconnect FET can be
used to eliminate leakage current through the LEDs when the part is off and also to disconnect the LED when
the input voltage exceeds the forward voltage drop of the LED. The maximum output current the LM3224 can
supply in this configuration is shown in Table 1.
Figure 25 is another method of driving a high current white LED. This circuit has a higher component count but
allows the switcher to remain on continuously for torch mode reducing stress on the supply. The two FETs also
double for a disconnect function as described above. In this circuit the device and the torch enable FET are
turned on setting a lower current through the LED. When flash is needed the flash enable FET is turned on to
increase the current for the amount of time desired. The minimum ensured maximum output current for this
circuit is the same as for Figure 24.
Table 1. Maximum LED Drive current
(FSW=1.25MHz, L=4.7µH, LED VFMAX=4V (VOUT=5.26V)
VIN LED Drive Current (mA)
4.2 1077
4.1 1047
4.0 1017
3.9 987
3.8 958
3.7 929
3.6 900
3.5 871
3.4 842
3.3 814
3.2 785
3.1 757
3.0 729
2.9 701
2.8 673
2.7 646
Table 2. Some Recommended Inductors (Others May Be Used)
Manufacturer Inductor Contact Information
Coilcraft DO3316 and DT3316 series www.coilcraft.com
800-3222645
TDK SLF10145 series www.component.tdk.com
847-803-6100
Pulse P0751 and P0762 series www.pulseeng.com
Sumida CDRH8D28 and CDRH8D43 series www.sumida.com
18 Submit Documentation Feedback Copyright © 2004–2013, Texas Instruments Incorporated
Product Folder Links: LM3224
2.9V - 4.2V
LM3224
L
10 PH
CC
1 nF
D
RC
30k
CIN
22 PF
COUT
22PF
ceramic
30k
SHDN
3
VIN
6SW
5
FSLCT 7
FB 2
GND
4
VC
1
Battery or
Power Source
SS
8
160k
8V, 500mA
CC2
68 pF
2.7V - 4.2V
LM3224
L
4.7 PH
CC
1 nF
D
RC
20k
CIN
22 PF
COUT
10 PF
ceramic
6.98k
SHDN
3
VIN
6SW
5
FSLCT 7
FB 2
GND
4
VC
1
Battery or
Power Source
SS
8
20.5k
5V, 650 mA
LM3224
www.ti.com
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
Table 3. Some Recommended Input And Output Capacitors (Others May Be
Used)
Manufacturer Capacitor Contact Information
Vishay Sprague 293D, 592D, and 595D series tantalum www.vishay.com
407-324-4140
Taiyo Yuden High capacitance MLCC ceramic www.t-yuden.com
408-573-4150
ESRD seriec Polymer Aluminum Electrolytic
Cornell Dubilier www.cde.com
SPV and AFK series V-chip series
MuRata High capacitance MLCC ceramic www.murata.com
Figure 26. 1.25MHz, 5V Output
Figure 27. 1.25MHz, 8V Output
Copyright © 2004–2013, Texas Instruments Incorporated Submit Documentation Feedback 19
Product Folder Links: LM3224
2.9V - 4.2V
LM3224
L
15 PH
CC
1.5 nF
D
RC
40.2k
CIN
22 PF
COUT
22PF
ceramic
27.4k
SHDN
3
VIN
6SW
5
FSLCT 7
FB 2
GND
4
VC
1
Battery or
Power Source
SS
8
301k
15V, 220mA
2.9V - 4.2V
LM3224
L
10 PH
CC
1.5 nF
D
RC
40.2k
CIN
22 PF
COUT
22PF
ceramic
39.2k
SHDN
3
VIN
6SW
5
FSLCT 7
FB 2
GND
4
VC
1
Battery or
Power Source
SS
8
332k
12V, 300mA
LM3224
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
www.ti.com
Figure 28. 1.25MHz, 12V Output
Figure 29. 1.25MHz, 15V Output
20 Submit Documentation Feedback Copyright © 2004–2013, Texas Instruments Incorporated
Product Folder Links: LM3224
LM3224
www.ti.com
SNVS277C –DECEMBER 2004–REVISED MARCH 2013
REVISION HISTORY
Changes from Revision B (March 2013) to Revision C Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 20
Copyright © 2004–2013, Texas Instruments Incorporated Submit Documentation Feedback 21
Product Folder Links: LM3224
PACKAGE OPTION ADDENDUM
www.ti.com 11-Apr-2013
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish MSL Peak Temp
(3)
Op Temp (°C) Top-Side Markings
(4)
Samples
LM3224MM-ADJ/NOPB ACTIVE VSSOP DGK 8 1000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 SEKB
LM3224MMX-ADJ/NOPB ACTIVE VSSOP DGK 8 3500 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM SEKB
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) Multiple Top-Side Markings will be inside parentheses. Only one Top-Side Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a
continuation of the previous line and the two combined represent the entire Top-Side Marking for that device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM3224MM-ADJ/NOPB VSSOP DGK 8 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1
LM3224MMX-ADJ/NOPB VSSOP DGK 8 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 11-Oct-2013
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM3224MM-ADJ/NOPB VSSOP DGK 8 1000 210.0 185.0 35.0
LM3224MMX-ADJ/NOPB VSSOP DGK 8 3500 367.0 367.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 11-Oct-2013
Pack Materials-Page 2
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