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Applicati on Not e AN -1177
IRPLLED13
90-250VAC Offline LED Driver
using IRS2980
By Peter B. Green
Table of Contents
Page
1. Introduction ...................................................................................... 2
2. Constant Current Control ................................................................. 3
3. High Voltage Regulator .................................................................... 6
4. High Side Differential Current Sense ............................................... 7
5. Thermal Co nsi der a ti ons ................................................................... 8
6. IRPLLED13 Circuit Schematic ......................................................... 9
7. Bill of Mate rials ................................................................................. 10
8. PCB Layout ...................................................................................... 11
9. Test Results ..................................................................................... 12
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EVALUATION BOARD - IRPLLED13
1. In tr o d uction
Solid state light sources are now available that offer viable alternatives to
Fluorescent and HID lamps and far surpass incandescent lamps. Luminous
efficacy (expr ess ed in Lumens per Watt) has now reached levels enabling LEDs
to be used for general illumination. High brightness LEDs also possess the
added advantages of longer operating life span up to 50000 hours and greater
robustness than other less efficient light sources making them suitable for
outside applications such as street lighting.
High power LEDs are ideally driven with constant regulated DC current,
requiring a "driver" or "converter" to provide the required current from an AC or
DC power source. A simple single stage power converter based around the
IRS2980 LED driver IC provides a controlled current output over a wide AC line
or DC voltage input range.
The IRPLLED13 evaluation board is an off line non-isolated constant current
Buck regulator LED driver designed to supply a nominal 350mA DC output
current. The LED output voltage can be up to 90% of the input voltage,
operating from an AC line input voltage between 90 and 250VAC 50/60Hz. The
IRPLLED13 demo board is not designed for use with dimmers.
Important Safety Information
The IRPLLED13 does not provide galvanic isolation of the LED drive output
from the line input. Therefore if the system is supplied directly from a non-
isolated input, an electrical shock hazard exists at the LED outputs and these
should not be touched during operation.
It is recommended that for laboratory evaluation the IRPLLED13 board be used
with an isolated AC or DC input supply. The IRS2980 series Buck topology is
suitable only for final applications where isolation is either not necessary or
provided elsewhere in the system.
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Figure 1: IRPLLED13 Block Diagram
2. Constant current control
The IRS2980 is a hysteretic Buck controller operating in continuous conduction
mode (CCM) and using a low side switching MOSFET as the controlled switch
and a fast recovery diode as the uncontr ol l ed switch connected to the positive
DC bus. This mode of operation includes a differential floating high side current
sense circuit, which is used to hysteretically control the output current by sensing
the voltage drop across a sense resistor and regulating the average to 0.5V. The
IRS2980 is designed for use in current regulated circuits and not voltage
regulated circuits.
VBUS
COM
1
2
3
4
8
7
6
5
IRS2980
HV
VS
COM ADIM
RAMP
OUT
CS
VCC
ADIM
RCS
RG
LBUCK
MBUCK
CRAMP
CVCC
CBUS
DBUCK
CHVS
CDIM
RDIM
RF
CF
Figure 2: IRS2980 Basic Schematic
Figure 2 illustrates how the current is sensed by diff erentially measuring the
voltage between the HV and CS inputs, RF and CF have been added to provide
noise filt er i ng . When the MOSFET (MBUCK) is switched on the current in the
inductor LBUCK rises linearly according to the relationship:
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dt
di
LbuckVoutVin .=−
[1]
W here Vin is the bus voltage rectified from the AC line voltage and Vout is the
combined series voltage of the string of LEDs making up the load.
W hen the voltage at HV rises to 0.55V with respect to CS the gate drive to
MBUCK switches off. When the MBUCK is off the inductor current flows instead
through DBUCK. During this period the current decreases linearly according to
the relationship:
dt
di
LbuckVout .−=
[2]
W hen the voltage at HV falls to 0.45V with respect to CS the gate drive to
MBUCK switches on. The cycle repeats continuously to provide an average
current in LBUCK which supplies the LED load. The frequency and duty cycle are
dependent on the input and output voltages and the value of the LBUCK as can
be inferr ed fro m the eq uati o ns .
The output current can be set by selecting the appropriate value of RCS
according to the relationship:
RCS
VCS
avgIout =)(
[3]
where VCS is nominally 0.5V, therefore for an RCS of 1.5Ohms, the output
current will be nominally 333mA. In practice there are some additional
propagation delays in the circuit which give rise to some error in the current
regulation and some variation over input voltage, however the perform ance is
acceptable for most LED applications as shown in the test results section.
Accuracy of regulation and amplitude of the current ripple are tradeoffs against
inductor size and frequency.
The IRS2980 based LED driver is designed to operate up to 150k Hz. This is
necessary in order to limit the VCC current consumption since the internal high
voltage regulator can supply only a limited current (ICC) dominat ed b y gate drive
current. Gate current charges and discharges the MOSFET input capacitance
during each switching cycle and therefore increases with frequency.
Using hysteretic regulati on to cont rol the LED current some overshoot occurs
due to propagat ion delays and well as a small undershoot. These vary depending
on di/dt of the ripple current, which is a function of input and output voltage,
inductor value and frequency as well as RC filter valu es (RF and CF). The
average current is maintained over a wide input and output voltage range. Since
the IRS2980S uses hysteretic current control to switch the Buck MOSFET on and
off, the LED current (which is equal to the inductor current) is maintained half
way between minimum and maximum currents. Because of this the switching
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frequency and duty cycle vary to meet the operating conditions imposed by the
input and output voltages, output current and inductor value.
The following diagram shows the rise and fall of the LED current as the MOSFET
switches on and off:
Ton Toff
t
ILED
ILED_AV
VO
(MOSFET GATE)
Figure 3: MOSFET gate drive and inductor/LED current.
The operat ion of the IRS2980S based Buck LED driver can be mode l ed at an
operating point as follows:
First determine the rising and falling slopes of the inductor current.
L
VV
dt
di LEDDCBUS −
=+ _
[4]
LVV
dt
di
FLED
−
=−
[5]
These slo pes ar e then us ed to det er mi ne the amo unt of ov er s hoot a nd
undershoot based on the delay caused by the RC filter, gate charge and
propagation delays.
+⋅++⋅≈ dt
di
t
Q
CRi DR
G
FFOS )
18.0
(
[6]
−⋅++⋅≈ dt
di
t
Q
CRi
DF
G
FFUS
)
26.0
(
[7]
Where,
RF and CF are the current sense filter com ponents,
L is the inductor value,
QG is the MOSFET gate charge,
tDR and tDF are propag ati on del ay s an d ca n be omitted for firs t or der
approximation.
VF is the forward voltage drop of the Buck diode which may assumed as 0.7V.
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Now that the overshoot and undershoot have been determined the on and off
time, frequency and duty cycle can then be calculated.
LEDDCBUS
US
OS
LED
on VV i
iI
L
t−
+
+⋅
⋅
=
_
2.0
[8]
FLED
USOSLED
off VV iiIL
t+
++⋅⋅
=2.0
[9]
offon
SW
tt
f+
=1
(switching frequency) [10]
offon
on
tt t
d+
=
(duty cycle) [11]
Operating freq uency and duty cycle vary since there is a high ripple content in
the DC bus voltage. This is a res ult of the passive valley fill c irc uit used for pow er
factor correction (PFC). The LED current is regulated dynamically by the
IRS2980 continually changing frequency and duty cycle to maintain a constant
average. The passive valle y fill circuit provides power factor between 0.85 and
0.9 depending on the AC input voltage and load, with two capacitors and three
diodes as shown in figure 4. It should be noted that the line current THD will be
about 50% using this technique but this is acceptable for many applications.
Buck LED
Driver
Bus Voltage Waveform
AC Line Input
Figure 4: Passive valley f ill PFC circuit
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Figure 5: Passive valley fill PFC bus voltage (blue) and line input current (red)
waveforms.
The efficiency of the IRPLLED13 like all Buck converters, varies depending on
input and output voltage.
3. High Voltage Regulator
The IRS2980 contains an internal high voltage regulator to supply VCC from the
high voltage DC bus. Figure 2 shows that pin 1 (HV) is connected directly the DC
bus. Current is supplied to the VCC supply at pin 2 through an internal current
source capable of operating up to 450V. The internal regulator can supply up to
3mA, which is sufficient to supply VCC for most MOSFET gate capacitances and
frequencies normally required in an LED driver. ICC can be reduced by selecting
a MOSFET with a low gate capacitance (25nC or less) and selecting an
inductance (LBUCK) that will allow the regulator to operate at a reduced
frequency. A regulator operating at 60kHz for example will require much lower
ICC than one operating at 120kHz. As explained earlier this is a tradeoff against
inductor size. It is also im portant to consider the temperature rise of the IRS2980.
Since the internal regulator operates linearly the associated power loss is
dependent on bus voltage and ICC.
More care must be taken at higher bus voltages to minimize frequency and ICC
to minimize the IC operating temper at ur e. Th e addi ti on o f heat sinking in the form
of areas of copper on the PCB or thermally conductive potting compounds can
significantly reduce temperature. Inductor values are generally larger for 220V off
line AC applications than for 120V in order to reduce switching frequency, which
lowers power dissipation in the circuit.
The VCC current (ICC) drawn by the IRS2980S can be estimated from the
following formula:
SWGCC fQ
mAI⋅+≈ 1
[12]
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therefore power dissipation due to the high voltage regulator can be calculated
as:
CCAVDCBUSREG IVP ⋅= __
[13]
4. High side differential current sense
The IRS2980 uses a floating differential current sense circuit to measure the LED
current in the high side of the supply circuit. The Buck regulator configuration
uses a low side switch unlike the IRS25401 which uses a high side switch and
low side current sense. In order to realize average current control the current
must be sensed both when the MOSFET (MBUCK) is switched on and when it is
switched off and therefore must be sensed at the high side. In order to
accomplish this the hysteretic current sensing circuitry within the IRS2980 is
situated within a floating high side well, constructed by means of International
Rectifiers HVIC technology. A floating supply voltage (nominally 8V) for the
circuitry contained within this high voltage well is developed between the HV and
VS pins of the IC. The supply is provided by a current source located between
VS and COM.
The high side contains a comparator with defined hysteresis connected to a
-0.5V reference with respect to HV. The output from the comparator is
transferred through high voltage level shift circuitry to the gate driver circuitry,
which is referenced to COM.
The floating high side current sense incorporated in the IRS2980S is able to
operate up to 450V and withstand voltage surges up to 600V. A internal bias
supply is derived between the HV and VS pins by a 1mA current source pulling
down on VS so that a supply voltage is produced across the external capacitor
CHVS to supply the high side circuitry. A value of 22nF is recommended for
CHVS. The internal bias supply also dissipates some power, which can be
calculat e d fro m the formula:
mA
VP
AVDCBUSBIAS
1)10(
__
⋅−
=
[14]
In order for the high side current sense circuitry to function, a minimum bus
voltage of 30V is required to provide adequate bias supply current and stand off
voltage.
5. Thermal considerations
Since the IRS2980S dissipates some power during normal operation,
temperature rise of the IC die must be considered as part of the design process.
The SO8 IC package has a maximum power rating (PD) of 625mW, therefore the
sum of PREG and PBIAS should not exc eed thi s value. The junction temperature
should remain below 125°C to ensure operation within specifications.
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The junction temperature is normally 10°C above the case temperature for an
SO8 package therefore the case temperature should not exceed 115°C at
maximum ambient.
In order to reduce the junction temperature rise thermal relief has been added
around the IRS2980S on the IRPLLED13 PCB layout. With adequat e thermal
relief the die temperature rise is greatly reduced. An area of copper has been
placed on the opposite side of the PCB to the IC in the same position with
several through hole vias added underneath the IC to conduct heat through to
the other side. In addition an area of this thermal relief area has been left without
solder resist to aid dissipation.
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6. IRPLLED13 Circuit Schematic
1VOUT+
1VOUT-
COM
4HV 1
VS
2
VCC
3
ADIM 5
RAMP 6
OUT 7
CS 8
IC1
IRS2980
R2
1.5 / 1W
R5
10
C6
0.1uF
L2
3.3mH / 0.5A
M1
IRFR812
D1
MURS120-13
C4
1nF
R4
100
C7
22nF
1
VAC1
1
VAC2
3
1
4
2
BR1
DF10S
D2
S1G-13F
D3
S1G-13F
D4
S1G-13F
C3
0.1uF / 630V
R3
10/1W
C1
100nF / 250V
+
C2
10uF / 200V
+
C5
10uF/200V
R1
10 / 1W
L1
1mH / 0.5A
Figure 4: IRPLLED13 Complete Schematic
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7. Bill of Materials
Item Description Part Number Manufacurer Quantity Reference
1 IC, LED Controller IRS2980S
International
Rectifier
1 IC1
2
Rectifier ,1A , 400V,
SMA
S1G-13-F Diodes Inc 3 D2,D3,D4
3
Diode, 1A, 600V, 35n S,
SMB
MURHS160T3G On Semi 1 D1
4
Bridge, 1000V, 1.5A,
4SDIP
DF10S Fairchild 1 BR1
5
MOSFET, 500V,
2.2Ohm, DPAK
IRFR812
International
Rectifier
1 M1
6
Capacitor, 100nF,
250VAC, Radial
B32652A6104J Epcos 1 C1
7
Capacitor, 1nF, 50V,
5%, 1206
1206A102JAT2A AVX 1 C4
8
Capacitor, 0.1uF, 50V,
10%, 1206
GRM319R71H10
4KA01D
Murata 1 C6
9
Capacitor, 10uF,
200VDC, 20%
EKXG201ELL100
MJ16S
United Chem 2 C2, C5
10
Capacitor, 0.10uF,
630V, 1812, X7R
GRM43DR72J10
4KW01L
Murata 1 C3
11
Capacitor, 22nF, 50V,
1206
12065C223KAT2
A
AVX 1 C7
12
Resistor, 1.5Ohm, 1W,
5%, 2512
ERJ-1TYJ1R5U Panasonic 1 R2
13
Resistor, 10Ohm, 1W,
5%, Axial
PR01000101009
JR500
Vishay 2 R1, R3
14
Resistor, 1kOhm,
0.25W, 5%, 1206
ERJ-8GEYJ102V Panasonic 1 R4
15
Resistor, 10, 0.25W,
5%, 1206
ERJ-8GEYJ100V Panasonic 1 R5
16
Inductor, 1mH, 0.55A,
1.68Ohm
B82477G4105M Epcos 1 L1
17
Inductor, 3.3mH, 0.44A,
Radial
TSL1315RA-
332JR44-PF
TDK 1 L2
18
Test point, 0.063"D
Yellow
5009 Keystone 2
19
Test point, 0.063"D
Red
5005 Keystone 1
20
Test point, 0.063"D
Black
5006 Keystone 1
21 PCB
IRPLLED13 Rev
D
1
22
23
24
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8. PCB Layout
Top Overlay Top Copper
Bottom Overlay Bottom Copper
Figure 5: PCB Layout
Layout Considerations
It is very important when laying out the PCB for the IRS2 980 based LED driver to
consider the fol l owing points:
1. CVCC (C6) and CHVS (C8) must be as close to IC1 as possible.
2. The feedback path should be kept to a minimum length and separated as
much as possibl e from high frequency switching traces to minimize noise
at the CS input.
3. The current sense filter components RF (R4) and CF (C4) should be
locate d close to t he IR S298 0 w i th short di r ect traces .
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4. It is essential that all signal and power grounds should be kept separated
from each other to prevent noise from entering the control environment.
Signal and power grounds should be connected together at one point only,
which must be at the COM pin of the IRS2980. The IRS2980 may not
operate i n a stabl e ma nner if these guidelines are not followed!
All low side components associated with the IC should be connected to
the IC signal ground (COM) with the shortest path possible.
5. All traces carrying the load current need to be sized accordingly.
6. Gate drive traces should also be kept to a minimum length.
9. Test Results
Measurements were carried out using a variable AC supply and a load of 7 white
LEDs being driven at a nominal 350mA.
Calculated
Measured
Line Voltage
Minimum
Maximum
Minimum
Maximum
120VAC
64.3
73.9
65.8
72.2
220VAC
73.1
78.3
65.0
72.8
Table 1: Predicted frequency and measured freq uency (kHz):
The minimum frequency occurs when the bus voltage is at its lowest which is
estimated to be half the peak voltage in the passive valley fill circuit. The
maximu m freq ue ncy occurs when the bus voltage reaches its peak. Table 1
shows that a first order calculation described in section 2 gives a frequency close
to the actual frequency measured. The frequency is likely to vary from board to
board due to component tolerances.
AC Input
Voltage
(V)
Input
Power
(W)
Output
Voltage
(V)
Output
Current
(mAav)
Ripple
(mApp)
Output
Power
(W)
80
8.58
20.1
339
62
6.81
100
8.57
20.1
339
68
6.81
120
8.67
20.1
340
71
6.83
140
8.79
20.1
341
73
6.85
160
8.83
20.1
342
80
6.87
180
9.97
20.0
343
80
6.87
200
9.12
20.0
344
82
6.89
220
9.26
20.0
345
81
6.90
240
9.40
20.0
346
82
6.93
260
9.55
20.0
348
84
6.95
Table 2: IRPLLED13 T est Resu lts
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Figure 6: LED Current at 120VAC Input
Figure 7: LED Current at 220VAC Input
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The case temperature of the IRS2980 (IC1), the IRFR812 (M1) and the Buck
diode (D1) were measured with a thermocouple attached to each device after 30
minutes of operation with the board positioned with the discrete component side
facing up.
AC Line Voltage (Vrms)
120
220
AC Line Current (mArms)
114
65
Input Power (W)
12.21
12.98
Power Factor
0.892
0.896
Output Voltage (Vdc)
30.5
30.5
Output Current (mAdc)
340.9
352.6
Output Power (W)
10.40
10.65
Frequency (kHz)
60.1
50.2
IC1 Case Temperature (ºC)
56.5
75.0
M1 Case Temperature (ºC)
57.0
71.3
D1 Case Temper ature (ºC)
66.5
74.6
Table 3: Component Temperatures
Component temperatures should be kept below their maximum rating and as low
as possible at maxim um ambient temperature. Reducing frequency by increasing
the inductor size or increasing the value of filter resistor R4 reduces power
dissipation due to switching losses. Thermally conductive potting material is often
used in LED products, which conducts heat away from the components.
If the value of R4 is increased to low er freq uency rather than increasing inductor
size, the output current ripple increases. This leads to larger variation in current
with input voltage because of the increased delay. It is important to consider that
the current sens e filter is necessary f or the circuit to work correctly. Without this
filter noise transients caused by high dv/dt resulting from switching cause fals e
triggering of the hysteretic circuit, which causes the IRS2980 to det e ct a fault and
shut dow n. The values of C4 and R4 required for correct operation depe nd on
the MOSFET parasitic capacitances, the diode reverse recovery time and how
well the PCB layout shields the IFB input from noise. C4 must be pl a ced as close
to the IRS2980 as possible with short traces.
IR WORLD HEADQUARTERS: 101 N. Sepulveda Blvd., El Segundo, Calif orni a 90245 Tel: (310) 252-7105
Data and specifications subject to change without notice. 2/17/2014