April 2009 Rev 3 1/47
47
LED7707
6-rows 85 mA LEDs driver with boost regulator
for LCD panels backlight
Features
■Boost section
– 4.5 V to 36 V input voltage range
– Internal power MOSFET
– Internal +5 V LDO for device supply
– Up to 36 V output voltage
– Constant frequency peak current-mode
control
– 250 kHz to 1 MHz adjustable switching
frequency
– External synchronization for multi-device
application
– Pulse-skip power saving mode at light load
– Programmable soft-start
– Programmable OVP protection
– Stable with ceramic output capacitors
– Thermal shutdown
■Backlight driver section
– Six rows with 85 mA maximum current
capability (adjustable)
– Rows disable option
– Less than 10 μs minimum dimming on-time
– ±2 % current matching between rows
– LED failure (open and short-circuit)
detection
Applications
■LCD monitors and TV panels
■PDAs panel backlight
■GPS panel backlight
Description
The LED7707 consists of a high efficiency
monolithic boost converter and six controlled
current generators (rows) specifically designed to
supply LEDs arrays used in the backlighting of
LCD panels. The device can manage an output
voltage up to 36 V (i.e. 10 white LEDs per row).
The generators can be externally programmed to
sink up to 85 mA and can be dimmed via a PWM
signal (1 % dimming duty-cycle at 1 kHz can be
managed). The device allows to detect and
manage the open and shorted LED faults and to
let unused rows floating. Basic protections (output
over-voltage, internal MOSFET over-current and
thermal shutdown) are provided.
VFQFPN-24 4x4
Table 1. Device summary
Order codes Package Packaging
LED7707 VFQFPN-24 4x4 (exposed pad) Tu b e
LED7707TR Tape and reel
www.st.com
Contents LED7707
2/47
Contents
1 Typical application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
2 Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.1 Connections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.2 Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
3 Electrical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
3.1 Maximum rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
3.2 Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
4 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
5 Operation description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
5.1 Boost section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
5.1.1 Functional description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
5.1.2 Enable function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
5.1.3 Soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
5.1.4 Over-voltage protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
5.1.5 Switching frequency selection and synchronization . . . . . . . . . . . . . . . 14
5.1.6 Slope compensation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
5.1.7 Boost current limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
5.1.8 Thermal protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
5.2 Backlight driver section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
5.2.1 Current generators . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
5.2.2 PWM dimming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
5.3 Fault management . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
5.3.1 FAULT pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
5.3.2 MODE pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
5.3.3 Open LED fault . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
5.3.4 Shorted LED fault . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
LED7707 Contents
3/47
6 Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
6.1 System stability . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
6.1.1 Loop compensation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
6.2 Thermal considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
6.3 Component selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
6.3.1 Inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
6.3.2 Capacitors selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
6.3.3 Flywheel diode selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
6.4 Design example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
6.4.1 Switching frequency setting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
6.4.2 Row current setting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
6.4.3 Inductor choice . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
6.4.4 Output capacitor choice . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
6.4.5 Input capacitor choice . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
6.4.6 Over-voltage protection divider setting . . . . . . . . . . . . . . . . . . . . . . . . . 35
6.4.7 Compensation network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
6.4.8 Boost current limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
6.4.9 Power dissipation estimate . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
6.5 Layout consideration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
7 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
8 Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
9 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
Typical application circuit LED7707
4/47
1 Typical application circuit
Figure 1. Application circuit
ROW1
ROW2
ROW3
ROW4
ROW5
ROW6
LDO5
BILIM
RILIM
SS
SGND
SLOPE
VIN
LX
OVSEL
SWF
AVCC
FAULT
EN
SYNC
MODE
PGND
DIM
VIN VOUT
COMP
+5V
Up to 10 WLEDs per row
Enable
Dimming
Fault
LED7707
L
MLCC
Sync Output
Faults Management Selection
OVP selection
Switching Frequency selection
Slope Compensation
Rows current selection
Internal MOS OCP
AM00579v1
LED7707 Pin settings
5/47
2 Pin settings
2.1 Connections
Figure 2. Pin connection (through top view)
FAULT
LED7706
1
6
712
13
18
19
24
VIN
SLOPE
SGND
ROW1
ROW2 LX
DIM
EN
SYNC
SS
ROW3
ROW4
ROW5
ROW6
PGND
OVSEL
COMP
RILIM
FSW
BILIM
MODE
AVCC
LDO5
Pin settings LED7707
6/47
2.2 Pin description
Table 2. Pin functions
N° Pin Function
1COMP
Error amplifier output. A simple RC series between this pin and ground is
needed to compensate the loop of the boost regulator.
2RILIM
Output generators current limit setting. The output current of the rows can be
programmed connecting a resistor to SGND.
3 BILIM Boost converter current limit setting. The internal MOSFET current limit can
be programmed connecting a resistor to SGND.
4FSW
Switching frequency selection and external sync input. A resistor to SGND is
used to set the desired switching frequency. The pin can also be used as
external synchronization input. See Section 5.1.5 on page 14 for details.
5MODE
Current generators fault management selector. It allows to detect and manage
LEDs failures. See Section 5.3.2 on page 22 for details.
6 AVCC + 5 V analog supply. Connect to LDO5 through a simple RC filter.
7LDO5
+ 5 V LDO output and power section supply. Bypass to SGND with a
1 µF ceramic capacitor.
8 VIN Input voltage. Connect to the main supply rail.
9SLOPE
Slope compensation setting. A resistor between the output of the boost
converter and this pin is needed to avoid sub-harmonic instability.
Refer to Section 6.1 on page 25 for details.
10 SGND Signal ground. Supply return for the analog circuitry and the current
generators.
11 ROW1 Row driver output #1.
12 ROW2 Row driver output #2.
13 ROW3 Row driver output #3.
14 ROW4 Row driver output #4.
15 ROW5 Row driver output #5.
16 ROW6 Row driver output #6.
17 PGND Power ground. Source of the internal power MOSFET.
18 OVSEL Over-voltage selection. Used to set the desired 0 V threshold by an external
divider. See Section 5.1.4 on page 14 for details.
19 LX Switching node. Drain of the internal power MOSFET.
20 DIM Dimming input. Used to externally set the brightness by using a PWM signal.
21 EN Enable input. When low, the device is turned off. If tied high or left open, the
device is turned on and a soft-start sequence takes place.
22 FAULT Fault signal output. Open drain output. The pin goes low when a fault condition
is detected (see Section 5.3.1 on page 22 for details).
23 SYNC Synchronization output. Used as external synchronization output.
24 SS Soft-start. Connect a capacitor to SGND to set the desired soft-start duration.
LED7707 Electrical data
7/47
3 Electrical data
3.1 Maximum rating
3.2 Thermal data
Table 3. Absolute maximum ratings (1)
1. Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the
device. Exposure to absolute maximum rated conditions for extended periods may affect device reliability.
Symbol Parameter Value Unit
VAVCC AVCC to SGND -0.3 to 6
V
VLDO5 LDO5 to SGND -0.3 to 6
PGND to SGND -0.3 to 0.3
VIN VIN to PGND -0.3 to 40
VLX LX to SGND -0.3 to 40
LX to PGND -0.3 to 40
RILIM, BILIM, SYNC, OVSEL, SS to SGND -0.3 to VAVCC + 0.3
EN, DIM, SW, MODE, FAULT to SGND -0.3 to 6
ROWx to PGND/ SGND -0.3 to 40
SLOPE to VIN VIN - 0.3 to VIN + 6
SLOPE to SGND -0.3 to 40
Internal switch maximum RMS current
(flowing through LX node) 2.0 A
PTOT Power dissipation @ TA = 25 °C 2.3 (2)
2. Power dissipation referred to the device mounted on the demonstration board described in section 5.5
W
Maximum withstanding voltage range test condition:
CDF-AEC-Q100-002- “human body model” acceptance
criteria: “normal performance”
±1000 V
Table 4. Thermal data
Symbol Parameter Value Unit
R
thJA
Thermal resistance junction to ambient 42 °C/W
T
STG
Storage temperature range -50 to 150 °C
T
J
Junction operating temperature range -40 to 150 °C
Electrical characteristics LED7707
8/47
4 Electrical characteristics
VIN = 12 V; TJ = 25 °C and LDO5 connected to AVCC if not otherwise specified (a)
a. Specification referred to TJ from 0 °C to +85 °C. Specification over the 0 to +85 °C TJ range are assured by
design, characterization and statistical correlation.
Table 5. Electrical characteristics
Symbol Parameter Test condition Min. Typ. Max. Unit
Supply section
VIN Input voltage range 4.5 36 V
VBST Boost section output voltage 36 V
VLDO5 LDO output and IC supply
voltage
EN high
ILDO5 = 0 mA 4.4 5 5.5 V
VAVCC
IIN,Q Operating quiescent current
RRILIM = 51 kΩ, RBILIM = 220 kΩ,
RSLOPE = 680 kΩ
DIM tied to SGND.
1mA
IIN,SHDN Operating current in shutdown EN low 20 30 μA
VUVLO,ON
LDO5 under voltage lock out
upper threshold 4.0 4.3
V
VUVLO,OFF
LDO5 under voltage lock out
lower threshold 3.5 3.7
LDO linear regulator
Line regulation 6 V ≤ VIN ≤ 36 V, ILDO5 = 30 mA 30 mV
LDO dropout voltage ILDO5 = 10 mA (-10 % drop) 80 120
LDO maximum output current VLDO5 > VUVLO,ON 25 40 60 mA
VLDO5 < VUVLO,OFF 20 30
Boost section
tON,min Minimum switching on-time 200 ns
fSW Default switching frequency FSW connected to AVCC 570 660 770 kHz
Minimum FSW sync frequency 220
FSW sync input low level 240
mVFSW sync input high level 350
FSW sync input hysteresis 30
FSW sync min ON time 270 %
SYNC output duty-cycle FSW connected to AVCC
(Internal oscillator selected) 34 40 %
SYNC output high level ISYNC = 10 μAVAVCC
-20V mV
SYNC output low level ISYNC = -10 μA 20
LED7707 Electrical characteristics
9/47
Symbol Parameter Test condition Min. Typ. Max. Unit
Power switch
KBLX current coefficient RBILIM = 600 kΩ1⋅1061.2⋅1061.4⋅106V
RDS(on) Internal MOSFET on-resistance 280 500 mΩ
OC and OV protections
VTH,OVP
Over-voltage protection
reference threshold 1.145 V
Soft-start and power management
EN, Turn-on threshold 1.6
V
EN, Turn-off threshold 0.8
DIM, high level threshold 1.3
DIM, low level threshold 0.8
EN, pull-up current 2.5 μA
SS, charge current 4 5 6
SS, end-of-startup threshold 1.8 2.4 2.6
V
SS, reduced switching
frequency release threshold 0.8
Current generators section
KRCurrent generators gain 1850 V
ΔKR(1) Current generators gain
accuracy ±2.0 %
VIFB Feedback regulation voltage 700 750 mV
Vrowx,
FAULT
LED short circuit detection
threshold MODE tied to SGND 4.0 V
VFAULT,
LOW
FAULT pin low-level voltage IFAULT,SINK = 4 mA 250 380 mV
Thermal shutdown
TSHDN
Thermal shutdown
turn-off temperature 150 °C
Thermal shutdown hysteresis 30
1. IROW = KR / RRILIM, ΔIROW/IROW ≈ ΔKR/KR+ ΔRRILIM/RRILIM
Table 5. Electrical characteristics (continued)
Operation description LED7707
10/47
5 Operation description
The device can be divided into two sections: the boost section and the backlight driver
section. These sections are described in the next paragraphs.
Figure 3 provides an overview of the internal blocks of the device.
Figure 3. Simplified block diagram
LDO5
BILIM
COMP
SYNC
FSW
RILIM
EN
FAULT
ROW4
PGND
LX
ROW6
DIM
VIN
ROW1
SLOPE
AVCC
ROW3
CONTROL
LOGIC
ROW2
+
_
Thermal
Shutdown
ROW5
OVSEL
+
_
SGND
Ext Sync
Detector OSC
OVP
Boost_EN
1.2V
4V
MODE
Boost_EN
0.7V
Min Voltage
Selector
UVLO
UVLO
Detector
+5V
LDO
Ramp
Generator
Current Sense
1.172V
Soft Start
OVP
CTRL1
CTRL6
V
ROW1
V
ROW2
V
ROW5
V
ROW4
V
ROW3
V
ROW6
CTRL2
V
TH,FLT
CTRL5
CTRL3
CTRL4
CTRL2
UVLO
CTRL3
CTRL4
CTRL5
CTRL6
Current
Generator 2
Current
Generator 3
Current
Generator 4
Current
Generator 5
Current
Generator 6
Current
Generator 1
Current Limit
++
LOGIC
Boost
Control
Logic
ZCD
+
_
g
m
+
_
I to V
I to V
SS
MODE
÷2
Prot_EN
Prot_EN
LDO5
BILIM
COMP
SYNC
FSW
RILIM
EN
FAULT
ROW4
PGND
LX
ROW6
DIM
VIN
ROW1
SLOPE
AVCC
ROW3
CONTROL
LOGIC
ROW2
+
_
Thermal
Shutdown
ROW5
OVSEL
+
_
+
_
SGND
Ext Sync
Detector OSC
OVP
Boost_EN
1.2V
4V
MODE
Boost_EN
0.7V
Min Voltage
Selector
UVLO
UVLO
Detector
+5V
LDO
Ramp
Generator
Current Sense
1.172V
Soft Start
OVP
CTRL1
CTRL6
V
ROW1
V
ROW2
V
ROW5
V
ROW4
V
ROW3
V
ROW6
CTRL2
V
TH,FLT
CTRL5
CTRL3
CTRL4
CTRL2
UVLO
CTRL3
CTRL4
CTRL5
CTRL6
Current
Generator 2
Current
Generator 3
Current
Generator 4
Current
Generator 5
Current
Generator 6
Current
Generator 1
Current Limit
++
LOGIC
Boost
Control
Logic
ZCD
+
_
g
m
+
_
I to V
I to V
SS
MODE
÷2
Prot_EN
Prot_EN
LED7707 Operation description
11/47
5.1 Boost section
5.1.1 Functional description
The LED7707 is a monolithic LEDs driver for the backlight of LCD panels and it consists of a
boost converter and six PWM-dimmable current generators.
The boost section is based on a constant switching frequency, peak current-mode
architecture. The boost output voltage is controlled such that the lowest row's voltage,
referred to SGND, is equal to an internal reference voltage (700 mV typ.). The input voltage
range is from 4.5 V up to 36 V. In addition, the LED7707 has an internal LDO that supplies
the internal circuitry of the device and is capable to deliver up to 40 mA. The input of the
LDO is the VIN pin.
The LDO5 pin is the LDO output and the supply for the power MOSFET driver at the same
time. The AVCC pin is the supply for the analog circuitry and should be connected to the
LDO output through a simple RC filter in order to improve the noise rejection.
Figure 4. AVCC filtering
Two loops are involved in regulating the current sunk by the generators.
The main loop is related to the boost regulator and uses a constant frequency peak current-
mode architecture to regulate the power rail that supplies the LEDs (Figure 5), while an
internal current loop regulates the same current (flowing through the LEDs) at each row
according to the set value (RILIM pin).
Figure 5. Main loop and current loop diagram
V
IN
0.7V
PWM
Minimum voltage drop
selector
COMP
LX ROWx
SGND
RILIM
E/A
Slope
Error amplifier
AM00582v1
Operation description LED7707
12/47
A dedicated circuit automatically selects the lowest voltage drop among all the rows and
provides this voltage to the main loop that, in turn, regulates the output voltage. In fact, once
the reference generator has been detected, the error amplifier compares its voltage drop to
the internal reference voltage and varies the COMP output. The voltage at the COMP pin
determines the inductor peak current at each switching cycle. The output voltage of the
boost regulator is thus determined by the total forward voltage of the LEDs strings (see
Figure 6):
Equation 1
where the first term represents the highest total forward voltage drop over N active rows and
the second is the voltage drop across the leading generator (700 mV typ.).
The device continues to monitor the voltage drop across all the rows and automatically
switches to the current generator having the lowest voltage drop.
Figure 6. Calculation of the output voltage of the boost regulator
5.1.2 Enable function
The LED7707 is enabled by the EN pin. This pin is active high and, when forced to SGND,
the device is turned off. This pin is connected to a permanently active 2.5 µA current source;
when sudden device turn-on at power-up is required, this pin must be left floating or
connected to a delay capacitor. Starting from an ON state, when the LED7707 is turned off,
it quickly discharges the Soft-Start capacitor and turns off the power-MOSFET, the current
generators and the LDO. The power consumption is thus reduced to 20 µA only.
In applications where the dimming signal is used to turn on and off the device, the EN pin
can be connected to the DIM pin as shown in Figure 7.
mV700)V(maxV j,F
m
1j
N
1i
BST
LEDS
ROWS += Σ
=
=
700 mV
I
LED
Row with the highest voltage
drop across LEDs
V
BOOST
V
IN
max Σ
V
F
Leading
generator
Boost
controller
Current
generators
section
AM00583v1
LED7707 Operation description
13/47
Figure 7. External sync waveforms
5.1.3 Soft-start
The soft-start function is required to perform a correct start-up of the system, controlling the
inrush current required to charge the output capacitor and to avoid output voltage overshoot.
The soft-start duration is set connecting an external capacitor between the SS pin and
ground. This capacitor is charged with a 5 μA (typ.) constant current, forcing the voltage on
the SS pin to ramp up. When this voltage increases from zero to nearly 1.2 V, the current
limit of the power MOSFET is proportionally released from zero to its final value. However,
because of the limited minimum on-time of the switching section, the inductor might saturate
due to current runaway. To solve this problem the switching frequency is reduced to one half
of the nominal value at the beginning of the soft-start phase. The nominal switching
frequency is restored after the SS pin voltage has crossed 0.8 V.
Figure 8. Soft-start sequence waveforms in case of floating rows
During the soft-start phase the floating rows detection is also performed. In presence of one
or more floating rows, the voltage across the involved current generator drops to zero. This
voltage becomes the inverting input of the error amplifier through the minimum voltage drop
selector (see Figure 5). As a consequence the error amplifier is unbalanced and the loop
DIM
EN
SGND
LED7707
220k 100n
BAS69
AM00584v1
AM00585v1
AVCC
t
ss
SS pin voltage
t
Protections turn active
1.2V
0.8V
Current limit
EN pin voltage
100%
Output voltage
OVP
95% of
OVP
Floating ROWs detection
2.4V
Nominal switching
frequency release
AVCC
t
ss
SS pin voltage
t
Protections turn active
1.2V
0.8V
Current limit
EN pin voltage
100%
Output voltage
OVP
95% of
OVP
Floating ROWs detection
2.4V
Nominal switching
frequency release
Operation description LED7707
14/47
reacts by increasing the output voltage. When it reaches the floating row detection (FRD)
threshold (which coincides with the OVP threshold, see Section 5.1.4), the floating rows are
managed according to Ta b l e 6 (see Section 5.3 on page 21). After the SS voltage reaches a
2.4 V threshold, the start-up finishes and all the protections turn active. The soft-start
capacitor CSS can be calculated according to equation 2.
Equation 2
Where ISS = 5 µA and tSS is the desired soft-start duration.
5.1.4 Over-voltage protection
An adjustable over-voltage protection is available. It can be set feeding the OVSEL pin with a
partition of the output voltage. The voltage of the central tap of the divider is thus compared
to a fixed 1.145 V threshold. When the voltage of the OVSEL pin exceeds the OV threshold,
the switching activity is suspended. It is resumed as OVSEL returns below the OV threshold.
A 10 mV hysteresis is provided. No device turn-off is performed. Normally, the value of the
high-side resistors of the divider is in the order of 100 kΩ to reduce the output capacitor
discharge when the boost converter is off (during the off phase of the dimming cycle),
whereas the low-side resistor can be calculated as:
Equation 3
An additional filtering capacitor CF (typically in the 100 pF-330 pF range) may be required to
improve noise rejection at the OVSEL pin (see Figure 9).
Figure 9. OVP threshold setting
5.1.5 Switching frequency selection and synchronization
The switching frequency of the boost converter can be set in the 250 kHz-1 MHz range by
connecting the FSW pin to ground through a resistor. Calculation of the setting resistor is
made using equation 4 and should not exceed the 100 kΩ-400 kΩ range.
4.2
tI
CSSSS
SS
⋅
≅
V145.1V4V
V145.1
RR
MAX,OUT
12 −+
⋅=
AM00586v1AM00586v1
LX
OVSEL
SGND
LED7707
V
OUT
V
IN
R
1
R
2
C
F
C
OUT
LED7707 Operation description
15/47
Equation 4
In addition, when the FSW pin is tied to AVCC, the LED7707 uses a default 660 kHz fixed
switching frequency, allowing to save a resistor in minimum component-count applications.
Figure 10. Multiple device synchronization
The FSW pin can also be used as synchronization input, allowing the LED7707 to operate
both as master or slave device. If a clock signal with a 220 kHz minimum frequency is
applied to this pin, the device locks synchronized. The signal provided to the FSW pin must
cross the 270 mV threshold in order to be recognized. The minimum pulse width which
allows the synchronizing pulses to be detected is 270 ns. An Internal time-out allows
synchronization as long as the external clock frequency is greater than 220 kHz.
Keeping the FSW pin voltage lower than 270 mV for more than 4.5 µs results in a stop of the
device switching activity. Normal operation is resumed as soon as FSW rises above the
mentioned threshold and the soft-start sequence is repeated.
The SYNC pin is a synchronization output and provides a 35 % (typ.) duty-cycle clock when
the LED7707 is used as master or a replica of the FSW pin when used as slave. It is used to
connect multiple devices in a daisy-chain configuration or to synchronize other switching
converters running in the system with the LED7707 (master operation). When an external
synchronization clock is applied to the FSW pin, the internal oscillator is over-driven: each
switching cycle begins at the rising edge of clock, while the slope compensation (Figure 11)
ramp starts at the falling edge of the same signal. Thus, to prevent sub-harmonic instability
(see Section 5.1.6), the external synchronization clock is required to have a 40 % maximum
duty-cycle when the boost converter is working in continuous-conduction mode (CCM) in
order to assure that the slope compensation is effective (starts with duty-cycle lower than
40%)
5.2
F
RSW
FSW =
SYNC
SGND
LED7707
AVCC
R
FSW
FSW SYNC
SGND
LED7707
SLAVE
FSW
MASTER
Sync Out
SYNC
AM00587v1
Operation description LED7707
16/47
Figure 11. External sync waveforms
5.1.6 Slope compensation
The constant frequency, peak current-mode topology has the advantage of very easy loop
compensation with output ceramic caps (reduced cost and size of the application) and fast
transient response. In addition, the intrinsic peak-current measurement simplifies the
current limit protection, avoiding undesired saturation of the inductor.
On the other side, this topology has a drawback: there is an inherent open loop instability
when operating with a duty-ratio greater than 0.5. This phenomenon is known as “Sub-
Harmonic Instability” and can be avoided by adding an external ramp to the one coming
from the sensed current. This compensating technique, based on the additional ramp, is
called “slope compensation”. In Figure 12, where the switching duty-cycle is higher than 0.5,
the small perturbation ΔIL dies away in subsequent cycles thanks to the slope compensation
and the system reverts to a stable situation.
The SLOPE pin allows to properly set the amount of slope compensation connecting a
simple resistor RSLOPE between the SLOPE pin and the output. The compensation ramp
starts at 35% (typ.) of each switching period and its slope is given by the following equation:
Equation 5
Where KS = 5.8 ⋅1010 s-1, VBE = 2 V (typ.) and SE is the slope ramp in [A/s].
To avoid sub-harmonic instability, the compensating slope should be at least half the slope
of the inductor current during the off-phase when the duty-cycle is greater than 50%. The
value of RSLOPE can be calculated according to equation 6.
Slave SYNC pin voltage
270mV threshold
FSW pin voltage (ext. sync)
Slave LX pin voltage
270ns minimum
AM00588v1
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−−
=
SLOPE
BEINOUT
SE R
VVV
KS
LED7707 Operation description
17/47
Equation 6
Figure 12. Effect of slope compensation on small inductor current perturbation (D > 0.5)
)VV(
)VVV(LK2
R
INOUT
BEINOUTS
SLOPE −
−−⋅⋅⋅
≤
AM00589v1
T
SW
Programmed inductor peak current with
slope compensation (SE)
Inductor current (CCM)
t
I
BOOST, PEAK
0.35
·T
SW
ΔI
L
Inductor current
perturbation
T
SW
Programmed inductor peak current with
slope compensation (SE)
Inductor current (CCM)
t
I
BOOST, PEAK
0.35
·T
SW
ΔI
L
Inductor current
perturbation
Operation description LED7707
18/47
5.1.7 Boost current limit
The design of the external components, especially the inductor and the flywheel diode, must
be optimized in terms of size relying on the programmable peak current limit. The LED7707
improves the reliability of the final application giving the way to limit the maximum current
flowing into the critical components. A simple resistor connected between the BILIM pin and
ground sets the desired value. The voltage at the BILIM pin is internally fixed to 1.23 V and
the current limit is proportional to the current flowing through the setting resistor, according
to the following equation:
Equation 7
where
The maximum allowed current limit is 5 A, resulting in a minimum setting resistor
RBILIM > 240 kΩ. The maximum guaranteed RMS current in the power switch is 2 A.
In a boost converter the RMS current through the internal MOSFET depends on both the
input and output voltages, according to equations 8a (DCM) and 8b (CCM).
The current limitation works by clamping the COMP pin voltage proportionally to RBILIM.
Peak inductor current is limited to the above threshold decreased by the slope
compensation contribution.
Equation 8 a
Equation 8 b
5.1.8 Thermal protection
In order to avoid damage due to high junction temperature, a thermal shutdown protection is
implemented. When the junction temperature rises above 150 °C (typ.), the device turns off
both the control logic and the boost converter and holds the FAULT pin low. The LDO is kept
alive and normal operation is automatically resumed after the junction temperature has
been reduced by 30 °C.
BILIM
B
PEAK,BOOST R
K
I=
KB1.2 106V⋅=
3
D
LF
DV
I
SW
IN
rms,MOS ⋅
⋅
=
() ()()
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
⋅⋅
+
−
=3
2
SWOUT
OUT
2
OUTrms,MOS D1D
LfI
V
12
1
D1
D
II
LED7707 Operation description
19/47
5.2 Backlight driver section
5.2.1 Current generators
The LED7707 is a LEDs driver with six channels (rows); each row is able to drive multiple
LEDs in series (max. 36 V) and to sink up to 85 mA maximum current, allowing to manage
different kinds of LEDs.
The LEDs current can be set by connecting an external resistor (RRILIM) between the RILIM
pin and ground. The voltage across the RILIM pin is internally set to 1.23 V and the rows
current is proportional to the RILIM current according to the following equation:
Equation 9
Where KR = 1850 V.
The graph in Figure 13 better shows the relationship between IROW and RRILIM and helps to
choose the correct value of the resistor to set the desired row current.
Figure 13. Row current vs RRILIM
The maximum current mismatch between the rows is ± 2 % @ Irowx = 60 mA.
The LED7707 allows parallelism different rows if required by the application. If the maximum
current provided by a single row (85 mA) is not enough for the load, two or more current
generators can be connected together, as shown in Figure 14. To keep the parallelism
generators stable, the row current should be higher than 40 mA.The connection between
channels in parallel must be done as close as possible to the device in order to minimize
parasitic inductance.
RILIM
R
ROWx R
K
I=
AM00590v1
Operation description LED7707
20/47
Figure 14. Rows parallelism for higher current
ROW0
ROW1
ROW2
ROW3
ROW4
ROW5
VCC
BILIM
RILIM
SS
SGND
SLOPE
VIN
LX
OVSEL
SWF
AVCC
FAULT
EN
SYNC
MODE
PGND
DIM
V
IN
COMP
High Current WLEDs
Enable
Dimming
Fault
LED7707
Sync Output
Faults Management Selection
AM00591v1
LED7707 Operation description
21/47
5.2.2 PWM dimming
The brightness control of the LEDs is performed by a pulse-width modulation of the rows
current. When a PWM signal is applied to the DIM pin, the current generators are turned on
and off mirroring the DIM pin behavior. Actually, the minimum dimming duty-cycle depends
on the dimming frequency.
The real limit to the PWM dimming is the minimum on-time that can be managed for the
current generators; this minimum on-time is approximately 10 μs.
Thus, the minimum dimming duty-cycle depends on the dimming frequency according to the
following formula:
Equation 10
For example, at a dimming frequency of 1 kHz, 1% of dimming duty-cycle can be managed.
During the off-phase of the PWM signal the boost converter is paused and the current
generators are turned off. The output voltage can be considered almost constant because of
the relatively slow discharge of the output capacitor. During the start-up sequence (see
Section 5.1.3 on page 13) the dimming duty-cycle is forced to 100% to detect floating rows
regardless of the applied dimming signal.
Figure 15. PWM dimming waveforms
5.3 Fault management
The main loop keeps the row having the lowest voltage drop regulated to about 700 mV.
This value slightly depends on the voltage across the remaining active rows. After the soft-
start sequence, all protections turn active and the voltage across the active current
generators is monitored to detect shorted LEDs.
DIMmin,DIM fs10D
⋅
μ
=
10µs minimum on-time10µs minimum on-time
Operation description LED7707
22/47
5.3.1 FAULT pin
The FAULT pin is an open-collector output, (with 4 mA current capability) active low, which
gives information regarding faulty conditions eventually detected. This pin can be used
either to drive a status LED or to warn the host system.
The FAULT pin status is strictly related to the MODE pin setting (see Ta bl e 6 for details).
5.3.2 MODE pin
The MODE pin is a digital input and can be connected to AVCC or SGND in order to choose
the desired fault detection and management. The LED7707 can manage a faulty condition
in two different ways, according to the application needs. Ta bl e 6 summarizes how the
device detects and handles the internal protections related to the boost section (over-
current, over-temperature and over-voltage) and to the current generators section (open and
shorted LEDs).
5.3.3 Open LED fault
In case a row is not connected or a LED fails open, the device has two different behaviors
according to the MODE pin status. If the MODE pin is high (i.e. connected to AVCC), the
FAULT pin is set high as soon as the device recognizes the event; the open row is excluded
from the control loop and the device continues to work properly with the remaining rows.
Thus, if less than six rows are used in the application, the MODE pin must be set high.
Connecting the MODE pin to SGND, the LED7707 behaves in a different manner: as soon
as an open row is detected the FAULT pin is tied low and the device is turned off. The
internal logic latches this status: to restore the normal operation, the device must be
restarted by toggling the EN pin or performing a power-on reset (POR occurs when the
voltage at the LDO5 pin falls below the lower UVLO threshold and subsequently rises above
the upper one).
Table 6. Faults management summary
FAULT MODE to GND MODE to VCC
Internal MOSFET
over-current
FAULT pin HIGH
Power MOS turned OFF
Output over-voltage FAULT pin LOW
Device turned OFF, latched condition
Thermal shutdown FAULT pin LOW. device turned OFF.
Automatic restart after 30 C temperature drop.
LED short circuit
FAULT pin LOW, device turned
OFF (100s masking time),
latched condition (Vth = 4.0 V)
-
Open row(s)
FAULT pin LOW
Device turned OFF at first
occurrence, latched condition
FAULT pin HIGH faulty row(s)
disconnected.
LED7707 Operation description
23/47
Figure 16 shows an example of open channel detection in case of MODE connected to
AVCC.
At the point marked as “1” in Figure 16, the row opens (row current drops to zero). From this
point on the output voltage is increased as long as the output voltage reaches the floating
row detection threshold (see Section 5.1.3 on page 13). Then (point marked as “2”) the
faulty row is disconnected and the device keeps on working only with the remaining rows.
Figure 16. Open channel detection (MODE to AVCC)
5.3.4 Shorted LED fault
When a LED is shorted, the voltage across the related current generator increases of an
amount equal to the missing voltage drop of the faulty LED. Since the feedback voltage on
each active generator is constantly compared with a fault threshold VTH,FAULT
, the device
detects the faulty condition and acts according to the MODE pin status.
A 100 µs masking time is introduced to support ESD capacitors eventually connected
across the LEDs strings.
If the MODE pin is low, the fault threshold is VTH,FAULT = 4.0 V. When the voltage across a
row is higher than this threshold for more than 100 μs, the FAULT pin is set low and the
device is turned off. The internal logic latches this status until the EN pin is toggled or a POR
is performed.
In case the MODE pin is connected to AVCC, the LED short-circuit protection is disabled.
The LED7707 simply keeps on regulating the set current without affecting the FAULT pin.
Despite the higher power dissipation, this option is useful to avoid undesired triggering of
the shorted-LED protection simply due to the high voltage drop spread across the LEDs.
Figure 17 shows an example of shorted LED detection in case MODE is connected to GND.
1
2
11
22
Operation description LED7707
24/47
At the point marked as “1” in Figure 17 one LED fails becoming a short-circuit. The voltage
across the current generator of the channel where the failed LED is connected increases by
an amount equal to the forward voltage of the faulty LED. Since the voltage across the
current generator is above the threshold (4 V), the device is turned off and the fault pin is set
low (point “2”). Note that, once a new dimming cycle starts (point “3”), the device waits the
masking time (approximately 100 μs) and then sets the FAULT pin low and turns off.
Figure 17. Shorted LED detection (MODE to GND)
1
2
masking time
3
11
22
masking time
33
LED7707 Application information
25/47
6 Application information
6.1 System stability
The boost section of the LED7707 is a fixed frequency, current-mode converter. During
normal operation, a minimum voltage selection circuit compares all the voltage drops across
the active current generators and provides the minimum one to the error amplifier. The
output voltage of the error amplifier determines the inductor peak current in order to keep its
inverting input equal to the reference voltage (700 mV typ). The compensation network
consists of a simple RC series (RCOMP - CCOMP) between the COMP pin and ground.
The calculation of RCOMP and CCOMP is fundamental to achieve optimal loop stability and
dynamic performance of the boost converter and is strictly related to the operating
conditions.
6.1.1 Loop compensation
The compensation network can be quickly calculated using equations 11 to 16. Once both
RCOMP and CCOMP have been determined, a fine-tuning phase may be required in order to
get the optimal dynamic performance from the application.
The first parameter to be fixed is the switching frequency. Normally, a high switching
frequency allows reducing the size of the inductor and positively affects the dynamic
response of the converter (wider bandwidth) but increases the switching losses. For most of
applications, the fixed value (660 kHz) represents a good trade-off between power
dissipation and dynamic response, allowing to save an external resistor at the same time. In
low-profile applications, the inductor value is often kept low to reduce the number of turns;
an inductor value in the 4.7 µH-15 µH range is a good starting choice.
In order to avoid instability due to interaction between the DC-DC converter's loop and the
current generators' loop, the bandwidth of the boost should not exceed the bandwidth of the
current generators. A unity-gain frequency (fU) in the order of 30-40 kHz is acceptable. Also,
take care not to exceed the CCM-mode right half-plane zero (RHPZ).
Equation 11
Equation 12
Equation 13 a
SWU F2.0f
⋅
≤
L2
I
V
V
V
2.0
L2
RM
2.0f OUT
OUT
2
OUT
min,IN
2
U⋅π
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
⋅=
⋅π
⋅≤
OUT
min,IN
V
V
M=
Application information LED7707
26/47
Equation 13b
Where VIN,min is the minimum input voltage and IOUT is the overall output current.
Note that, the lower the inductor value (and the higher the switching frequency), the higher
the bandwidth can be achieved. The output capacitor is directly involved in the loop of the
boost converter and must be large enough to avoid excessive output voltage drop in case of
a sudden line transition from the maximum to the minimum input voltages.
However a more significant requirement concerns the output voltage ripple.
The output capacitor should be chosen in accordance with the following expression:
Equation 14
where ΔVOUT, max is the maximum acceptable output voltage ripple, IL, peak is the peak
inductor current, TOFF is the off-time of the switching cycle (for an extensive explanation see
Section 6.4.4 on page 33).
Once the output capacitor has been chosen, the RCOMP can be calculated as:
Equation 15
Where GM = 2.7 S and gEA = 375 µS
Equation 15 places the loop bandwidth at fU. Then, the CCOMP capacitor is determined to
place the frequency of the compensation zero 5 times lower than the loop bandwidth:
Equation 16
Where fZ = fU/5.
In most of the applications an experimental approach is also very valid to compensate the
circuit. A simple technique to optimize different applications is to choose CCOMP = 4.7 nF
and to replace RCOMP with a 10 kΩ trimmer adjusting its value to properly damp the output
transient response. Insufficient damping will result in excessive ringing at the output and
poor phase margin.
Figure 18 (a and b) give an example of compensation adjustment for a typical application.
OUT
OUT
I
V
R=
(
)
max,OUT
OFFOUTpeak,L
OUT V2
TII
CΔ⋅
⋅−
>
MgG
Cf2
R
EAM
U
COMP ⋅⋅
⋅
⋅
π
=
COMPZ
COMP Rf2
1
C⋅⋅π
=
LED7707 Application information
27/47
Figure 18. Poor phase margin (a) and properly damped (b) load transient responses
Figure 19. Load transient response measurement set-up
a) b)
ROW1
ROW2
ROW3
ROW4
ROW5
ROW6
VCC
BILIM
RILIM
SS
SGND
SLOPE
VIN
LX
OVSEL
FSW
AVCC
FAULT
EN
SYNC
MODE
PGND
DIM
V
IN
= 12V V
BOOST
IN
C
COMP
+5V
Up to 10 WLEDs per row
LED7707
6.8μH
2 x 4.7μF
MLCC
100mA
500Hz
R
L
=V
BST
AM00592v1
Application information LED7707
28/47
6.2 Thermal considerations
In order to prevent the device from exceeding the thermal shutdown threshold (150 °C), it is
important to estimate the junction temperature through the following equation:
Equation 17
where TA is the ambient temperature, Rth,JA is the equivalent thermal resistance junction to
ambient and PD,tot is the power dissipated by the device.
The Rth,JA measured on the application demonstration board (described in Section 6.5) is
42 °C/W.
The PD,tot has several contributions, listed below.
a) Conduction losses due to the RDS(on) of the internal power switch, equal to:
Equation 18
where D is defined as:
Equation 19
and DDIM is the duty cycle of the PWM dimming signal.
b) Switching losses due to the power MOSFET turn on and off, calculated as:
Equation 20
where tr and tf are the power MOSFET rise time and fall time respectively.
c) Current generators losses. This contribution is strictly related to the LEDs used in
the application. Only the contribution of the leading current generator (“master”
current generator) can be predicted, regardless of the LEDs forward voltage:
Equation 21
where IROW is the current flowing through the row, whereas VIFB is the voltage across the
master current generator (typically 700 mV).
The voltages across the other current generators depend on the spread of the LEDs forward
voltage. The worst case for power dissipation (maximum forward voltage LEDs in the master
row, minimum forward voltage LEDs in all other rows) can be estimated as:
tot,DJA,thAJ PRTT
⋅
+
=
DIM
2
INDSoncond,D DDIRP ⋅⋅⋅=
OUT
IN
V
V
1D −=
DIM
fr
swINOUTsw,D D
2
)tt(
fIVP ⋅
+
⋅⋅⋅=
DIMIFBROWMaster,GEN DVIP
⋅
⋅
=
LED7707 Application information
29/47
Equation 22
where nROWs is the number of active rows, ΔVf,LEDs is the spread of the LEDs forward
voltage and nLEDs is the number of LEDs per row.
d) LDO losses, due to the dissipation of the 5 V linear regulator:
Equation 23
The LED7707 is housed in a 24 leads 4x4-VFQFPN package with exposed pad that allows
good thermal performance. However it is also important to design properly the
demonstration board layout in order to assure correct heat dissipation.
Figure 20 shows a picture of the LED7707 application demonstration board taken using an
infrared camera. The chip temperature, in those application conditions, is kept below 50 °C.
Figure 20. Demonstration board thermographic analysis
(
)
(
)
DIMLEDsLEDs,fIFBROWsROWGEN DnVV1nIP
⋅
⋅
Δ
+
⋅
−
⋅
=
(
)
LDOLDOINLDO,D IVVP
⋅
−
=
64°C 50°C
VIN = 12V
IROW = 60mA
VOUT = 30V
FSW = 660kHz
DDIM = 100%
Tamb = 25°C 64°C 50°C
VIN = 12V
IROW = 60mA
VOUT = 30V
FSW = 660kHz
DDIM = 100%
Tamb = 25°C
Application information LED7707
30/47
6.3 Component selection
6.3.1 Inductor selection
Being the LED7707 mostly dedicated to backlighting, real-estate applications dictate severe
constrain in selecting the optimal inductor. The inductor choice must take into account
different parameters like conduction losses (DCR), core losses (ferrite or iron-powder),
saturation current and magnetic-flux shielding (core shape and technology).
The switching frequency of the LED7707 can be set in the 200 kHz-1 MHz range, allowing a
wide selecting room for the inductance value. Low switching frequencies takes to high
inductance value, resulting in significant DCR and size. On the other hand, high switching
frequencies result in significant core losses. The suggested range is 4.7-22 µH, even if the
best trade off between the different loss contributions varies from manufacturer to
manufacturer.
A 6.8 µH inductor has been experimentally found as the most suitable for applications
running at a 660 kHz switching frequency.
6.3.2 Capacitors selection
The input and output capacitors should have very low ESR (ceramic capacitors) in order to
minimize the ripple voltage. The boost converter of the LED7707 has been designed to
support ceramic capacitors. The required capacitance depends on the programmed LED
current and the minimum dimming frequency (the boost converter is off when the DIM pin is
low and the output capacitor is slowly discharged). Considering the worst case (i.e. 200 Hz
dimming frequency and 85 mA/channel), two 4.4 µF MLCCs are suitable for almost all
applications. Particular care must be taken when selecting the rated voltage and the
dielectric type of the output capacitors: 50 V rated MLCC may show a significant
capacitance drop when biased, especially in case of Y5V dielectric.
As in most of boost converters, the input capacitor is less critical, although it is necessary to
reduce the switching noise on the supply rail. The input capacitor is also important for the
internal LDO of the LED7707 and must be kept as close as possible to the chip. The rated
voltage of the input capacitor can be chosen according to the supply voltage range; a 10 µF
X5R MLCC is recommended.
Table 7. Recommended inductors
Manufacturer Part number Description Size
Coilcraft LPS6235-682MLC 6.8 μH, 75 mΩ, 2.7 A 6x6 mm
Coilcraft XPL7030-682ML 6.8 μH, 60 mΩ, 5.8 A 7x7 mm
Wurth 7440650068 6.8 μH, 33 mΩ, 3.6 A 10x10 mm
Table 8. Recommended capacitors
Manufacturer Part number Description Package Notes
Taiyo Yuden UMK325BJ106KM-T Ceramic, 35V, X5R, 20 % SMD 1210 CIN
Murata GRM31CR71H225KA88B Ceramic, 50V, X7R, 20 % SMD 1206 COUT
LED7707 Application information
31/47
6.3.3 Flywheel diode selection
The flywheel diode must be a Schottky type to minimize the losses. This component is
subject to an average current equal to the output one and must sustain a reverse voltage
equal to the maximum output rail voltage. Considering all the channels sinking 75 mA each
(i.e. 450 mA output current) and the maximum output voltage (36 V), the STP1L40M
(If,ave = 1 A, Vr = 40 V) diode is a good choice. Smaller diodes can be used in applications
involving lower output voltage and/or lower output current.
6.4 Design example
In order to help the design of an application using the LED7707, in this section a simple
step-by-step design example is provided.
A possible application could be the LED backlight in a 17” LCD panel using the LED7707.
Here below the possible application conditions are listed:
●VIN = 12 ± 10 %
●4 strings of 42 white LEDs (60 mA) each (arranged in 6 rows, 7LEDs per row)
●VF, L E D s = 3.5 V ± 200 mV
6.4.1 Switching frequency setting
To reduce the number of the external components, the default switching frequency is
selected (660 kHz typ.) by connecting the FSW pin to AVCC pin.
However, in case a different switching frequency is required, a resistor from FSW pin and
ground can be connected, according to the equation (5) in section 4.1.5.
6.4.2 Row current setting
Considering the equation 9 in Section 5.2.1, the RRILIM resistor can be calculated as:
Equation 24
The closest standard commercial value is 30 kΩ. The actual value of the row current will be
a little lower (61.7 mA).
6.4.3 Inductor choice
The boost section, as all DC-DC converters, can work in CCM (continuous conduction
mode) or in DCM (discontinuous conduction mode) depending on load current, input and
output voltage and other parameters, among which the inductor value.
In a boost converter it is usually preferable to work in DCM.
Once the load, the input and output voltage, and the switching frequency are fixed, the
inductor value defining the boundary between DCM and CCM operation can be calculated
as:
Ω=== k83.30
mA60
V1850
I
K
R
ROW
R
RILIM
Application information LED7707
32/47
Equation 25
where D is the duty-cycle defined as:
Equation 26
whereas R0 is:
Equation 27
and
Equation 28
The output voltage in the above calculations is considered as the maximum value (LED with
the maximum forward voltage connected to the leading generator):
Equation 29
Considering the input voltage range, the lower LB will be at the lower input voltage. Hence
the condition to assure the DCM operation becomes:
Equation 30
An inductor value of 4.7 µH could be a suitable value, considering also a margin from the
boundary condition.
It is important to highlight that the inductor choice involves not only the value itself but the
saturation current (higher than the current limit, see Section 6.4.4), the rated RMS current
(the compliance with the saturation current might be not enough; also the thermal
performances must be taken into account), the DCR (which affects the efficiency) and the
size (in some application might be a strict requirement).
However the DCR can’t be reduced keeping the size small. Hence a trade off between these
two requirements must be achieved according to the application.
(
)
SW
2
0
BF2
D1DR
L⋅
−⋅⋅
=
⎩
⎨
⎧
=
=
=−= V2.13V@50.0
V8.10V@59.0
V
V
1D
max,IN
min,IN
OUT
IN
Ω== 74
I
V
R
OUT
OUT
0
mA360I6I ROWOUT
=
⋅
=
V6.26mV700V7V max,LEDs,Fmax,OUT
=
+
⋅
=
(
)
H6.5VLL min,INB
μ
=
<
LED7707 Application information
33/47
6.4.4 Output capacitor choice
The choice of the output capacitor is mainly affected by the desired output voltage ripple.
Since the voltage across the LEDs can be considered almost constant, this ripple is
transferred across the current generators, affecting their dynamic response.
The output ripple can be estimated as (neglecting the contribution of ESR of COUT
, very low
in case of MLCC):
Equation 31
where IL, peak is the inductor peak current (see Figure 21) calculated as:
Equation 32
whereas D, working in DCM, is:
Equation 33
defining M as:
Equation 34
Figure 21. Inductor current in DCM operation
(
)
`
C2
TII
V
OUT
OFFOUTpeak,L
OUT ⋅
⋅−
=Δ
⎩
⎨
⎧
=
=
=
⋅
⋅
=V2.13V@A762.1
V8.10V@A915.1
LF
DV
I
max,IN
min,IN
sw
IN
peak,L
⎩
⎨
⎧
=
=
=
−⋅⋅⋅
=V2.13V@414.0
V8.10V@55.0
R
)1M(MLF2
D
max,IN
min,IN
0
sw
⎩
⎨
⎧
=
=
=V2.13V@015.2
V8.10V@463.2
V
V
M
max,IN
min,IN
IN
OUT
I
L, peak
I
L
t
T
SW
= 1/F
SW
T
ON
T
OFF
AM00593v1
Application information LED7707
34/47
TOFF can be calculated as:
Equation 35
defining D2 as:
Equation 36
The worst case for the output voltage ripple is when input voltage is lower (VIN,min = 10.8 V).
A simple way to select the COUT value is fixing a maximum voltage ripple.
In order to affect as less as possible the current generators, it would be better to fix the
maximum ripple lower than the typical voltage across the generators.
For example considering ΔVOUT lower than 70 mV (i.e. the 10 % of the voltage across the
leading generator), the required capacitance is:
Equation 37
A margin from the calculated value should be taken into account because of the
capacitance drop due to the applied voltage when MLCCs are used.
One 10 µF MLCC (or two 4.7 µF MLCCs) can be a good choice for this application.
In case a dimming duty cycle different from 100% is used, a further contribution to the
capacitor discharge (during the off time of the dimming cycle) should be considered.
6.4.5 Input capacitor choice
The input capacitor of a boost converter is less critical than the output capacitor, due to the
fact that the inductor is in series with the input, and hence, the input current waveform is
continuous.
A low ESR capacitor is always recommended.
A capacitor of 10 µF is tentatively a good choice for most of the applications.
⎩
⎨
⎧
=
=
=⋅= V2.13V@ns2.618
V8.10V@ns7.569
DTT
max,IN
min,IN
2SWOFF
()
⎩
⎨
⎧
=
=
=
−⋅
⋅⋅⋅
=V2.13V@408.0
V8.10V@376.0
1MR
MLF2
D
max,IN
min,IN
0
SW
2
(
)
F33.6
V2
TII
C
max,OUT
OFFOUTpeak,L
OUT μ=
Δ⋅
⋅−
>
LED7707 Application information
35/47
6.4.6 Over-voltage protection divider setting
The over-voltage protection (OVP) divider provides a partition of the output voltage to the
OVSEL pin. The OVP divider setting not only fixes the OVP threshold, but also the open-
channel detection threshold.
The proper OVP divider setting can be calculated by the equation (3):
Equation 38
where VOUT, MAX is the maximum output voltage considering the worst case (all LEDs with
the maximum VF = VF, m a x = 3.7 V on the same row):
Equation 39
R1 can be chosen is in the order of hundreds of kilo-ohms to reduce the leakage current in
the resistor divider. For example, setting R1 = 510 kΩ leads to R2 = 21.89 kΩ. The closest
standard commercial value is R2 = 22 kΩ.
6.4.7 Compensation network
For the compensation network, the suggestions provided in Section 6.1 are always valid.
In this condition, tentatively the following value of R3 and C8 (see Figure 24) are usually a
good choice for the loop stability:
R3 = 2.4 kΩ
C8 = 4.7 nF
6.4.8 Boost current limit
The boost current limit is set to protect the internal power switch against excessive current.
The slope compensation may reduce the programmed current limit. Hence, to take into
account this effect, as a rule of thumb, the current limit can be set as twice as much the
maximum inductor peak current (see Section 6.4.4):
IBOOST, PEAK > 3.83 A
Therefore, using equation (7) and choosing IBOOST, PEAK = 4 A, RBILIM will be:
Equation 40
V145.1V4V
V145.1
RR
MAX,OUT
12 −+
⋅=
V6.26mV700VnLEDV max,FOVP,OUT
=
+
⋅
=
Ω== k300
I
K
R
PEAK,BOOST
B
BILIM
Application information LED7707
36/47
6.4.9 Power dissipation estimate
As explained in section 5.2, there are several contributions to the total power dissipation.
Neglecting the power dissipated by the LDO (surely less significant compared with the other
contributions), equation (18), (20), (21) and (22) help to estimate the overall power
dissipation.
Before starting the power dissipation estimate it is important to highlight that the following
calculations are considering the worst case (the actual value of the dissipated power would
require measurements). Therefore the power dissipation is estimated according to the
following assumptions:
1. Minimum input voltage (10.8 V), which leads to maximum input current (and also D will
have the higher value, see Section 6.4.4);
2. Maximum RDS(on) of the internal power MOSFET;
3. LEDs in the row of the leading generator will have the maximum forward voltage,
whereas all other LEDs in the other rows will have the minimum forward voltage.
4. 100 % dimming signal duty cycle is considered.
The conduction and switching losses on the internal power switch can be calculated as:
Equation 41
Equation 42
where tr = tf = 15 ns
The power dissipation related to the current generators is given by:
Equation 43
Equation 44
Equation 45
The junction temperature can be estimated by equation (18) considering TA = 25 °C:
Equation 46
mW216DDIRP DIM
2
INDSoncond,D =⋅⋅⋅=
mW233D
2
)tt(
fIVP DIM
fr
swINOUTsw,D =⋅
+
⋅⋅⋅=
mW42DVIP DIMIFBROWMaster,GEN
=
⋅
⋅
=
()
(
)
mW630DnVV1nIP DIMLEDsLEDs,fIFBROWsROWGEN =
⋅
⋅
Δ
+
⋅
−
⋅=
W12.1PPPPP GENMaster,GENsw,Dcond,Dtot,D
=
+
+
+
≅
C72PRTT tot,DJA,thAmbJ °
=
⋅
+
=
LED7707 Application information
37/47
In order to estimate also the efficiency, other contributions to the power dissipation must be
added to PD, tot (which represents only the power dissipated by the device), that is:
Equation 47
where VF, Diode = 0.4 V
Equation 48
where DCR = 80 mΩ (typical DCR of the recommended inductors).
Therefore the total dissipated power is:
Equation 49
Considering the input power as the result of input voltage multiplied by the input current, the
estimated efficiency is:
Equation 50
Note: It is important to remind that the previous calculations consider the worst case, especially for
the power dissipated on the current generators.
Statistical analysis (confirmed by bench measurements) shows that the series connection of
more LEDs on each channel leads to compensation effects.
The hypothesis 3 above mentioned is thus rather unlikely.
Therefore PGEN is significantly lower and the overall efficiency is typically around 90 %.
mW133DIVP 2INDiode,FDiode,DISS
=
⋅
⋅
=
mW63IDCRIDCRP 2
IN
2
RMS,IndInd,DISS =⋅≅⋅=
W316.1PPPP Ind,DISSDiode,DISStot,DTOTAL,DISS
=
+
+
=
862.0
P
PP
IN
TOT,DISSIN =
−
=η
Application information LED7707
38/47
6.5 Layout consideration
1. A careful PCB layout is important for proper operation. In this section some guidelines
are provided in order to achieve a good layout.
2. The device has two different ground pins: signal ground (SGND) and power ground
(PGND). The PGND pin handles the switching current related to the boost section; for
this reason the PCB traces should be kept as short as possible and with adequate
width.
3. The signal ground is the return for the device supply and the current generators and
can be connected to the thermal pad.
4. The heat dissipation area (adequate to the application conditions) should be placed
backside respect to the device and with the lowest thermal impedance possible (i.e.
PCB traces in the backside should be avoided). The dissipation area is thermally and
electrically connected to the thermal pad by several vias (nine vias are recommended).
5. The signal and power grounds must be connected together in a single point as close as
possible to the PGND pin to reduce ground loops.
6. The R-C components of the compensation network should be placed as close as
possible to the COMP pin in order to avoid noise issue and instability of the
compensation.
7. Noise sensitive signals (i.e. feedbacks and compensation) should be routed as short as
possible to minimize noise collection. The LED7707 pinout makes it easy to separate
power components (e.g. inductor, diode) from signal ones.
8. The LX switching node should have and adequate width for high efficiency.
9. The critical power path inductor-LX-PGND must be as short as possible by mounting
the inductor, the diode and COUT as close as possible each other.
10. The capacitors of the compensated divider connected to the OVSEL pin should be
placed as close as possible to the OVSEL pin.
11. In order to assure good performance in terms of row current accuracy/mismatch, the
PCB traces from the rows pins to the LEDs should have similar length and width.
12. The capacitors of the filter connected to LDO5 and VIN pins should be mounted as
close as possible to the mentioned pins
Figure 22 and Figure 23 shows the demonstration board layout (top view and bottom view
respectively).
Application information LED7707
40/47
Figure 24. LED7707 demonstration board schematic
Table 9. LED7707 demonstration board component list
Component Description Package Part number MFR Value
C1 Ceramic, 35 V
X5R, 20 % SMD 1210 UMK325BJ106KM-T Taiyo Yuden 10 µF
C2,C3 Ceramic, 50 V
X7R, 20 %
SMD 1206 GRM31CR71H475KA88B Murata 4.7 µF
C4 SMD 1206 GRM31CR71H225KA88B N.M.
C5
Ceramic, 25 V
X5R, 20 % SMD 0603 Standard
1 µF
C6 100 nF
C7 3.3 nF
C8 4.7 nF
C9 N.M.
C10 220 pF
C11 4.7 nF
C12 N.M.
C13 15 pF
R1
Chip resistor
0.1 W, 1 % SMD 0603 Standard
510 kΩ
R2 16 kΩ
R3 2.4 kΩ
R4 4.7 Ω
R5 330 kΩ
R6 24 kΩ
R7 Chip resistor
0.1 W, 1 % SMD 0603 Standard 360 kΩ
LED7707 Application information
41/47
Component Description Package Part number MFR Value
R8 680 kΩ
R9, R10 100 kΩ
R11 1.2 kΩ
R12 N.M.
R13 N.M.
L1 6u8, 60 mΩ, 5.8 A 7x7 mm XPL7030-682ML Coilcraft 6.8 µF
D1 Schottky, 40 V, 1 A DO216-AA STPS1L40M ST STPS1L40M
D2 Red LED, 3 mA SMD 0603 Standard
D3 Signal Schottky SOD-523 BAS69 N.M.
U1 Integrated circuit QFN4x4 LED7707 ST LED7707
J2 PCB pad jumper
J8 Header 8 SIL 8 Standard
SW1, SW2 Jumper 3 SIL 3 Standard
SW3 Push button 6x6 mm FSM4JSMAT TYCO
Table 9. LED7707 demonstration board component list (continued)
Electrical characteristics LED7707
42/47
7 Electrical characteristics
Figure 25. Efficiency versus DIM duty cycle,
VIN = 12 V, 6 rows, 10 white LEDs
(60 mA) in series, FSW = 660 kHz
Figure 26. Efficiency versus DIM duty cycle,
VIN = 18 V, 6 rows, 10 white LEDs
(60 mA) in series, FSW = 660 kHz
Figure 27. Efficiency versus DIM duty cycle,
VIN = 24 V, 6 rows, 10 white LEDs
(60 mA) in series, FSW = 825 kHz
Figure 28. Efficiency versus DIM duty cycle,
VIN = 24 V, 6 rows, 10 white LEDs
(60 mA) in series, FSW = 825 kHz
LED7707 Electrical characteristics
43/47
Figure 29. Soft-start waveforms (EN, SS, and
VOUT monitored)
Figure 30. Boost section switching signals
(LX, SYNC and inductor current
monitored), VIN = 12 V, 10 LEDs
Figure 31. Dimming waveforms
(FDIM = 200 Hz)
Figure 32. Dimming waveforms (FDIM = 1 kHz)
Package mechanical data LED7707
44/47
8 Package mechanical data
In order to meet environmental requirements, ST offers these devices in different grades of
ECOPACK® packages, depending on their level of environmental compliance. ECOPACK®
specifications, grade definitions and product status are available at: www.st.com.
ECOPACK is an ST trademark.
LED7707 Package mechanical data
45/47
Table 10. VFQFPN-24 4 mm x 4 mm mechanical data
Dim.
mm
Min Typ Max
A 0.80 0.90 1.00
A1 0.00 0.02 0.05
A3 0.20
b 0.18 0.25 0.30
D 3.85 4.00 4.15
D2 2.40 2.50 2.60
E 3.85 4.00 4.15
E2 2.40 2.50 2.60
e0.50
L 0.30 0.40 0.50
ddd 0.08
Figure 33. Package dimensions
LED7707
47/47
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