MIC4723YMLTR - Micrel, Inc.

MIC4723

3A 2MHz Integrated Switch

Buck Regulator

MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.

Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com

June 2008 M9999-060308-E

General Description

The Micrel MIC4723 is a high efficiency PWM buck (step-

down) regulator that provides up to 3A of output current.

The MIC4723 operates at 2.0MHz and has proprietary

internal compensation that allows a closed loop bandwidth

of over 200KHz.

The low on-resistance internal p-channel MOSFET of the

MIC4723 allows efficiencies over 92%, reduces external

components count and eliminates the need for an

expensive current sense resi stor.

The MIC4723 operates from 2.7V to 5.5V input and the

output can be adjusted down to 1V. The devices can

operate with a maximum duty cycle of 100% for use in low-

dropout conditions.

The MIC4723 is available in the exposed pad 12-pin

3mm x 3mm MLF® and 10-pin ePAD MSOP packages with

a junction operating range from –40°C to +125°C.

Data sheets and support documentation can be found on

Micrel’s web site at: www.micrel.com.

Features

• 2.7/3.0V to 5.5V supply voltage

• 2.0MHz PWM mode

• Output current to 3A

• Up to 94% efficiency

• 100% maximum duty cycle

• Adjustable output voltage option down to 1V

• Ultra-fast transient response

• Ultra-small external components

Stable with a 1µH inductor and a 4.7µF output

capacitor

• Fully integrated 3A MOSFET switch

• Micropower shutdown

• Thermal shutdown and current limit protection

• Pb-free 12-pin 3mm x 3mm MLF® package

• Pb-free 10-pin ePAD MSOP package

• –40°C to +125°C junction temperature range

Applications

• FPGA/DSP/ASIC applications

• General point of load

• Broadband communications

• DVD/TV recorders

• Point of sale

• Printers/Scanners

• Set top boxes

• Computing peripherals

• Video cards

___________________________________________________________________________________________________________

Typical Application

3A 2MHz Buck Regulator 76

0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

3.3VOUT Efficiency

4.5VIN

5.5VIN

5VIN

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Ordering Information

Part Number Voltage Temperature Range Package Lead Finish

MIC4723YML Adj. –40° to +125°C 12-Pin 3x3 MLF® Pb-Free

MIC4723YMME Adj. –40° to +125°C 10-Pin ePAD MSOP Pb-Free

Note

MLF® is a GREEN RoHS compliant package. Lead finish is NiPdAu. Mold compound is Halogen Free.

Pin Configur ation

BIAS EN

VIN

PGND

SGND

VIN

PGOOD

FB NC

FB EN65

1SW

VIN

SGND

BIAS

10 SW

VIN

PGND

PGOOD

4EP

12-Pin 3mm x 3mm MLF (ML) 10-Pin ePAD MSOP (MME)

Pin Description

Pin Number

MLF-12 Pin Number

MSOP-10 Pin Name Pin Function

1, 12 1, 10 SW Switch (Output): Internal power P-Channel MOSFET output switch.

2, 11 2, 9 VIN Supply Voltage (Input): Supply voltage for the source of the internal P-channel

MOSFET and driver. Requires b ypass capacitor to GND.

3 8 PGND Power Ground. Provides the ground return path for the high-side drive current.

4 3 SGND

Signal (Analog) Ground. Provides return path for control circuitry and internal

reference.

5 4 BIAS

Internal circuit bias supply. Must be bypassed with a 0.1µF ceramic cap acitor to

SGND.

6 5 FB

Feedback. Input to the error amplifier, connect to the external resistor divider network

to set the output voltage.

7, 10 — NC No Connect. Not internally connected to di e. This pin can be tied to any other pin if

desired.

8 6 EN

Enable (Input). Logic level low, will shutdown the device, reducing the current draw to

less than 5µA.

9 7 PGOOD

Power Good. Open drain output that is pulled to ground when the output voltage is

within ±7.5% of the set regulation voltage.

EP EP GND Connect to ground.

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Absolute Maximum Ratings(1)

Supply Voltage (VIN).......................................................+6V

Output Switch Voltage (VSW)..........................................+6V

Output Switch Current (ISW)............................................11A

Logic Input Voltage (VEN) ..................................–0.3V to VIN

Storage Temperature (Ts).........................–60°C to +150°C

Operating Ratings(2)

Supply Voltage (VIN).....................................+2.7V to +5.5V

Logic Input Voltage (VEN).......................................0V to VIN

Junction Temperature (TJ)........................–40°C to +125°C

Junction Thermal Resistance

3mm x 3mm MLF-12 (θJA).................................60°C/W

3mm x 3mm MLF-12 (θJc) .................................28°C/W

ePAD MSOP-10 (θJA)........................................76°C/W

ePAD MSOP-10 (θJc) ........................................28°C/W

Electrical Characteristics(4)

VIN = VEN = 3.6V; L = 1µH; COUT = 4.7µF; TA = 25°C, unless noted. Bold values indicate –40°C< TJ < +125°C.

Parameter Condition Min Typ Max Units

Supply Voltage Range MIC4723YML

MIC4723YMME 2.7

3.0 5.5

5.5 V

Under-Voltage Lockout

Threshold (turn-on) 2.45 2.55 2.65 V

UVLO Hysteresis 100 mV

Quiescent Current VFB = 0.9 * VNOM (not switching) 570 900 µA

Shutdown Current VEN = 0V 2 10 µA

[Adjustable] Feedback

Voltage ± 2% (over temperature) ILOAD = 100mA 0.98 1.02 V

FB pin input current 1 nA

Current Limit in PWM Mode VFB = 0.9 * VNOM 3.5 5 A

Output Voltage Line

Regulation VOUT > 2V; VIN = VOUT+500mV to 5.5V; ILOAD= 100mA

VOUT < 2V; VIN = 2.7V to 5.5V; ILOAD= 100mA 0.07 %

Output Voltage Load

Regulation 20mA < ILOAD < 3A 0.2 %

Maximum Duty Cycle VFB ≤ 0.4V 100 %

PWM Switch ON-

Resistance ISW = 50mA; VFB = GND (High Side Switch) 95 200

300 mΩ

mΩ

Oscillator Frequency 1.8 2 2.2 MHz

Enable Threshold 0.5 0.85 1.3 V

Enable Hysteresis 50 mV

Enable Input Current 0.1 2.3 µA

Power Good Range ±7 ±10 %

Power Good Resistance IPGOOD = 500µA 150 250 Ω

Over-Temperature

Shutdown 160

°C

Over-Temperature

Hysteresis 25 °C

Notes:

1. Exceeding the absolute maximum rating ma y damage the device.

2. The device is not guaranteed to function outside its operating rating.

3. Specification for packaged product only.

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Typical Characteristics

0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

3.3VOUT Effici ency

4.5VIN

5.5VIN

5VIN

0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

2.5VOUT Effici ency

5VIN

5.5VIN

4.5VIN

100

0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

2.5VOUT Efficiency

3.3VIN

3.6VIN

3VIN

0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

1.8VOUT Effici ency

4.5VIN

5.5VIN

5VIN

0 0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

1.8VOUT Effici ency

3.3VIN

3.6VIN

3VIN

0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

1.5VOUT Efficiency

4.5VIN

5.5VIN

5VIN

0 0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

1.5VOUT Effici ency

3.3VIN

3.6VIN

3VIN

0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

1.2VOUT Effici ency

4.5VIN

5.5VIN

5VIN

0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

1.2VOUT Efficiency

3.3VIN

3.6VIN

3VIN

0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

1VOUT Effi ciency

4.5VIN

5.5VIN

5VIN

0 0.5 1 1.5 2 2.5 3

OUTPUT CURRENT (A)

MIC4723

1VOUT Effi ciency

3VIN

3.6VIN

3.3VIN

Load Regulatio n

3.3V

0.990

0.995

1.000

1.005

1.010

0 0.5 1 1.5 2 2.5 3

OUTPUT VOLTAGE (V)

OUTPUT CURRENT (A)

VIN = 3.3V

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Typical Characteristics (continue)

0.9990

0.9992

0.9994

0.9996

0.9998

1.0000

1.0002

1.0004

1.0006

1.0008

1.0010

2.7 3.2 3.7 4.2 4.7 5.2

SUPPLY VOLTAGE (V)

Line Regulation

0.990

0.992

0.996

0.998

1.000

1.002

1.004

1.006

1.008

1.010

20 40 60 80

TEMPERATURE (°C)

Feedback Voltage

vs. Temperature

0.994

VIN = 3.3V

1.5

1.6

1.7

1.8

2.0

2.1

2.2

2.3

2.4

2.5

20 40 60 80

TEMPERATURE (°C)

Frequency

vs. Temperature

VIN = 3.3V

1.9

0.2

0.4

0.6

0.8

1.0

1.2

012345

SUPPLY VOLTAGE (V)

Feedback Voltage

vs. Supply Volt age

VEN = VIN

100

200

300

400

500

600

700

800

0123456

SUPPLY VOLTAGE (V)

Quiescent Current

vs. Supply Volta ge

VEN = VIN

100

105

110

115

120

2.7 3.2 3.7 4.2 4.7 5.2

SUPPLY VOLTAGE (V)

RDSON

vs. Supply Voltage

100

120

140

160

20 40 60 80

TEMPERATURE (°C)

RDSON

vs. Temperature

VIN = 3.3V

0.2

0.4

0.6

0.8

1.0

1.2

2.7 SUPPLY VOLTAGE (V)

Enable Threshold

vs. Supply Voltage

3.73.2 4.2 4.7

0.2

0.4

0.6

0.8

1.2

Enable Threshold

vs. Temperature

20 40 60 80

TEMPERATURE (°C)

VIN = 3.3V

1.0

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Functional Characteristics

Micrel, Inc. MIC4723

June 2008 7 M9999-060308-E

Functional Diagram

VIN

BIAS

PGOOD

PGND

Enable and

Control Logic

PWM

Control

P-Channel

Current Limit

SGND

1.0V

Soft

Start

Bias,

UVLO,

Thermal

Shutdown

HSD

MIC4723 Block Diagram

Micrel, Inc. MIC4723

June 2008 8 M9999-060308-E

Pin Description

VIN

Two pins for VIN provide power to the source of the

internal P-channel MOSFET along with the current

limiting sensing. The VIN operating voltage range is from

2.7V to 5.5V for the MIC4723YML or 3.0V to 5.5V for the

MIC4723YMME. Due to the high switching speeds, a

10µF capacitor is recommended close to VIN and the

power ground (PGND) for each pin for bypassing.

Please refer to layout recommendations.

BIAS

The bias (BIAS) provides power to the internal reference

and control sections of the MIC4723. A 10Ω resistor

from VIN to BIAS and a 0.1µF from BIAS to SGND is

required for clean op eration.

The enable pin provides a logic level control of the

output. In the off state, supply current of the device is

greatly reduced (typically <1µA). Do not drive the enable

pin above the supply voltage.

The feedback pin (FB) provides the control path to

control the output. For adjustable versions, a resistor

divider connecting the feedback to the output is used to

adjust the desired output voltage. The output voltage is

calculated as follows:

⎟

⎠

⎞

⎜

⎝

⎛+×= 1

VV REFOUT

where VREF is equal to 1.0V.

A feedforward capacitor is recommended for most

designs using the adjustable output voltage option. To

reduce current draw, a 10K feedback resistor is

recommended from the output to the FB pin (R1). Also, a

feedforward capacitor should be connected between the

output and feedback (across R1). The large resistor

value and the parasitic capacitance of the FB pin can

cause a high frequency pole that can reduce the overall

system phase margin. By placing a feedforward

capacitor, these effects can be significantly reduced.

Feedforward capacitance (CFF) can be calculated as

follows:

200kHzR12 1

CFF ××

The switch (SW) pin connects directly to the inductor

and provides the switching current necessary to operate

in PWM mode. Due to the high speed switching on this

pin, the switch node should be routed away from

sensitive nodes. This pin also connects to the cathode of

the free-wheeling diode.

PGOOD

Power good is an open drain pull down that indicates

when the output voltage has reached regulation. When

power good is low, then the output voltage is within

±10% of the set regulation voltage. For output voltages

greater or less than 10%, the PGOOD pin is high. This

should be connected to the input supply through a pull

up resistor. A delay can be added by placing a capacitor

from PGOOD to ground.

PGND

Power ground (PGND) is the ground path for the

MOSFET drive current. The current loop for the power

ground should be as small as possible and separate

from the Signal ground (SGND) loop. Refer to the layout

considerations for more details.

SGND

Signal ground (SGND) is the ground path for the biasing

and control circuitry. The current loop for the signal

ground should be separate from the power ground

(PGND) loop. Refer to the layout considerations for more

details.

Micrel, Inc. MIC4723

June 2008 9 M9999-060308-E

Application Information

The MIC4723 is a 3A PWM non-synchronous buck

regulator. By switching an input voltage supply, and

filtering the switched voltage through an Inductor and

capacitor, a regulated DC voltage is obtained. Figure 1

shows a simplified example of a non-synchronous buck

converter.

Figure 1. Example of non-synchronous buck converter

For a non-synchronous buck converter, there are two

modes of operation; continuous and discontinuous.

Continuous or discontinuous refer to the inductor

current. If current is continuously flowing through the

inductor throughout the switching cycle, it is in

continuous operation. If the inductor current drops to

zero during the off time, it is in discontinuous operation.

Critically continuous is the point where any decrease in

output current will cause it to enter discontinuous

operation. The critically continuous load current can be

calculated as follows;

L22.0MHz

IIN

OUT

OUT ××

⎥

⎦

⎤

⎢

⎣

⎡−

Continuous or discontinuous operation determines how

we calculate peak inductor current.

Continuous Operation

Figure 2 illustrates the switch voltage and inductor

current during continuous operation.

Figure 2. Continuous Operation

The output voltage is regulated by pulse width

modulating (PWM) the switch voltage to the average

required output voltage. The switching can be broken up

into two cycles; On and Off.

During the on-time, Figure 3 illustrates the high side

switch is turned on, current flows from the input supply

through the inductor and to the output. The inductor

current is:

Figure 3. On-Time

charged at the rate;

(

)

LVV OUTIN

−

To determine the total on-time, or time at which the

inductor charges, the duty cycle needs to be calculated.

The duty cycle can be calculated a s;

OUT

and the On time is;

2.0MHz

TON =

Therefore, peak to peak ripple current is;

()

L2.0MHz V

IIN

OUT

OUTIN

pkpk ×

×−

−

Since the average peak to peak current is equal to the

load current. The actual peak (or highest current the

inductor will see in a steady-state condition) is equal to

the output current plus ½ the peak-to-peak current.

()

L2.0MHz2 V

II IN

OUT

OUTIN

OUTpk ××

×−

Figure 4 demonstrates the off-time. During the off-time,

the high-side internal P-channel MOSFET turns off.

Since the current in the inductor has to discharge, the

current flows through the free-wheeling Schottky diode

to the output. In this case, the inductor discharge rate is

(where VD is the diode forward voltage);

Micrel, Inc. MIC4723

June 2008 10 M9999-060308-E

()

LVV DOUT +

−

The total off time can be calculated as;

2.0MHz

TOFF −

Figure 4. Off-Time

Discontinuous Operation

Discontinuous operation is when the inductor current

discharges to zero during the off cycle. Figure 5

demonstrates the switch voltage and inductor currents

during discontinuou s operation.

Figure 5. Discontinuous Operation

When the inductor current (IL) has completely

discharged, the voltage on the switch node rings at the

frequency determined by the parasitic capacitance and

the inductor value. In Figure 5, it is drawn as a DC

voltage, but to see actual operation (with ringing) refer to

the functional characteristics.

Discontinuous mode of operation has the advantage

over full PWM in that at light loads, the MIC4723 will skip

pulses as nessasary, reducing gate drive losses,

drastically improving light load efficiency .

Efficiency Considerations

Calculating the efficiency is as simple as measuring

power out and dividing it by the power in;

100

Efficiency IN

OUT ×=

Where input power (PIN) is;

INININ IVP

and output power (POUT) is cal culated as;

OUTOUTOUT IVP

The Efficiency of the MIC4723 is determined by several

factors.

• Rdson (Internal P-channel Resistance)

• Diode conduction losses

• Inductor Conduction losses

• Switching losses

Rdson losses are caused by the current flowing through

the high side P-channel MOSFET. The amount of power

loss can be approximated by;

DIRP 2

OUTDSONSW ××=

Where D is the duty cycle.

Since the MIC4723 uses an internal P-channel

MOSFET, Rdson losses are inversely proportional to

supply voltage. Higher supply voltage yields a higher

gate to source voltage, reducing the Rdson, reducing the

MOSFET conduction losses. A graph showing typical

Rdson vs input supply voltage can be found in the typical

characteristics section of this datash eet.

Diode conduction losses occur due to the forward

voltage drop (VF) and the output current. Diode power

losses can be approximate d as follows;

()

D1IVP OUTFD −×

For this reason, the Schottky diode is the rectifier of

choice. Using the lowest forward voltage drop will help

reduce diode conduction losses, and improve efficiency.

Duty cycle, or the ratio of output voltage to input voltage,

determines whether the dominant factor in conduction

losses will be the internal MOSFET or the Schottky

diode. Higher duty cycles place the power losses on the

high side switch, and lower duty cycles place the power

Micrel, Inc. MIC4723

June 2008 11 M9999-060308-E

losses on the Schottky diode.

Inductor conduction losses (PL) can be calculated by

multiplying the DC resistance (DCR) times the square of

the output current;

OUTL IDCRP ×=

Also, be aware that there are additional core losses

associated with switching current in an inductor. Since

most inductor manufacturers do not give data on the

type of material used, approximating core losses

becomes very difficult, so verify inductor temperature

rise.

Switching losses occur twice each cycle, when the

switch turns on and when the switch turns off. This is

caused by a non-ideal world where switching transitions

are not instantaneous, and neither are currents. Figure 6

demonstrates how switching losses due to the

transitions dissipate power in the switch.

Figure 6. Switch ing Transition Losses

Normally, when the switch is on, the voltage across the

switch is low (virtually zero) and the current through the

switch is high. This equates to low power dissipation.

When the switch is off, voltage across the switch is high

and the current is zero, again with power dissipation

being low. During the transitions, the voltage across the

switch (VS-D) and the current through the switch (IS-D) are

at middle, causing the transition to be the highest

instantaneous power point. During continuous mode,

these losses are the highest. Also, with higher load

currents, these losses are higher. For discontinuous

operation, the transition losses only occur during the “off”

transition since the “on” transitions there is no current

flow through the inductor.

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June 2008 12 M9999-060308-E

Component Selection

Input Capacitor

A 10µF ceramic is recommended on each VIN pin for

bypassing. X5R or X7R dielectrics are recommended for

the input capacitor. Y5V dielectrics lose most of their

capacitance over temperature and are therefore, not

recommended. Also, tantalum and electrolytic capacitors

alone are not recommended due their reduced RMS

current handling, reliability, and ESR increases.

An additional 0.1µF is recommended close to the VIN

and PGND pins for high frequency filtering. Smaller case

size capacitors are recommended due to their lower

ESR and ESL. Please refer to layout recommendations

for proper layout of the input capacitor.

Output Capacitor

The MIC4723 is designed for a 4.7µF output capacitor.

X5R or X7R dielectrics are recommended for the output

capacitor. Y5V dielectrics lose most of their capacitance

over temperature and are therefore not recommended.

In addition to a 4.7µF, a small 0.1µF is recommended

close to the load for high frequency filtering. Smaller

case size capacitors are recommended due to there

lower equivalent series ES R and ESL.

The MIC4723 utilizes type III voltage mode internal

compensation and utilizes an internal zero to

compensate for the double pole roll off of the LC filter.

For this reason, larger output capacitors can create

instabilities. In cases where a 4.7µF output capacitor is

not sufficient, other values of capacitance can be used

but the original LC filter pole frequency determined by

CO = 4.7µF + L = 1µH (which is approximately 73.4KHz)

must remain fixed. Increasing COUT forces L to

decrease and vice versa.

Inductor Selection

The MIC4723 is designed for use with a 1µH inductor.

Proper selection should ensure the inductor can handle

the maximum average and peak currents required by the

load. Maximum current ratings of the inductor are

generally given in two methods; permissible DC current

and saturation current. Permissible DC current can be

rated either for a 40°C temperature rise or a 10% to 20%

loss in inductance. Ensure the inductor selected can

handle the maximum operating current. When saturation

current is specified, make sure that there is enough

margin that the peak current will not saturate the

inductor.

Diode Selection

Since the MIC4723 is non-synchronous, a free-wheeling

diode is required for proper operation. A Schottky diode

is recommended due to the low forward voltage drop

and their fast reverse recovery time. The diode should

be rated to be able to handle the average output current.

Also, the reverse voltage rating of the diode should

exceed the maximum input voltage. The lower the

forward voltage drop of the diode the better the

efficiency. Please refer to the layout recommendations to

minimize switching noise.

Feedback Resistors

The feedback resistor set the output voltage by dividing

down the output and sending it to the feedback pin. The

feedback voltage is 1.0V. Calculating the set output

voltage is as follows;

⎟

⎠

⎞

⎜

⎝

⎛+= 1

VV FBOUT

Where R1 is the resistor from VOUT to FB and R2 is the

resistor from FB to GND. The recommended feedback

resistor values for common output voltages are available

in the bill of materials on page 19. Although the range of

resistance for the FB resistors is very wide, R1 is

recommended to be 10K. This minimizes the effect the

parasitic capacitance of the FB node.

Feedforward Capaci tor (CFF)

A capacitor across the resistor from the output to the

feedback pin (R1) is recommended for most designs.

This capacitor can give a boost to phase margin and

increase the bandwidth for transient response. Also,

large values of feedforward capacitance can slow down

the turn-on characteristics, reducing inrush current. For

maximum phase boost, CFF can be calculated as follows;

R1200kHz2 1

CFF ××

Bias filter

A small 10Ω resistor is recommended from the input

supply to the bias pin along with a small 0.1µF ceramic

capacitor from bias to ground. This will bypass the high

frequency noise generated by the violent switching of

high currents from reaching the internal reference and

control circuitry. Tantalum and electrolytic capacitors are

not recommended for the bias, these types of capacitors

lose their ability to filter at high frequencies.

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Loop Stability and Bode Analysis

Bode analysis is an excellent way to measure small

signal stability and loop response in power supply

designs. Bode analysis monitors gain and phase of a

control loop. This is done by breaking the feedback loop

and injecting a signal into the feedback node and

comparing the injected signal to the output signal of the

control loop. This will require a network analyzer to

sweep the frequency and compare the injected signal to

the output signal. The most common method of injection

is the use of transformer. Figure 7 demonstrates how a

transformer is used to inject a signal into the feedback

network.

Figure 7. Transformer Injection

A 50Ω resistor allows impedance matching from the

network analyzer source. This method allows the DC

loop to maintain regulation and allow the network

analyzer to insert an AC signal on top of the DC voltage.

The network analyzer will then sweep the source while

monitoring A and R for an A/R measurement. While this

is the most common method for measuring the gain and

phase of a power supply, it does have significant

limitations. First, to measure low frequency gain and

phase, the transformer needs to be high in inductance.

This makes frequencies <100Hz require an extremely

large and expensive transformer. Conversely, it must be

able to inject high frequencies. Transformers with these

wide frequency ranges generally need to be custom

made and are extremely expensive (usually in the tune

of several hundred dollars!). By using an op-amp, cost

and frequency limitations used by an injection

transformer are completely eliminated. Figure 8

demonstrates using an op-amp in a summing amplifier

configuration for signal injection.

Network Analyzer

Source

+8V R1

1k R4

Feedback Output

Netwo

Analyzer

“A” Input

Network

Analyzer

“R” Input MIC922BC5

Figure 8. Op Amp Injection

R1 and R2 reduce the DC voltage from the output to the

non-inverting input by half. The network analyzer is

generally a 50Ω source. R1 and R2 also divide the AC

signal sourced by the network analyzer by half. These

two signals are “summed” together at half of their

original input. The output is then gained up by 2 by R3

and R4 (the 50Ω is to balance the network analyzer’s

source impedance) and sent to the feedback signal. This

essentially breaks the loop and injects the AC signal on

top of the DC output voltage and sends it to the

feedback. By monitoring the feedback “R” and output

“A”, gain and phase are measured. This method has no

minimum frequency. Ensure that the bandwidth of the

op-amp being used is much greater than the expected

bandwidth of the power supplies control loop. An op-amp

with >100MHz bandwidth is more than sufficient for most

power supplies (which includes both linear and

switching) and are more common and significantly

cheaper than the injection transformers previously

mentioned. The one disadvantage to using the op-amp

injection method; is the supply voltages need to below

the maximum operating voltage of the op-amp. Also, the

maximum output voltage for driving 50Ω inputs using the

MIC922 is 3V. For measuring higher output voltages,

1MΩ input impedance is required for the A and R

channels. Remember to always measure the output

voltage with an oscilloscope to ensure the measurement

is working properly. You should see a single sweeping

sinusoidal waveform without distortion on the output. If

there is distortion of the sinusoid, reduce the amplitude

of the source signal. You could be overdriving the

feedback causing a large signal response.

Micrel, Inc. MIC4723

June 2008 14 M9999-060308-E

The following Bode analysis show the small signal loop

stability of the MIC4723, it utilizes type III compensation.

This is a dominant low frequency pole, followed by 2

zeros and finally the double pole of the inductor

capacitor filter, creating a final 20dB/decade roll off.

Bode analysis gives us a few important data points;

speed of response (Gain Bandwidth or GBW) and loop

stability. Loop speed or GBW determines the response

time to a load transient. Faster response times yield

smaller voltage deviations to load steps.

Instability in a control loop occurs when there is gain and

positive feedback. Phase margin is the measure of how

stable the given system is. It is measured by determining

how far the phase is from crossing zero when the gain is

equal to 1 (0dB).

-30

-20

-10

GAIN (dB)

FREQUENCY (Hz)

Bode Plot

VIN=3.3V, VOUT=1.8V, IOUT=3A

-105

-70

-35

105

140

175

210

PHASE (°)

100 1k 10k 100k 1M

L=1µH

COUT = 4.7µF

R1 = 10k

R2 = 12.4k

CFF = 82pF

GAIN

PHASE

Typically for 3.3Vin and 1.8Vout at 3A;

• Phase Margin=47 Degrees

• GBW=156KHz

Gain will also increase with input voltage. The following

graph shows the increase in GBW for an increase in

supply voltage.

-30

-20

-10

GAIN (dB)

FREQUENCY (Hz)

Bode Plot

VIN=5V, VOUT=1.8V, IOUT=3A

-105

-70

-35

105

140

175

210

PHASE (°)

100 1k 10k 100k 1M

L=1µH

COUT = 4.7µF

R1 = 10k

R2 = 12.4k

CFF = 82pF

GAIN

PHASE

5Vin, 1.8Vout at 3A load;

• Phase Margin=43.1 Degrees

• GBW= 218KHz

Being that the MIC4723 is non-synchronous; the

regulator only has the ability to source current. This

means that the regulator has to rely on the load to be

able to sink current. This causes a non-linear response

at light loads. The following plot shows the effects of the

pole created by the nonlinearity of the output drive

during light load (discontinuou s) conditions.

-30

-20

-10

GAIN (dB)

FREQUENCY (Hz)

Bode Plot

VIN=3.3V,VOUT=1.8V,IOUT=50mA

-105

-70

-35

105

140

175

210

PHASE (°)

100 1k 10k 100k 1M

L=1µH

COUT = 4.7µF

R1 = 10k

R2 = 12.4k

CFF = 82pF GAIN

PHASE

3.3Vin, 1.8Vout Iout=50mA;

• Phase Margin=90.5 Degrees

• GBW= 64.4KHz

Feed Forward Capacitor

The feedback resistors are a gain reduction block in the

overall system response of the regulator. By placing a

capacitor from the output to the feedback pin, high

frequency signal can bypass the resistor divider, causing

a gain increase up to unity gain.

-10

-9

-8

-7

-6

-5

-4

-3

-2

-1

GAIN (dB)

FREQUENCY (Hz)

Gain and Phase

vs. Frequency

PHASE BOOST (°)

100 1k 10k 100k 1M

L=1µH

COUT = 4.7µF

R1 = 10k

R2 = 12.4k

CFF = 82pF

GAIN

PHASE

The graph above shows the effects on the gain and

phase of the system caused by feedback resistors and a

feedforward capacitor. The maximum amount of phase

boost achievable with a feedforward capacitor is

graphed below.

Micrel, Inc. MIC4723

June 2008 15 M9999-060308-E

12345

PAHSE BOOST (°)

OUTPUT VOLTAGE (V)

Max. Amount of Phase Boost

Obtainable using CFF vs. Output

Voltage

VREF = 1V

By looking at the graph, phase margin can be affected to

a greater degree with higher output voltages.

The next bode plot shows the phase margin of a 1.8V

output at 3A without a feedforward capacitor.

-30

-20

-10

GAIN (dB)

FREQUENCY (Hz)

Bode Plot

VIN=3.3V, VOUT=1.8V, IOUT=3A

-105

-70

-35

105

140

175

210

PHASE (°)

100 1k 10k 100k 1M

L=1µH

COUT = 4.7µF

R1 = 10k

R2 = 12.4k

CFF = 0pF

GAIN

PHASE

As one can see, the typical phase margin, using the

same resistor values as before without a feedforward

capacitor results in 33.6 degrees of phase margin. Our

prior measurement with a feedforward capacitor yielded

a phase margin of 47 degrees. The feedforward

capacitor has given us a phase boost of 13.4 degrees

(47 degrees- 33.6 Degrees = 13.4 Degrees).

Micrel, Inc. MIC4723

June 2008 16 M9999-060308-E

Output Impeda nc e and Transient

Response

Output impedance, simply stated, is the amount of

output voltage deviation vs. the load current deviation.

The lower the output impedance, the better.

OUT

OUT I

Z∆

∆

Output impedance for a buck regulator is the parallel

impedance of the output capacitor and the MOSFET and

inductor divided by the gain;

COUT

LDSON

TOTAL X

GAIN XDCRR

Z++

To measure output impedance vs. frequency, the load

current must be load current must be swept across the

frequencies measured, while the output voltage is

monitored. Figure 9 shows a test set-up to measure

output impedance from 10Hz to 1MHz using the

MIC5190 high speed controller.

Figure 9. Output Impedan ce Measur ement

By setting up a network analyzer to sweep the feedback

current, while monitoring the output of the voltage

regulator and the voltage across the load resistance,

output impedance is easily obtainable. To keep the

current from being too high, a DC offset needs to be

applied to the network analyzer’s source signal. This can

be done with an external supply and 50Ω resistor. Make

sure that the currents are verified with an oscilloscope

first, to ensure the integrity of the signal measurement. It

is always a good idea to monitor the A and R

measurements with a scope while you are sweeping it.

To convert the network analyzer data from dBm to

something more useful (such as peak to peak voltage

and current in our case);

707.0 250Ω1mW10

V10

dBm

×××

=∆

and peak to peak current;

LOAD

dBm

R707.0 250Ω1mW10

I×

×××

=∆

The following graph shows output impedance vs

frequency at 3A load current sweeping the AC current

from 10Hz to 10MHz, at 1A peak to peak amplitude.

0.001

0.01

0.1

OUTPUT IMPEDANCE (Ohms)

FREQUENCY (Hz)

Output Impedance

vs. Frequency

100 1k 10k 100k 1M

VOUT=1.8V

L=1µH

COUT=4.7µF + 0.1µ

5VIN

3.3VIN

From this graph, one can see the effects of bandwidth

and output capacitance. For frequencies <200KHz, the

output impedance is dominated by the gain and

inductance. For frequencies >200KHz, the output

impedance is dominated by the capacitance. A good

approximation for transient response can be calculated

from determining the frequency of the load step in amps

per second;

A/sec

Then, determine the output impedance by looking at the

output impedance vs frequency graph. Then calculating

the voltage deviation times the load step;

OUTOUTOUT ZIV ×

∆

The output impedance graph shows the relationship

between supply voltage and output impedance. This is

caused by the lower Rdson of the high side MOSFET

and the increase in gain with increased supply voltages.

This explains why higher supply voltages have better

transient response.

COUT

LDSON

TOTAL X

GAIN XDCRR

Z↑

++↓

=↓

Micrel, Inc. MIC4723

June 2008 17 M9999-060308-E

Ripple measurements

To properly measure ripple on either input or output of a

switching regulator, a proper ring in tip measurement is

required. Standard oscilloscope probes come with a

grounding clip, or a long wire with an alligator clip.

Unfortunately, for high frequency measurements, this

ground clip can pick-up high frequency noise and

erroneously inject it into the measured output ripple.

The standard evaluation board accommodates a home

made version by providing probe points for both the

input and output supplies and their respective grounds.

This requires the removing of the oscilloscope probe

sheath and ground clip from a standard oscilloscope

probe and wrapping a non-shielded bus wire around the

oscilloscope probe. If there does not happen to be any

non-shielded bus wire immediately available, the leads

from axial resistors will work. By maintaining the

shortest possible ground lengths on the oscilloscope

probe, true ripple measure ments can be obtained.

Micrel, Inc. MIC4723

June 2008 18 M9999-060308-E

MIC4723 Schematic and BOM for 3A Output

Item Part Number Manufacturer Description Qty

C2012JB0J106K TDK

GRM219R60J106KE19 Murata

C1a,C1b

08056D106MAT AVX

10µF Ceramic Capacitor X5R 0805 6.3V 2

C2 0402ZD104MA T AVX 0.1µF Ceramic Capacitor X5R 0402 10V 1

C2012JB0J475K TDK

GRM188R60J475KE19 Murata

06036D475MAT AVX

4.7µF Ceramic Capacitor X5R 0603 6.3V 1

C4 VJ0403A820KXAA Vishay VT 82pF Ceramic Capacitor 0402 1

D1 SSA33L Vishay Semi 3A Schottky 30V SMA 1

RLF7030-1R0N6R4 TDK 1µH Inductor 8.8mΩ 7.1mm(L) x 6.8mm (W)x 3.2mm(H) 1

744 778 9001 Wurth Elektronik 1µH Inductor 12mΩ 7.3mm(L)x7.3mm(W)x3.2mm(H) 1

IHLP2525AH-01 1 Vishay Dale 1µH Inductor 17.5mΩ 6.47mm(L)x6.86mm(W)x1.8mm(H) 1

R1,R4 CRCW04021002F Vishay Dale 10KΩ1% 0402 resistor 1

CRCW04026651F 6.65kΩ 1 % 04 02 For 2.5VOUT

CRCW04021242F 12.4kΩ 1% 0402 For 1.8 VOUT

CRCW04022002F 20kΩ 1% 0402 For 1.5 VOUT

CRCW04024022F

Vishay Dale

49.9kΩ 1% 0402 For 1.2 VOUT

Open For 1.0 VOUT

R3 CRCW040210R0F Vishay Dale 10Ω1% 0402 resistor 1

U1 MIC4723YML Micrel, Inc. 3 A 2MHz Integrated Switch Buck Regulator 1

Notes:

1. TDK: www.tdk.com

2. Murata: www.murata.com

3. AVX: www.avx.com

4. Vishay: www.vishay.com

5. Wurth Elektronik: www.we-online.com

6. Micrel, Inc: www.micrel.com

Micrel, Inc. MIC4723

June 2008 19 M9999-060308-E

Package Information

12-Pin 3mm x 3mm MLF® (ML)

10-Pin ePAD MSOP (MME)

Micrel, Inc. MIC4723

June 2008 20 M9999-060308-E

MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA

TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com

The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its

use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.

Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product

can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant

into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A

Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully

indemnify Micrel for any damages resulting from such use or sale.