AD8346ARUZ-REEL7 - ANALOG DEVICES

0.8 GHz to 2.5 GHz

Quadrature Modulator

AD8346

Rev. A

Information furnished by Analog Devices is believed to be accurate and reliable.

However, no responsibility is assumed by Analog Devices for its use, nor for any

infringements of patents or other rights of third parties that may result from its use.

Specifications subject to change without notice. No license is granted by implication

or otherwise under any patent or patent rights of Analog Devices. Trademarks and

registered trademarks are the property of their respective companies.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.

Tel: 781.329.4700 www.analog.com

FEATURES

High accuracy

1 degree rms quadrature error @ 1.9 GHz

0.2 dB I/Q amplitude balance @ 1.9 GHz

Broad frequency range: 0.8 GHz to 2.5 GHz

Sideband suppression: −46 dBc @ 0.8 GHz

Sideband suppression: −36 dBc @ 1.9 GHz

Modulation bandwidth: dc to 70 MHz

0 dBm output compression level @ 0.8 GHz

Noise floor: −147 dBm/Hz

Single 2.7 V to 5.5 V supply

Quiescent operating current: 45 mA

Standby current: 1 μA

16-lead TSSOP

APPLICATIONS

Digital and spread spectrum communication systems

Cellular/PCS/ISM transceivers

Wireless LAN/wireless local loop

QPSK/GMSK/QAM modulators

Single-sideband (SSB) modulators

Frequency synthesizers

Image reject mixer

FUNCTIONAL BLOCK DIAGRAM

8BIAS

PHASE

SPLITTER

AD8346

IBBP

IBBN

COM1

LOIN

LOIP

VPS1

ENBL

QBBP

QBBN

COM4

VPS2

VOUT

COM3

COM2

05335-001

Figure 1.

GENERAL DESCRIPTION

The AD8346 is a silicon RFIC I/Q modulator for use from

0.8 GHz to 2.5 GHz. Its excellent phase accuracy and amplitude

balance allow high performance direct modulation to RF.

The differential LO input is applied to a polyphase network

phase splitter that provides accurate phase quadrature from

0.8 GHz to 2.5 GHz. Buffer amplifiers are inserted between

two sections of the phase splitter to improve the signal-to-

noise ratio. The I and Q outputs of the phase splitter drive the

LO inputs of two Gilbert-cell mixers. Two differential V-to-I

converters connected to the baseband inputs provide the

baseband modulation signals for the mixers. The outputs of

the two mixers are summed together at an amplifier which is

designed to drive a 50 Ω load.

This quadrature modulator can be used as the transmit mod-

ulator in digital systems such as PCS, DCS, GSM, CDMA, and

ISM transceivers. The baseband quadrature inputs are directly

modulated by the LO signal to produce various QPSK and

QAM formats at the RF output.

Additionally, this quadrature modulator can be used with direct

digital synthesizers in hybrid phase-locked loops to generate

signals over a wide frequency range with millihertz resolution.

The AD8346 comes in a 16-lead TSSOP package, measuring

6.5 mm × 5.1 mm × 1.1 mm. It is specified to operate over a

−40°C to +85°C temperature range and a 2.7 V to 5.5 V supply

voltage range. The device is fabricated on Analog Devices’ high

performance 25 GHz bipolar silicon process.

AD8346

Rev. A | Page 2 of 20

TABLE OF CONTENTS

Specifications..................................................................................... 3

Absolute Maximum Ratings............................................................ 4

ESD Caution.................................................................................. 4

Pin Configuration and Function Descriptions............................. 5

Equivalent Circuits ........................................................................... 6

Typical Performance Characteristics ............................................. 7

Circuit Description......................................................................... 10

Overview...................................................................................... 10

LO Interface................................................................................. 10

V-to-I Converter......................................................................... 10

Mixers .......................................................................................... 10

Differential-to-Single-Ended Converter ................................. 10

Bias ............................................................................................... 10

Basic Connections...................................................................... 11

LO Drive...................................................................................... 11

RF Output.................................................................................... 11

Interface to AD9761 TXDAC® .................................................. 12

AC-Coupled Interface ............................................................... 13

Evaluation Board ............................................................................ 14

Characterization Setups................................................................. 16

SSB Setup..................................................................................... 16

CDMA Setup............................................................................... 17

Outline Dimensions ....................................................................... 18

Ordering Guide .......................................................................... 18

REVISION HISTORY

6/05—Rev. 0 to Rev. A

Updated Format..................................................................Universal

Changes to Figures 30, 31, 32........................................................ 14

Update Outline Dimensions ......................................................... 18

Changes to Ordering Guide .......................................................... 18

3/99—Revision 0: Initial Version

AD8346

Rev. A | Page 3 of 20

SPECIFICATIONS

VS = 5 V; TA = 25°C; LO frequency = 1900 MHz; LO level = –10 dBm; BB frequency = 100 kHz; BB inputs are dc-biased to 1.2 V; BB input

level = 1.0 V p-p each pin for 2.0 V p-p differential drive; LO source and RF output load impedances are 50 Ω, dBm units are referenced

to 50 Ω unless otherwise noted.

Table 1.

Parameters Conditions Min Typ Max Unit

RF OUTPUT

Operating Frequency 0.8 2.5 GHz

Quadrature Phase Error See Figure 35 for setup 1 Degree rms

I/Q Amplitude Balance See Figure 35 for setup 0.2 dB

Output Power I and Q channels in quadrature −13 −10 −6 dBm

Output VSWR 1.25:1

Output P1 dB −3 dBm

Carrier Feedthrough −42 −35 dBm

Sideband Suppression −36 −25 dBc

IM3 Suppression −60 dBc

Equivalent Output IP3 20 dBm

Output Noise Floor 20 MHz offset from LO −147 dBm/Hz

RESPONSE TO CDMA IS95 BASEBAND SIGNALS

ACPR (Adjacent Channel Power Ratio) See Figure 35 for setup −72 dBc

EVM (Error Vector Magnitude) See Figure 35 for setup 2.5 %

Rho (Waveform Quality Factor) See Figure 35 for setup 0.9974

MODULATION INPUT

Input Resistance 12 kΩ

Modulation Bandwidth −3 dB 70 MHz

LO INPUT

LO Drive Level −12 −6 dBm

Input VSWR 1.9:1

ENABLE

ENBL HI Threshold 2.0 V

ENBL LO Threshold 0.5 V

ENBL Turn-On Time Settle to within 0.5 dB of final SSB

output power

2.5 μs

ENBL Turn-Off Time Time for supply current to drop below

2 mA

12 μs

POWER SUPPLIES

Voltage 2.7 5.5 V

Current Active (ENBL HI) 35 45 55 mA

Current Standby (ENBL LO) 1 20 μA

AD8346

Rev. A | Page 4 of 20

ABSOLUTE MAXIMUM RATINGS

Table 2.

Parameter Min Rating

Supply Voltage VPS1, VPS2 5.5 V

Input Power LOIP, LOIN (relative to 50 Ω) 10 dBm

Min Input Voltage IBBP, IBBN, QBBP, QBBN 0 V

Max Input Voltage IBBP, IBBN, QBBP, QBBN 2.5 V

Internal Power Dissipation 500 mW

θJA 125°C/W

Operating Temperature Range −40°C to +85°C

Storage Temperature Range −65°C to +150°C

Lead Temperature (Soldering 60 sec) 300°C

Stresses above those listed under Absolute Maximum Ratings

may cause permanent damage to the device. This is a stress

rating only; functional operation of the device at these or any

other condition s above those indicated in the operational

section of this specification is not implied. Exposure to absolute

maximum rating conditions for extended periods may affect

device reliability.

ESD CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the

human body and test equipment and can discharge without detection. Although this product features

proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy

electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance

degradation or loss of functionality.

AD8346

Rev. A | Page 5 of 20

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

IBBP QBBP

AD8346

TOP VIEW

(Not to Scale)

IBBN QBBN

COM1 COM4

LOIN VPS2

LOIP VOUT

VPS1 COM3

ENBL COM2

05335-002

Figure 2. Pin Configuration

Table 3. Pin Function Descriptions

Pin No. Mnemonic Description

Equivalent

Circuit

1 IBBP I Channel Baseband Positive Input Pin. Input should be dc-biased to approximately 1.2 V.

Nominal characterized ac swing is 1 V p-p (0.7 V to 1.7 V). This makes the differential input

2 V p-p when IBBN is 180 degrees out of phase from IBBP.

Circuit A

2 IBBN I Channel Baseband Negative Input Pin. Input should be dc-biased to approximately 1.2 V.

Nominal characterized ac swing is 1 V p-p (0.7 V to 1.7 V). This makes the differential input

2 V p-p when IBBN is 180 degrees out of phase from IBBP.

Circuit A

3 COM1 Ground Pin for the LO phase splitter and LO buffers.

4 COM1 Ground Pin for the LO phase splitter and LO buffers.

5 LOIN LO Negative Input Pin. Internal dc bias (approximately VPS1 to 800 mV) is supplied. This

pin must be ac coupled.

Circuit B

6 LOIP LO Positive Input Pin. Internal dc bias (approximately VPS1 to 800 mV) is supplied. This pin

must be ac-coupled.

Circuit B

7 VPS1 Power Supply Pin for the bias cell and LO buffers. This pin should be decoupled using

local 100 pF and 0.01 μF capacitors.

8 ENBL Enable Pin. A high level enables the device; a low level puts the device in sleep mode. Circuit C

9 COM2 Ground Pin for the input stage of output amplifier.

10 COM3 Ground Pin for the output stage of output amplifier.

11 VOUT 50 Ω DC-Coupled RF Output. User must provide ac coupling on this pin. Circuit D

12 VPS2 Power Supply Pin for baseband input voltage to current converters and mixer core. This

pin should be decoupled using local 100 pF and 0.01 μF capacitors.

13 COM4 Ground Pin for baseband input voltage to current converters and mixer core.

14 COM4 Ground Pin for baseband input voltage to current converters and mixer core.

15 QBBN Q Channel Baseband Negative Input. Input should be dc biased to approximately 1.2 V.

Nominal characterized ac swing is 1 V p-p. This makes the differential input 2 V p-p when

QBBN is 180° out of phase from QBBP.

Circuit A

16 QBBP Q Channel Baseband Positive Input. Input should be dc-biased to approximately 1.2 V.

Nominal characterized ac swing is 1 V p-p. This makes the differential input 2 V p-p when

QBBN is 180° out of phase from QBBP.

Circuit A

AD8346

Rev. A | Page 6 of 20

EQUIVALENT CIRCUITS

3kΩ

9kΩ

VPS2

INPUT

BUFFER TO MIXER

CORE

ACTIVE LOADS

05335-003

Figure 3. Circuit A

VPS1

LOIN

LOIP

PHASE

SPLITTER

CONTINUES

05335-004

Figure 4. Circuit B

40kΩ

30kΩ

VPS1

TO BIAS FOR

STARTUP/

SHUTDOWN

780Ω

75kΩ

ENBL

75kΩ

05335-005

Figure 5. Circuit C

43Ω

VPS2

OUT

05335-006

Figure 6. Circuit D

AD8346

Rev. A | Page 7 of 20

TYPICAL PERFORMANCE CHARACTERISTICS

LO FREQUENCY (MHz)

SSB POWER (dBm)

–6

–7

–8

–9

–10

–11

–12

–13

–14

–15 1200 1600 2000 2400800 1400 1800 22001000

T = 25°C

VP = 5.5V

VP = 5V

VP = 2.7V

VP = 3V

05335-007

Figure 7. Single Sideband (SSB) Output Power (POUT) vs. LO Frequency (FLO).

I and Q inputs driven in quadrature at baseband frequency

(FBB) = 100 kHz with differential amplitude of 2.00 V p-p.

SSB OUTPUT POWER (dBm)

–6

–7

–8

–9

–10

–11

–12

–13

LO = 1900MHz, –10dBm

LO = 800MHz, –10dBm

LO = 1900MHz, –6dBm

LO = 800MHz, –6dBm

TEMPERATURE (°C)

–40 –20 0 20304050607080–30 –10 10

05335-008

Figure 8. SSB POUT vs. Temperature. I and Q inputs driven in quadrature

with differential amplitude of 2.00 V p-p at FBB = 100 kHz.

TEMPERATURE (°C)

CARRIER FEEDTHROUGH (dBm)

–35

–40 –20 0 20304050607080

–37

–39

–41

–43

–45

–47

–49

–51 –30 –10 10

= 5V

= 5.5V

= 3V

= 2.7V

05335-009

Figure 9. Carrier Feedthrough vs. Temperature.

FLO = 1900 MHz, LO input level = –10 dBm.

BASEBAND FREQUENCY (MHz)

OUTPUT POWER VARIATION (dB)

0.1 1 100

–1

–2

–3

–4

–5

–6

–7

–8

05335-010

Figure 10. I and Q Input Bandwidth. FLO =1900 MHz, I or Q inputs

driven with differential amplitude of 2.00 V p-p.

LO FREQUENCY (MHz)

SSB OUTPUT P1dB (dBm)

800

–2

–4

–6

–8

–10

–12

1000

–14 1200 1400 1600 1800 2000 2200 2400

= 2.7V

T = –40°C

= 2.7V

T = +85°C

= 5V

T = –40°C

= 5V

T = +85°C

05335-011

Figure 11. SSB Output 1 dB Compression Point (OP 1 dB) vs. FLO.

I and Q inputs driven in quadrature at FBB = 100 kHz.

CARRIER FEEDTHROUGH (dBm/

AFTER NULLING TO <–60dBm @ 25°C)

PERCENTAGE

–90

–86

0–82–78–74–70–66–62–58–54–50–46

= +85

°C

= –40°C

05335-012

Figure 12. Histogram Showing Carrier Feedthrough Distributions

at the Temperature Extremes after Nulling at Ambient

at FLO = 1900 MHz, LO Input Level = –10 dBm.

AD8346

Rev. A | Page 8 of 20

SSB OUTPUT POWER (dBm)

–7

–8

–9

–10

–11

–12

–13

–14

–15

VP = 5V

VP = 3V

VP = 5.5V

VP = 2.7V

TEMPERATURE (°C)

–40 –20 0 20304050607080–30 –10 10

05335-013

Figure 13. SSB POUT vs. Temperature. FLO = 1900 MHz, I and Q inputs driven in

quadrature with differential amplitude of 2.00 V p-p at FBB = 100 kHz.

LO FREQUENCY (MHz)

CARRIER FEEDTHROUGH (dBm)

–36

–38

–40

–42

–44

–46

–48

–50

–52

–54

1200 1600 2000 2400800 1400 1800 22001000

T = 25°C

= 3V

= 5.5V

= 5V

= 2.7V

05335-014

Figure 14. Carrier Feedthrough vs. FLO.

LO input level = –10 dBm.

LO FREQUENCY (MHz)

SIDEBAND SUPPRESSION (dBc)

–32

–34

–36

–38

–40

–42

–44

–46

–48 1300 1700 2100 2500900 1500 1900 23001100

T = 25°C

= 3V

= 2.7V

= 5V

= 5.5V

05335-015

Figure 15. Sideband Suppression vs. FLO. VPOS = 2.7 V, I and Q inputs driven in

quadrature with differential amplitude of 2.00 V p-p at FBB = 100 kHz.

BASEBAND FREQUENCY (MHz)

SB SUPPRESSION (dBc)

–30

–32

–34

–36

–38

–40

–42

–44 2 4 6 8 10 12 14 16 18 20

05335-016

= 3V V

= 5.5V

= 2.7V

= 5V

Figure 16. Sideband Suppression vs. FBB. FLO = 1900 MHz, I and Q inputs

driven in quadrature with differential amplitude of 2.00 V p-p.

INPUT THIRD HARMONIC

DISTORTION (dBc)

–35

–40

–45

–50

–55

–60

–65

–70

05335-017

TEMPERATURE (°C)

–40 –20 0 20304050607080–30 –10 10

= 5.5V

= 2.7V

= 3V

= 5V

Figure 17. Third Harmonic Distortion vs. Temperature.

FLO =1900 MHz, I and Q inputs driven in quadrature with

differential amplitude of 2.00 V p-p at FBB = 100 kHz.

FREQUENCY (MHz)

–2

–4

–6

–8

–10

–12

–14

800 1200 1600 2000

–20

RETURN LOSS (dB)

2400

–16

–18

1400 1800 22001000

05335-018

T = +25

°C

T = +85°C

T = –40°C

Figure 18. Return Loss of LOIN Input vs. FLO.

VPOS = 5.0 V, LOIP pin ac-coupled to ground.

AD8346

Rev. A | Page 9 of 20

SB SUPPRESSION (dBc)

–30

–32

–34

–36

–38

–40

–42

–44

05335-019

TEMPERATURE (°C)

–40 –20 0 20304050607080–30 –10 10

= 5V

= 2.7V

= 3V

= 5.5V

Figure 19. Sideband Suppression vs. Temperature.

FLO = 1900 MHz, I and Q inputs driven in quadrature

with differential amplitude of 2.00 V p-p at FBB = 100 kHz.

BASEBAND DIFFERENTIAL INPUT

VOLTAGE (V

p-p)

INPUT THIRD HARMONIC

DISTORTION (dBc)

–30

0.5

–35

–40

–45

–50

–55

1.0 1.5 2.0 2.5 3.0

–60

–65

–70

–75

–80

–6

–8

–10

–12

–14

–16

–18

–20

–22

SSB OUTPUT POWER (dBm)

05335-020

SSB P

OUT

3RD HARMONIC

Figure 20. Third Harmonic Distortion and SSB Output

Power vs. Baseband Differential Input Voltage Level.

FLO = 1900 MHz, I and Q inputs driven in quadrature at FBB = 100 kHz.

FREQUENCY (MHz)

–5

–10

–15

–20

–25

–30

800 1200 1600 2000

–40

RETURN LOSS (dB)

2400

–35

1400 1800 22001000

05335-021

T = –40°C

T = +25°C

T = +85°C

Figure 21. Return Loss of VOUT Output vs. FLO.

VPOS = 2.7 V.

BASEBAND FREQUENCY (MHz)

INPUT THIRD HARMONIC

DISTORTION (dB)c

–40

–45

–50

–55

–60

–65 2468101214161820

05335-022

= 5V

= 5.5V

= 2.7V

= 3V

Figure 22. Third Harmonic Distortion vs. FBB.

FLO =1900 MHz, I and Q inputs driven in quadrature

with differential amplitude of 2.00 V p-p.

SUPPLY CURRENT (mA)

05335-023

TEMPERATURE (°C)

–40 –20 0 20 40 60 80

= 5V

= 2.7V

= 3V

= 5.5V

Figure 23. Power Supply Current vs. Temperature

FREQUENCY (MHz)

–5

–10

–15

–20

–25

–30

800 1200 1600 2000

–40

RETURN LOSS (dB)

2400

–35

1400 1800 22001000

05335-024

T = –40°C

T = +25°C

T = +85°C

Figure 24. Return Loss of VOUT Output vs. FLO.

VPOS = 5.0 V.

AD8346

Rev. A | Page 10 of 20

CIRCUIT DESCRIPTION

OVERVIEW

The AD8346 can be divided into the following sections: local

oscillator (LO) interface, mixer, voltage-to-current (V-to-I)

converter, differential-to-single-ended (D-to-S) converter, and

bias. A detailed block diagram of the part is shown in Figure 25.

The LO interface generates two LO signals, with 90° of phase

difference between them, to drive two mixers in quadrature.

Baseband voltage signals are converted into current form in

the V-to-I converters, feeding into two mixers. The output of

the mixers are combined to feed the D-to-S converter which

provides the 50 Ω output interface. Bias currents to each

section are controlled by the Enable (ENBL) signal. Detailed

descriptions of each section follows.

LO INTERFACE

The differential LO inputs allow the user to drive the LO differ-

entially in order to achieve maximum performance. The LO can

be driven single-endedly but the LO feedthrough performance

is degraded, especially towards the higher end of the frequency

range. The LO interface consists of interleaved stages of

polyphase network phase splitters and buffer amplifiers. The

phase-splitter contains resistors and capacitors connected in a

circular manner to split the LO signal into I and Q paths in

precise quadrature with each other. The signal on each path

goes through a buffer amplifier to make up for the loss and high

frequency roll-off. The two signals then go through another

polyphase network to enhance the quadrature accuracy. The

broad operating frequency range of 0.8 GHz to 2.5 GHz is

achieved by staggering the RC time constants in each stage of

the phase-splitters. The outputs of the second phase-splitter are

fed into the driver amplifiers for the mixers’ LO inputs.

V-TO-I CONVERTER

Each baseband input pin is connected to an op amp driving an

emitter follower. Feedback at the emitter maintains a current

proportional to the input voltage through the transistor. This

current is fed to the two mixers in differential form.

MIXERS

There are two double-balanced mixers, one for the in-phase

channel (I-channel) and one for the quadrature channel

(Q channel). Each mixer uses the gilbert cell design with four

cross-connected transistors. The bases of the transistors are

driven by the LO signal of the corresponding channel. The

output currents from the two mixers are summed together in

two resistors in series with two coupled on-chip inductors. The

signal developed across the R-L loads is sent to the D-to-S stage.

DIFFERENTIAL-TO-SINGLE-ENDED CONVERTER

The differential-to-single-ended converter consists of two

emitter followers driving a totem-pole output stage. Output

impedance is established by the emitter resistors in the output

transistors. The output of this stage is connected to the output

(VOUT) pin.

BIAS

A band gap reference circuit based on the Δ-VBE principle

generates the proportional-to-absolute-temperature (PTAT)

currents used by the different sections as references. The band

gap voltage is also used to generate a temperature-stable current

in the V-to-I converters to produce a temperature-independent

slew rate. When the band gap reference is disabled by pulling

down the ENBL pin, all other sections are shut off accordingly.

MIXER

V-TO-I V-TO-I

D-TO-S

BIAS CELL

AD8346

LOIN

LOIP

ENBL

QBBP QBBN

IBBNIBBP

PHASE

SPLITTER

PHASE

SPLITTER

05335-025

Figure 25. Detailed Block Diagram

AD8346

Rev. A | Page 11 of 20

BASIC CONNECTIONS

The basic connections for operating the AD8346 are shown in

Figure 27. A single power supply of between 2.7 V and 5.5 V is

applied to pins VPS1 and VPS2. A pair of ESD protection

diodes are connected internally between VPS1 and VPS2 so

these must be tied to the same potential. Both pins should be

individually decoupled using 100 pF and 0.01 μF capacitors,

located as close as possible to the device. For normal operation,

the enable pin, ENBL, must be pulled high. The turn-on

threshold for ENBL is 2 V. To put the device in its power-down

mode, ENBL must be pulled below 0.5 V. Pins COM1 to COM4

should all be tied to a low impedance ground plane.

The I and Q ports should be driven differentially. This is con-

venient as most modern high speed DACs have differential

outputs. For optimal performance, the drive signal should be a

2 V p-p (differential) signal with a bias level of 1.2 V, that is,

each input swings from 0.7 V to 1.7 V. The I and Q inputs have

input impedances of 12 kΩ. By dc coupling the DAC to the

AD8346 and applying small offset voltages, the LO feedthrough

can be reduced to well below its nominal value of −42 dBm

(see Figure 12).

LO DRIVE

The return loss of the LO port is shown in Figure 18. No add-

itional matching circuitry is required to drive this port from a

50 Ω source. For maximum LO suppression at the output, a

differential LO drive is recommended. In Figure 27, this is

achieved using a balun (M/A-COM Part Number ETC1-1-13).

The output of the balun is ac-coupled to the LO inputs which

have a bias level about 800 mV below supply. An LO drive

level of between −6 dBm and −12 dBm is required. For optimal

performance, a drive level of −10 dBm is recommended,

although a level of −6 dBm results in more stable temperature

performance (see Figure 8). Higher levels degrade linearity

while lower levels tend to increase the noise floor.

LOIP

LOIN

AD8346

100pF

05335-026

Figure 26. Single-Ended LO Drive

The LO terminal can be driven single-ended, as shown in

Figure 26 at the expense of slightly higher LO feedthrough.

LOIN is ac coupled to ground using a capacitor and LOIP is

driven through a coupling capacitor from a (single-ended)

50 Ω source (this scheme could also be reversed with LOIP

being ac-coupled to ground).

RF OUTPUT

The RF output is designed to drive a 50 Ω load, but must be ac-

coupled, as shown in Figure 27. If the I and Q inputs are driven

in quadrature by 2 V p-p signals, the resulting output power is

about −10 dBm (see Figure 7 for variations in output power

over frequency).

QBBP

IBBP

AD8346

QBBN

IBBN

COM4

COM1

COM4

COM1

VPS2

LOIN

VOUT

LOIP

COM3

VPS1

COM2

ENBL

0.01μFC3

100pF

ETC1-1-13

0.01μF

100pF

VOUT

05335-027

Figure 27. Basic Connections

AD8346

Rev. A | Page 12 of 20

INTERFACE TO AD9761 TXDAC®

Figure 28 shows a dc-coupled current output DAC interface.

The use of dual-integrated DACs, such as the AD9761 with

specified ±0.02 dB and ±0.004 dB gain and offset matching

characteristics, ensures minimum error contribution (over

temperature) from this portion of the signal chain. The use of a

precision thin-film resistor network sets the bias levels precisely

to prevent the introduction of offset errors, which increase LO

feedthrough. For instance, selecting resistor networks with a

0.1% ratio matching characteristics maintains 0.03 dB gain and

offset matching performance.

Using resistive division, the dc bias level at the I and Q inputs

to the AD8346 is set to approximately 1.2 V. Each of the four

current outputs of the DAC delivers a full-scale current of

10 mA, giving a voltage swing of 0 V to 1 V (at the DAC

output). This results in a 0.5 V p-p swing at the I and Q inputs

of the AD8346 (resulting in a 1 V p-p differential swing).

Note that the ratio matching characteristics of the resistive

network, as opposed to its absolute accuracy, is critical in

preserving the gain and offset balance between the I and Q

signal path.

By applying small dc offsets to the I and Q signals from the

DAC, the LO suppression can be reduced from its nominal

value of −42 dBm to as low as −60 dBm while holding to

approximately −50 dBm over temperature (see Figure 12 for

a plot of LO feedthrough over temperature for an offset

compensated circuit).

DAC

LATCH

IIOUTB

IOUTA

DAC

LATCH

QQOUTB

QOUTA

MUX

CONTROL

SELECT

WRITE

CLOCK

AD9761

DVDD DCOM AVDD

0.1

SET

SLEEP FS ADJ REFIO

DAC

DATA

INPUTS

FILTER

100

FILTER

100

500

0.1

634

PHASE

SPLITTER

VOUT

IBBP

IBBN

QBBP

QBBN

AD8346

LOIP

LOIN

VPS1 VPS2

0.5V p-p EACH PIN

WITH V

= 1.2V

+5V

05335-028

Figure 28. AD8346 Interface to AD9761 TxDAC

AD8346

Rev. A | Page 13 of 20

AC-COUPLED INTERFACE

An ac-coupled interface can also be implemented, as shown in

Figure 29. This is an advantage because there is almost no

voltage loss due to the biasing network, allowing the AD8346

inputs to be driven by the full 2 V p-p differential signal from

the AD9761 (each of the DAC’s 4 outputs delivering 1 V p-p).

As in the dc-coupled case, the bias levels on the I and Q inputs

should be set to as precise a level as possible, relative to each

other. This prevents the introduction of additional input offset

voltages. In Figure 29, the bias level on each input is set to

approximately 1.2 V. The 2.43 kΩ resistors should have a ratio

tolerance of 0.1% or better.

The network shown has a high-pass corner frequency of

approximately 14.3 kHz (note that the 12 kΩ input impedance

of the AD8346 has been factored into this calculation).

Increasing the resistors in the network or increasing the

coupling capacitance reduces the corner frequency further.

Note that the LO suppression can be manually optimized by

replacing a portion of the four top 2.43 kΩ resistors with

potentiometers. In this case, the bottom four resistors in the

biasing network no longer need to be precision devices.

2×

LATCH

IIOUTB

IOUTA

2×

LATCH

QQOUTB

QOUTA

MUX

CONTROL

SELECT

WRITE

CLOCK

AD9761

DVDD DCOM AVDD

0.1μF

SET

2kΩ

SLEEP FS ADJ REFIO

DAC

DATA

INPUTS

FILTER

100Ω

FILTER

100Ω

0.1μF

1kΩ

PHASE

SPLITTER

VOUT

IBBP

IBBN

QBBP

QBBN

AD8346

LOIP

LOIN

VPS1 VPS2

1V p-p EACH PIN

WITH V

= 1.2V

0.01μF

DAC 2.43kΩ

2.43kΩ

DAC

05335-029

Figure 29. AC-Coupled DAC Interface

AD8346

Rev. A | Page 14 of 20

EVALUATION BOARD

The schematic of the AD8346 evaluation board is shown in

Figure 30. This is a 4-layer FR4 board; the two center layers are

used as ground planes and the top and bottom layers are used

for signal and power. Figure 31 shows the layout and Figure 32

shows the silkscreen. The evaluation board circuit closely

follows the basic connections circuit shown in Figure 27.

Slide SW1 to the A position to connect the ENBL pin to +VS

via the 10 kΩ pull-up resistor REP. Slide SW1 to the B position

to disable the device by grounding the ENOP pin through the

49.9 Ω pull-down resistor REG. The device may be enabled via

an external voltage applied to the SMA connector ENOP or TP2.

All connectors are of the SMA type. The I and Q inputs are

provided with pads for implementing a simple RC filter

network. The local oscillator input is driven through a balun

(M/A-COM Part Number ETC1-1-13).

05335-030

100pF C3

0.01

CLOP

100pF

CLON

100pF

RLOP

OPEN

RLON

OPEN

RLOS

OPEN

IBBP

IBBN

COM1

LOIN

LOIP

VPS1

ENBL

QBBP

QBBN

COM4

VPS2

VOUT

COM3

COM2

TP2

ENOP

CVO

100pF VOUT

AD8346

RQP

CQP

OPEN

RQN

CIP

OPEN

RIP

RIN

CIN

OPEN

0.01

FC2

100pF

ETC1-1-13

RIS

OPEN RQS

OPEN

CQN

OPEN

REG

49.9k

REP

10k

SW1

Figure 30. Evaluation Board Schematic

AD8346

Rev. A | Page 15 of 20

05335-031

Figure 31. Layout of Evaluation Board

05335-032

Figure 32. Silkscreen of Evaluation Board

AD8346

Rev. A | Page 16 of 20

CHARACTERIZATION SETUPS

SSB SETUP

Two main setups were used to characterize this product. These

setups are shown in Figure 33 and Figure 35. Figure 33 shows

the setup used to evaluate the product as an SSB. The AD8346

motherboard had circuitry that converted the single-ended

I and Q inputs from the arbitrary function generator to differ-

ential inputs with a dc bias of approximately 1.2 V. In addition,

the motherboard also provided connections for power supply

routing. The HP34970A and its associated plug-in 34901 were

used to monitor power supply currents and voltages being

supplied to the AD8346 evaluation board (a full schematic of

the AD8346 evaluation board can be found in Figure 30).

The two HP34907 plug-ins were used to provide additional

miscellaneous dc and control signals to the motherboard. The

LO was driven by an RF signal generator (through the balun on

the evaluation board to present a differential LO signal to the

device) and the output was measured with a spectrum analyzer.

With the I channel driven with a sine wave and the Q channel

driven with a cosine wave, the lower sideband is the single

sideband output. The typical SSB output spectrum is shown in

Figure 34.

VPS1

GND

AD8346

MOTHERBOARD

I IN

QIN

D1 D2 D3

P1 IN IP QP QN

IEEE

34901 34907 34907

D1 D2 D3

HP34970A

AD8346

EVAL BOARD

IN IP QP

ENBL VOUT

RFOUT

IEEE

HP8648C

+15V MAX

COM

+25V MAX

–25V MAX

IEEE

HP3631

IEEE

PC CONTROLLER

HP8593E

RF I/P

CAL OUT

28VOLT

IEEE

SPECTRUM

ANALYZER

SWEEP OUT

OUTPUT 1

OUTPUT 2IEEE

TEKAFG2020

ARB FUNC. GEN

05335-033

Figure 33. Evaluation Board SSB Test Setup

CENTER 1.9GHz

–10

–20

–30

–40

–50

–60

–70

–80

–90

–

100 50kHz/ SPAN 500kHz

05335-034

Figure 34. Typical SSB Output Spectrum

AD8346

Rev. A | Page 17 of 20

CDMA SETUP

For evaluating the AD8346 with CDMA waveforms, the setup

shown in Figure 35 was used. This is essentially the same setup

as that used for the single sideband characterization, except that

the AFG2020 was replaced with the AWG2021 for providing the

I and Q input signals, and the spectrum analyzer used to monitor

the output was changed to an FSEA30 Rohde & Schwarz analyzer

with vector demodulation capability. The I/Q input signals for

these measurements were IS95 baseband signals generated with

Tektronix I/Q SIM software and downloaded to the AWG2021.

For measuring ACPR, the I/Q input signals used were generated

with Pilot (Walsh Code 00), Sync (WC 32), Paging (WC 01),

and 6 Traffic (WC 08, 09, 10, 11, 12, 13) channels active. The

I/Q SIM software was set for 32× oversampling and was using a

BS equifilter. Figure 36 shows the typical output spectrum for

this configuration. The ACPR was measured 885 kHz away

from the carrier frequency.

For performing EVM, Rho, phase, and amplitude balance

measurements, the I/Q input signals used were generated with

only the pilot channel (Walsh Code 00) active. The I/Q SIM

software was set for 32× oversampling using a CDMA equifilter.

VPS1

GND

AD8346

MOTHERBOARD

I IN

QIN

D1 D2 D3

P1 IN IP QP QN

IEEE

34901 34907 34907

D1 D2 D3

HP34970A

AD8346

EVAL BOARD

IN IP QP

ENBL VOUT

RFOUT

IEEE

HP8648C

+15V MAX

COM

+25V MAX

–25V MAX

IEEE

HP3631

IEEE

PC CONTROLLER

FSEA30

RF I/P IEEE

SPECTRUM

ANALYZER

OUTPUT 1

OUTPUT 2IEEE

TEKAFG2020

ARB FUNC. GEN

05335-035

Figure 35. Evaluation Board CDMA Test Setup

–20

CENTER 1.9GHz

–30

–40

–50

–60

–70

–80

–90

–

100

–

110

–

120 187.5kHz/ SPAN 1.875MHz

CH PWR = –20.7dBm

ACP UPR = –71.8dBc

ACP LWR = –71.7dBc

05335-036

Figure 36. Typical CDMA Output Spectrum

AD8346

Rev. A | Page 18 of 20

OUTLINE DIMENSIONS

16 9

PIN 1

SEATING

PLANE

8°

0°

4.50

4.40

4.30

6.40

BSC

5.10

5.00

4.90

0.65

BSC

0.15

0.05

1.20

MAX 0.20

0.09 0.75

0.60

0.45

0.30

0.19

COPLANARITY

0.10

COMPLIANT TO JEDEC STANDARDS MO-153AB

Figure 37.16-Lead Thin Shrink Small Outline Package [TSSOP]

(RU-16)

Dimensions shown in millimeters

ORDERING GUIDE

Model Temperature Range Package Description Package Option

AD8346ARU −40°C to +85°C 16-Lead Thin Shrink Small Outline Package (TSSOP) RU-16

AD8346ARU-REEL −40°C to +85°C 16-Lead (TSSOP) 13" Tape and Reel RU-16

AD8346ARU-REEL7 −40°C to +85°C 16-Lead (TSSOP) 7" Tape and Reel RU-16

AD8346ARUZ-REEL1 −40°C to +85°C 16-Lead (TSSOP) 13" Tape and Reel RU-16

AD8346ARUZ-REEL71−40°C to +85°C 16-Lead (TSSOP) 7" Tape and Reel RU-16

AD8346-EVAL Evaluation Board

1 Z = Pb-free part.

AD8346

Rev. A | Page 19 of 20

NOTES

AD8346

Rev. A | Page 20 of 20

NOTES

registered trademarks are the property of their respective companies.

C05335–0–6/05(A)