Dual, Interleaved, Step-Down
DC-to-DC Controller with Tracking
ADP1823
Rev. D
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FEATURES
Fixed-frequency operation: 300 kHz, 600 kHz, or
synchronized operation up to 1 MHz
Supply input range: 3.7 V to 20 V
Wide power stage input range: 1 V to 24 V
Interleaved operation results in smaller, low cost input
capacitor
All-N-channel MOSFET design for low cost
±0.85% accuracy at 0°C to 70°C
Soft start, thermal overload, current-limit protection
10 μA shutdown supply current
Internal linear regulator
Lossless RDSON current-limit sensing
Reverse current protection during soft start for handling
precharged outputs
Independent Power OK (POK) outputs
Voltage tracking for sequencing or DDR termination
Available in 5 mm × 5 mm, 32-lead LFCSP
APPLICATIONS
Telecommunications and networking systems
Medical imaging systems
Base station power
Set-top boxes
Printers
DDR termination
TYPICAL APPLICATION CIRCUIT
PV IN
GND
EN1
EN2
BST2
SW2
DH2
CSL2
DL2
PGND2
FB2
COMP2
FREQ
LDOSD
SYNC
TRK1
TRK2
VREG
BST1
DH1
SW1
CSL1
DL1
PGND1
FB1
COMP1
ADP1823
180µF
180µF
3900pF
390pF
IRFR3709Z IRFR3709Z
IRLR7807Z
IRLR7807Z
1.2V,
6A
1.8V,
8A
390pF
3900pF
IN = 12V
560µF560µF
2.2µH 2.2µH
1µF
0.47µF 0.47µF
4.53kΩ
4.53kΩ
1kΩ
2kΩ
2kΩ
2kΩ
2kΩ2kΩ
05936-001
Figure 1.
GENERAL DESCRIPTION
The ADP1823 is a versatile, dual, interleaved, synchronous,
PWM buck controller that generates two independent output
rails from an input of 3.7 V to 20 V, with a power input voltage
that ranges from 1 V to 24 V. Each controller can be configured
to provide output voltages from 0.6 V to 85% of the input voltage
and is sized to handle large MOSFETs for point-of-load regulators.
The two channels operate 180° out of phase, reducing stress on
the input capacitor and allowing smaller, low cost components.
The ADP1823 is ideal for a wide range of high power applications,
such as DSP and processor core I/O power, and general-purpose
power in telecommunications, medical imaging, PCs, gaming,
and industrial applications.
The ADP1823 operates at a pin-selectable, fixed switching
frequency of either 300 kHz or 600 kHz, minimizing external
component size and cost. For noise sensitive applications, it can
also be synchronized to an external clock to achieve switching
frequencies between 300 kHz and 1 MHz. The ADP1823 includes
soft start protection to prevent inrush current from the input
supply during startup, reverse current protection during soft
start for precharged outputs, as well as a unique adjustable
lossless current-limit scheme using external MOSFET sensing.
For applications requiring power supply sequencing, the
ADP1823 also provides tracking inputs that allow the output
voltages to track during startup, shutdown, and faults. This
feature can also be used to implement DDR memory bus
termination.
The ADP1823 is specified over the −40°C to +125°C junction
temperature range and is available in a 32-lead LFCSP.
ADP1823
Rev. D | Page 2 of 32
TABLE OF CONTENTS
Features .............................................................................................. 1
Applications....................................................................................... 1
Typical Application Circuit ............................................................. 1
General Description ......................................................................... 1
Revision History ............................................................................... 2
Specifications..................................................................................... 3
Absolute Maximum Ratings............................................................ 5
ESD Caution.................................................................................. 5
Functional Block Diagram .............................................................. 6
Pin Configuration and Function Descriptions............................. 7
Typical Performance Characteristics ............................................. 9
Theory of Operation ...................................................................... 13
Input Power ................................................................................. 13
Start-Up Logic............................................................................. 13
Internal Linear Regulator .......................................................... 13
Oscillator and Synchronization................................................ 13
Error Amplifier........................................................................... 14
Soft Start ...................................................................................... 14
Power OK Indicator ................................................................... 14
Tracking ....................................................................................... 14
MOSFET Drivers........................................................................ 15
Current Limit.............................................................................. 15
Applications Information.............................................................. 16
Selecting the Input Capacitor ................................................... 16
Selecting the MOSFETs ............................................................. 17
Setting the Current Limit .......................................................... 18
Feedback Voltage Divider ......................................................... 18
Compensating the Voltage Mode Buck Regulator................. 19
Soft Start ...................................................................................... 22
Voltage Tracking ......................................................................... 22
Coincident Tracking .................................................................. 23
Ratiometric Tracking................................................................. 23
Thermal Considerations............................................................ 24
PCB Layout Guidelines.................................................................. 25
LFCSP Considerations............................................................... 26
Application Circuits ....................................................................... 27
Outline Dimensions ....................................................................... 29
Ordering Guide .......................................................................... 29
REVISION HISTORY
10/07—Rev. C to Rev D
Changes to Table 1............................................................................ 3
Changes to Equation 33 and Type III Compensator Section ... 21
7/07—Rev. B to Rev C
Changes to Figure 34...................................................................... 27
5/07—Rev. A to Rev. B
Changes to Features Section............................................................ 1
Changes to General Description Section ...................................... 1
Changes to Power Supply and Logic Thresholds Sections.......... 3
Changes to Absolute Maximum Ratings Section......................... 5
Changes to Figure 17...................................................................... 11
Changes to Theory of Operation Section.................................... 13
Changes to Current Limit Section................................................ 15
Changes to Setting the Current Limit Section............................ 18
Changes to Compensating the Voltage Mode Buck
Regulator Section............................................................................ 19
Inserted Figure 25........................................................................... 19
Deleted Table 4................................................................................ 27
Changes to Application Circuits Section..................................... 27
Changes to Figure 34...................................................................... 27
11/06—Rev. 0 to Rev. A
Changes to Features and Applications Sections ............................1
Changes to Specifications Section...................................................3
Changes to Absolute Maximum Ratings Section..........................5
Replaced Theory of Operation Section ....................................... 13
Added Feedback Voltage Divider Section................................... 18
Changes to Ratiometric Tracking Section................................... 23
Replaced PCB Layout Guidelines Section................................... 25
Added Application Circuits Section ............................................ 29
Changes to Ordering Guide.......................................................... 31
4/06—Revision 0: Initial Version
ADP1823
Rev. D | Page 3 of 32
SPECIFICATIONS
IN = 12 V, ENx = FREQ = PV = VREG = 5 V, SYNC = GND, TJ = −40°C to +125°C, unless otherwise specified. All limits at temperature
extremes are guaranteed via correlation using standard statistical quality control (SQC). Typical values are at TA = 25°C.
Table 1.
Parameter Conditions Min Typ Max Unit
POWER SUPPLY
IN Input Voltage PV = VREG (using internal regulator) 5.5 20 V
IN = PV = VREG (not using internal regulator) 3.7 5.5 V
IN Quiescent Current Not switching, IVREG = 0 mA 1.5 3 mA
IN Shutdown Current EN1 = EN2 = GND 10 20 A
VREG Undervoltage Lockout Threshold VREG rising 2.4 2.7 2.9 V
VREG Undervoltage Lockout Hysteresis 0.125 V
ERROR AMPLIFIER
FB1, FB2 Regulation Voltage TA = 25°C, TRK1, TRK2 > 700 mV 597 600 603 mV
T
J = 0°C to 85°C, TRK1, TRK2 > 700 mV 591 609 mV
T
J = −40°C to +125°C, TRK1, TRK2 > 700 mV 588 612 mV
T
J = 0°C to 70°C, TRK1, TRK2 > 700 mV 595 605 mV
FB1, FB2 Input Bias Current 100 nA
Open-Loop Voltage Gain 70 dB
Gain-Bandwidth Product 20 MHz
COMP1, COMP2 Sink Current 600 A
COMP1, COMP2 Source Current 120 A
COMP1, COMP2 Clamp High Voltage 2.4 V
COMP1, COMP2 Clamp Low Voltage 0.75 V
LINEAR REGULATOR
VREG Output Voltage TA = 25°C, IVREG = 20 mA 4.85 5.0 5.15 V
IN = 7 V to 20 V, IVREG = 0 mA to 100 mA,
TA = −40°C to +85°C
4.75 5.0 5.25 V
VREG Load Regulation IVREG = 0 mA to 100 mA, IN = 12 V −40 mV
VREG Line Regulation IN = 7 V to 20 V, IVREG = 20 mA 1 mV
VREG Current Limit VREG = 4 V 220 mA
VREG Short-Circuit Current VREG < 0.5 V 50 140 200 mA
IN to VREG Dropout Voltage IVREG = 100 mA, IN < 5 V 0.7 1.4 V
VREG Minimum Output Capacitance 1 F
PWM CONTROLLER
PWM Ramp Voltage Peak SYNC = GND 1.3 V
DH1, DH2 Maximum Duty Cycle FREQ = GND (300 kHz) 85 90 %
DH1, DH2 Minimum Duty Cycle FREQ = GND (300 kHz) 1 3 %
SOFT START
SS1, SS2 Pull-Up Resistance SS1, SS2 = GND 90 kΩ
SS1, SS2 Pull-Down Resistance SS1, SS2 = 0.6 V 6 kΩ
SS1, SS2 to FB1, FB2 Offset Voltage SS1, SS2 = 0 mV to 500 mV −45 mV
SS1, SS2 Pull-Up Voltage 0.8 V
TRACKING
TRK1, TRK2 Common-Mode Input Voltage Range 0 600 mV
TRK1, TRK2 to FB1, FB2 Offset Voltage TRK1, TRK2 = 0 mV to 500 mV −5 +5 mV
TRK1, TRK2 Input Bias Current 100 nA
ADP1823
Rev. D | Page 4 of 32
Parameter Conditions Min Typ Max Unit
OSCILLATOR
Oscillator Frequency SYNC = FREQ = GND (fSW = fOSC) 240 300 370 kHz
SYNC = GND, FREQ = VREG (fSW = fOSC) 480 600 720 kHz
SYNC Synchronization Range1FREQ = GND, SYNC = 600 kHz to 1.2 MHz (fSW = fSYNC/2) 300 600 kHz
FREQ = VREG, SYNC = 1.2 MHz to 2 MHz (fSW = fSYNC/2) 600 1000 kHz
SYNC Minimum Input Pulse Width 200 ns
CURRENT SENSE
CSL1, CSL2 Threshold Voltage Relative to PGND −30 0 +30 mV
CSL1, CSL2 Output Current CSL1, CSL2 = PGND 44 50 56 A
Current Sense Blanking Period 100 ns
GATE DRIVERS
DH1, DH2 Rise Time CDH = 3 nF, VBST − VSW = 5 V 15 ns
DH1, DH2 Fall Time CDH = 3 nF, VBST − VSW = 5 V 10 ns
DL1, DL2 Rise Time CDL = 3 nF 15 ns
DL1, DL2 Fall Time CDL = 3 nF 10 ns
DH to DL, DL to DH Dead Time 40 ns
LOGIC THRESHOLDS
SYNC, FREQ, LDOSD Input High Voltage 2.2 V
SYNC, FREQ, LDOSD Input Low Voltage 0.4 V
SYNC, FREQ Input Leakage Current SYNC, FREQ = 0 V to 5.5 V 1 A
LDOSD Pull-Down Resistance 100 kΩ
EN1, EN2 Input High Voltage IN = 3.7 V to 20 V 2.0 V
EN1, EN2 Input Low Voltage IN = 3.7 V to 20 V 0.8 V
EN1, EN2 Current Source EN1, EN2 = 0 V to 3.0 V −0.05 −0.6 −1.5 A
EN1, EN2 Input Impedance to 5 V Zener EN1, EN2 = 5.5 V to 20 V 100 kΩ
THERMAL SHUTDOWN
Thermal Shutdown Threshold2 145
°C
Thermal Shutdown Hysteresis2 15
°C
POWER GOOD
FB1, UV2 Overvoltage Threshold VFB1, VUV2 rising 750 mV
FB1, UV2 Overvoltage Hysteresis 50 mV
FB1, UV2 Undervoltage Threshold VFB1, VUV2 rising 550 mV
FB1, UV2 Undervoltage Hysteresis 50 mV
POK1, POK2 Propagation Delay 8 s
POK1, POK2 Off Leakage Current VPOK1, VPOK2 = 5.5 V 1 A
POK1, POK2 Output Low Voltage IPOK1, IPOK2 = 10 mA 150 500 mV
UV2 Input Bias Current 10 100 nA
1 SYNC input frequency is 2× the single-channel switching frequency. The SYNC frequency is divided by 2, and the separate phases were used to clock the controllers.
2 Guaranteed by design and not subject to production test.
ADP1823
Rev. D | Page 5 of 32
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Rating
IN, EN1, EN2 −0.3 V to +20 V
BST1, BST2 −0.3 V to +30 V
BST1, BST2 to SW1, SW2 −0.3 V to +6 V
CSL1, CSL2 −1 V to +30 V
SW1, SW2 −2 V to +30 V
DH1 SW1 − 0.3 V to BST1 + 0.3 V
DH2 SW2 − 0.3 V to BST2 + 0.3 V
DL1, DL2 to PGND −0.3 V to PV + 0.3 V
PGND to GND ±2 V
LDOSD, SYNC, FREQ, COMP1,
COMP2, SS1, SS2, FB1, FB2, VREG,
PV, POK1, POK2, TRK1, TRK2
−0.3 V to +6 V
θJA 4-Layer
(JEDEC Standard Board)1, 2
45°C/W
Operating Ambient Temperature −40°C < TA < +85°C
Operating Junction Temperature3−55°C < TJ < +125°C
Storage Temperature Range −65°C to +150°C
1 Measured with exposed pad attached to PCB.
2 Junction-to-ambient thermal resistance (θJA) of the package is based on
modeling and calculation using a 4-layer board. The junction-to-ambient
thermal resistance is application and board-layout dependent. In applications
where high maximum power dissipation exists, attention to thermal
dissipation issues in board design is required. For more information, refer
to Application Note AN-772, A Design and Manufacturing Guide for the Lead
Frame Chip Scale Package (LFCSP).
3 In applications where high power dissipation and poor package thermal
resistance are present, the maximum ambient temperature may have to be
derated. Maximum ambient temperature (TA_MAX) is dependent on the
maximum operating junction temperature (TJ_MAX_OP = 125oC), the maximum
power dissipation of the device in the application (PD_MAX), and the junction-
to-ambient thermal resistance of the part/package in the application (θJA),
and is given by: TA_MAX = TJ_MAX_OP − (θJA × PD_MAX).
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
ADP1823
Rev. D | Page 6 of 32
FUNCTIONAL BLOCK DIAGRAM
IN
UVLO
LOGIC
SYNC
FREQ
LINEAR REG
REF
PV
LDOSD
VREG
VREG
QS
R
PWM
PV
GND
+
–
+
–
+
–
+
+
+
–
+
–
+
–
+
–
+
–
+
–
+
+
+
–
VREG
THERMAL
SHUTDOWN
0.75V
0.55V
0.6V
0.8V
ILIM2
CK1
FAULT2FAULT1
50µA
Q
QS
R
PWM
Q
ILIM1
CK1
RAMP1
CK2
RAMP2
OSCILLATOR
PHASE 1 = 0°
PHASE 2 = 180°
BST1
DH1
SW1
DL1
PGND1
CSL1
POK1
BST2
DH2
SW2
DL2
PGND2
CSL2
POK2
VREG
EN1
EN2
COMP1
FB1
TRK1
SS1
COMP2
FB2
TRK2
UV2
SS2
0.6V
0.8V
FAULT1
0.6V
0.8V
FAULT2
RAMP1
0.75V
0.55V
RAMP2
0.75V
0.55V
VREG
ILIM2
50µA
CK2
ADP1823
05936-002
BOTTOM PADDLE
OF LFCSP
Figure 2.
ADP1823
Rev. D | Page 7 of 32
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
PIN 1
INDICATOR
1FB1
2SYNC
3FREQ
4GND
5UV2
6FB2
7COMP2
8TRK2
24 POK1
23 BST1
22 DH1
21 SW1
20 CSL1
19 PGND1
18 DL1
17 PV
9
SS2
10
POK2
11
BST2
12
DH2
13
SW2
14
CSL2
15
PGND2
16
DL2
32 COMP1
31 TRK1
30 SS1
29 VREG
28 IN
27 LDOSD
26 EN2
25 EN1
TOP VIEW
(Not to Scale)
ADP1823
05936-003
Figure 3. Pin Configuration
Table 3. Pin Function Descriptions
Pin No. Mnemonic Description
1 FB1
Feedback Voltage Input for Channel 1. Connect a resistor divider from the buck regulator output to GND and
tie the tap to FB1 to set the output voltage.
2 SYNC
Frequency Synchronization Input. Accepts external signal between 600 kHz and 1.2 MHz or between 1.2 MHz
and 2 MHz depending on whether FREQ is low or high, respectively. Connect SYNC to ground if not used.
3 FREQ Frequency Select Input. Low for 300 kHz or high for 600 kHz.
4 GND
Ground. Connect to a ground plane directly beneath the ADP1823. Tie the bottom of the feedback dividers to
this GND.
5 UV2
Input to the POK2 Undervoltage and Overvoltage Comparators. For the default thresholds, connect UV2
directly to FB2. For some tracking applications, connect UV2 to an extra tap on the FB2 voltage divider string.
6 FB2
Feedback Voltage Input for Channel 2. Connect a resistor divider from the buck regulator output to GND and
tie the tap to FB2 to set the output voltage.
7 COMP2 Error Amplifier Output for Channel 2. Connect an RC network from COMP2 to FB2 to compensate Channel 2.
8 TRK2
Tracking Input for Channel 2. To track a master voltage, drive TRK2 from a voltage divider to the master
voltage. If the tracking function is not used, connect TRK2 to VREG.
9 SS2 Soft Start Control Input. Connect a capacitor from SS2 to GND to set the soft start period.
10 POK2
Open-Drain Power OK Output for Channel 2. Sinks current when UV2 is out of regulation. Connect a pull-up
resistor from POK2 to VREG.
11 BST2
Boost Capacitor Input for Channel 2. Powers the high-side gate driver, DH2. Connect a 0.22 F to 0.47 F
ceramic capacitor from BST2 to SW2 and a Schottky diode from PV to BST2.
12 DH2 High-Side (Switch) Gate Driver Output for Channel 2.
13 SW2 Switch Node Connection for Channel 2.
14 CSL2
Current Sense Comparator Inverting Input for Channel 2. Connect a resistor between CSL2 and SW2 to set
the current-limit offset.
15 PGND2 Ground for Channel 2 Gate Driver. Connect to a ground plane directly beneath the ADP1823.
16 DL2 Low-Side (Synchronous Rectifier) Gate Driver Output for Channel 2.
17 PV Positive Input Voltage for Gate Driver DL1 and Gate Driver DL2. Connect PV to VREG and bypass to ground
with a 1 µF capacitor.
18 DL1 Low-Side (Synchronous Rectifier) Gate Driver Output for Channel 1.
19 PGND1 Ground for Channel 1 Gate Driver. Connect to a ground plane directly beneath the ADP1823.
20 CSL1
Current Sense Comparator Inverting Input for Channel 1. Connect a resistor between CSL1 and SW1 to set
the current-limit offset.
21 SW1 Switch Node Connection for Channel 1.
22 DH1 High-Side (Switch) Gate Driver Output for Channel 1.
23 BST1
Boost Capacitor Input for Channel 1. Powers the high-side gate driver, DH1. Connect a 0.22 F to 0.47 F
ceramic capacitor from BST1 to SW1 and a Schottky diode from PV to BST1.
ADP1823
Rev. D | Page 8 of 32
Pin No. Mnemonic Description
24 POK1
Open-Drain Power OK Output for Channel 1. Sinks current when FB1 is out of regulation. Connect a pull-up
resistor from POK1 to VREG.
25 EN1
Enable Input for Channel 1. Drive EN1 high to turn on the Channel 1 controller, and drive it low to turn off
the Channel 1 controller. Enabling starts the internal LDO. Tie to IN for automatic startup.
26 EN2
Enable Input for Channel 2. Drive EN2 high to turn on the Channel 2 controller, and drive it low to turn off
the Channel 2 controller. Enabling starts the internal LDO. Tie to IN for automatic startup.
27 LDOSD
LDO Shut-Down Input. Only used to shut down the LDO in those applications where IN is tied directly to VREG.
Otherwise, connect LDOSD to GND or leave it open because it has an internal 100 kΩ pull-down resistor.
28 IN Input Supply to the Internal Linear Regulator. Drive IN with 5.5 V to 20 V to power the ADP1823 from the LDO.
For input voltages between 3.7 V and 5.5 V, tie IN to VREG and PV.
29 VREG
Output of the Internal Linear Regulator (LDO). The internal circuitry and gate drivers are powered from VREG.
Bypass VREG to the ground plane with a 1 F ceramic capacitor.
30 SS1 Soft Start Control Input. Connect a capacitor from SS1 to GND to set the soft start period.
31 TRK1
Tracking Input for Channel 1. To track a master voltage, drive TRK1 from a voltage divider to the master voltage.
If the tracking function is not used, connect TRK1 to VREG.
32 COMP1 Error Amplifier Output for Channel 1. Connect an RC network from COMP1 to FB1 to compensate Channel 1.
ADP1823
Rev. D | Page 9 of 32
LOAD CURRENT (A)
EFFICIENCY (%)
TYPICAL PERFORMANCE CHARACTERISTICS
95
70
02
0
90
85
80
75
51015
VIN = 5V
VIN = 20V
VIN = 15V
VIN = 12V
05936-004
LOAD CURRENT (A)
EFFICIENCY (%)
Figure 4. Efficiency vs. Load Current, VOUT = 1.8 V, 300 kHz Switching
95
70
02
0
90
85
80
75
51015
VOUT = 3.3V
VOUT = 1.8V
VOUT = 1.2V
05936-005
LOAD CURRENT (A)
EFFICIENCY (%)
Figure 5. Efficiency vs. Load Current, VIN = 12 V, 300 kHz Switching
94
78
02
0
05936-006
92
78
020
LOAD CURRENT (A)
EFFICIENCY (%)
SWITCHING FREQUENCY = 300kHz
92
90
88
86
84
82
80
51015
SWITCHING FREQUENCY = 300kHz
SWITCHING FREQUENCY = 600kHz
Figure 6. Efficiency vs. Load Current, VIN = 5 V, VOUT = 1.8 V
90
88
86
84
82
80
SWITCHING FREQUENCY = 600kHz
51015
05936-007
4.980
4.960
–40 85
TEMPERATURE (°C)
VREG VOLTAGE (V)
Figure 7. Efficiency vs. Load Current, VIN = 12 V, VOUT = 1.8 V
4.975
4.970
4.965
–15 10 35 60
05936-008
4.970
4.950
520
INPUT VOLTAGE (V)
VREG (V)
Figure 8. VREG Voltage vs. Temperature
4.968
4.966
4.964
4.962
4.960
4.958
4.956
4.954
4.952
8111417
05936-009
Figure 9. VREG vs. Input Voltage, 10 mA Load
ADP1823
Rev. D | Page 10 of 32
4.960
4.940
0 100
LOAD CURRENT (mA)
VREG (V)
4.956
4.952
4.948
4.944
20 40 60 80
05936-010
Figure 10. VREG vs. Load Current, VIN = 12 V
0
0 50 100 150 200 250
LOAD CURRENT (mA)
VREG OUTPUT (V)
1
2
4
5
3
05936-011
Figure 11. VREG Current-Limit Foldback
200ns/DIV
VREG, AC-COUPLED, 1V/DIV
SW1 PIN, V
OUT
= 1.8V, 10V/DIV
SW2 PIN, V
OUT
= 1.2V, 10V/DIV
T
05936-012
Figure 12. VREG Output During Normal Operation
0.6010
0.5980
–40 85
TEMPERATURE (°C)
FEEDBACK VOLTAGE (V)
0.6005
0.6000
0.5995
0.5990
0.5985
–15 10 35 60
05936-013
Figure 13. Feedback Voltage vs. Temperature, VIN = 12 V
330
260
–40 85
TEMPERATURE (°C)
FREQUENCY (Hz)
320
310
300
290
280
270
–15 10 35 60
05936-014
Figure 14. Switching Frequency vs. Temperature, VIN = 12 V
5
0
22
SUPPLY VOLTAGE (V)
SUPPLY CURRENT (mA)
0
4
3
2
1
5 8 11 14 17
05936-015
Figure 15. Supply Current vs. Supply Voltage
ADP1823
Rev. D | Page 11 of 32
LOAD OFF LOAD OFF
100µs/DIV
LOAD ON
V
OUT1
, AC-COUPLED,
100mV/DIV
T
0
5936-016
Figure 16. 1.5 A to 15 A Load Transient Response, VIN = 12 V
T
05936-017
SS1, 0.5V/DIV
V
OUT1
, 0.5V/DIV
SHORT CIRCUIT REMOVED
SHORT CIRCUIT APPLIED
INPUT CURRENT, 0.2A/DIV
4ms/DIV
Figure 17. Output Short-Circuit Response
T
400ns/DIV
SWITCH NODE CHANNEL 2
SWITCH NODE
CHANNEL 1
05936-018
Figure 18. Out-of-Phase Switching, Internal Oscillator
400ns/DIV
SW PIN, CHANNEL 2
SW PIN, CHANNEL 1
EXTERNAL CLOCK, FREQUENCY = 1MHz
T
05936-019
Figure 19. Out-of-Phase Switching, External 1 MHz Clock
200ns/DIV
SW PIN, CHANNEL 2
SW PIN, CHANNEL 1
EXTERNAL CLOCK, FREQUENCY = 2MHz
T
05936-020
Figure 20. Out-of-Phase Switching, External 2 MHz Clock
V
OUT1
, 2V/DIV
EN1, 5V/DIV
10ms/DIV
V
IN
= 12V
T
05936-021
Figure 21. Enable Pin Response, VIN = 12 V
ADP1823
Rev. D | Page 12 of 32
V
IN
, 5V/DIV
SOFT START, 1V/DIV
4ms/DIV
T
V
OUT
, 2V/DIV
05936-022
Figure 22. Power-On Response, EN Tied to VIN
FEEDBACK PIN
VOLTAGE, 200mV/DIV
TRACK PIN VOLTAGE,
200mV/DIV
20ms/DIV
T
05936-023
Figure 23. Output Voltage Tracking Response
40ms/DIV
V
OUT1
, 2V/DIV
V
OUT2
, 2V/DIV
EN2 PIN, 5V/DIV
EN1 = 5V
05936-024
Figure 24. Coincident Voltage Tracking Response
ADP1823
Rev. D | Page 13 of 32
THEORY OF OPERATION
The ADP1823 is a dual, synchronous, PWM buck controller
capable of generating output voltages down to 0.6 V and output
currents in the tens of amps. The switching of the regulators is
interleaved for reduced current ripple. It is ideal for a wide
range of applications, such as DSP and processor core I/O
supplies, general-purpose power in telecommunications,
medical imaging, gaming, PCs, set-top boxes, and industrial
controls. The ADP1823 controller operates directly from 3.7 V
to 20 V, and the power stage input voltage range is 1 V to 24 V,
which applies directly to the drain of the high-side external
power MOSFET. It includes fully integrated MOSFET gate
drivers and a linear regulator for internal and gate drive bias.
The ADP1823 operates at a fixed 300 kHz or 600 kHz switching
frequency. The ADP1823 can also be synchronized to an external
clock to switch at up to 1 MHz per channel. The ADP1823
includes soft start to prevent inrush current during startup, as
well as a unique adjustable lossless current limit.
The ADP1823 offers flexible tracking for startup and shutdown
sequencing. It is specified over the −40°C to +125°C temperature
range and is available in a space-saving, 5 mm × 5 mm,
32-lead LFCSP.
INPUT POWER
The ADP1823 is powered from the IN pin up to 20 V. The
internal low dropout linear regulator, VREG, regulates the IN
voltage down to 5 V. The control circuits, gate drivers, and
external boost capacitors operate from the LDO output. Tie
the PV pin to VREG and bypass VREG with a 1 μF or greater
capacitor.
The ADP1823 phase shifts the switching of the two step-down
converters by 180°, thereby reducing the input ripple current.
The phase shift reduces the size and cost of the input capacitors.
The input voltage should be bypassed with a capacitor close to
the high-side switch MOSFETs (see the Selecting the Input
Capacitor section). In addition, a minimum 0.1 µF ceramic
capacitor should be placed as close as possible to the IN pin.
The VREG output is sensed by the undervoltage lockout
(UVLO) circuit to be certain that enough voltage headroom
is available to run the controllers and gate drivers. As VREG
rises above about 2.7 V, the controllers are enabled. The IN
voltage is not directly monitored by UVLO. If the IN voltage is
insufficient to allow VREG to be above the UVLO threshold,
the controllers are disabled but the LDO continues to operate.
The LDO is enabled whenever either EN1 or EN2 is high, even
if VREG is below the UVLO threshold.
If the desired input voltage is between 3.7 V and 5.5 V, connect
IN directly to the VREG pin and the PV pin, and drive LDOSD
high to disable the internal regulator. The ADP1823 requires
that the voltage at VREG and PV be limited to no more than 5.5
V, which is the only application where the LDOSD pin is used.
Otherwise, it should be grounded or left open. LDOSD has an
internal 100 k pull-down resistor.
Although IN is limited to 20 V, the switching stage can run from
up to 24 V, and the BST pins can go to 30 V to support the gate
drive. Dissipation on the ADP1823 can be limited by running IN
from a low voltage rail while operating the switches from the high
voltage rail.
START-UP LOGIC
The ADP1823 features independent enable inputs for each
channel. Drive EN1 or EN2 high to enable their respective
controllers. The LDO starts when either channel is enabled.
When both controllers are disabled, the LDO is disabled and
the IN quiescent current drops to about 10 μA. For automatic
startup, connect EN1 and/or EN2 to IN. The enable pins are
20 V compliant, but they sink current through an internal
100 k resistor when the EN pin voltage exceeds about 5 V.
INTERNAL LINEAR REGULATOR
The internal linear regulator, VREG, is low dropout, which means
it can regulate its output voltage close to the input voltage.
VREG powers up the internal control and provides bias for
the gate drivers. It is guaranteed to have more than 100 mA of
output current capability, which is sufficient to handle the gate
drive requirements of typical logic threshold MOSFETs driven
at up to 1 MHz. VREG should always be bypassed with a 1 μF
or greater capacitor.
Because the LDO supplies the gate drive current, the output of
VREG is subject to sharp transient currents as the drivers switch
and the boost capacitors recharge during each switching cycle.
The LDO has been optimized to handle these transients without
overload faults. Due to the gate drive loading, using the VREG
output for other auxiliary system loads is not recommended.
The LDO includes a current limit well above the expected
maximum gate drive load. This current limit also includes a
short-circuit foldback to further limit the VREG current in the
event of a fault.
OSCILLATOR AND SYNCHRONIZATION
The ADP1823 internal oscillator can be set to either 300 kHz or
600 kHz. Drive the FREQ pin low for 300 kHz; drive it high for
600 kHz. The oscillator generates a start clock for each switching
phase and generates the internal ramp voltages for the PWM
modulation.
The SYNC input is used to synchronize the converter switching
frequency to an external signal. The SYNC input should be
driven with twice the desired switching frequency, as the SYNC
input is divided by 2, and the resulting phases are used to clock
the two channels alternately.
ADP1823
Rev. D | Page 14 of 32
If FREQ is driven low, the recommended SYNC input frequency
is between 600 kHz and 1.2 MHz. If FREQ is driven high, the
recommended SYNC frequency is between 1.2 MHz and
2 MHz. The FREQ setting should be carefully observed for
these SYNC frequency ranges, because the PWM voltage ramp
scales down from about 1.3 V based on the percentage of
frequency overdrive. Driving SYNC faster than recommended
for the FREQ setting results in a small ramp signal, which could
affect the signal-to-noise ratio and the modulator gain and
stability.
When an external clock is detected at the first SYNC edge, the
internal oscillator is reset and clock control shifts to SYNC.
The SYNC edges then trigger subsequent clocking of the PWM
outputs. The DH rising edges appear about 400 ns after the
corresponding SYNC edge, and the frequency is locked to the
external signal. Depending on the start-up conditions of Channel 1
and Channel 2, either Channel 1 or Channel 2 can be the first
channel synchronized to the rising edge of the SYNC clock. If
the external SYNC signal disappears during operation, the
ADP1823 reverts to its internal oscillator and experiences a
delay of no more than a single cycle of the internal oscillator.
ERROR AMPLIFIER
The ADP1823 error amplifiers are operational amplifiers. The
ADP1823 senses the output voltages through external resistor
dividers at the FB1 and FB2 pins. The FB pins are the inverting
inputs to the error amplifiers. The error amplifiers compare
these feedback voltages to the internal 0.6 V reference, and
the outputs of the error amplifiers appear at the COMP1 and
COMP2 pins. The COMP pin voltages then directly control
the duty cycle of each respective switching converter.
A series/parallel RC network is tied between the FB pins and
their respective COMP pins to provide the compensation for
the buck converter control loops. A detailed design procedure
for compensating the system is provided in the Compensating
the Voltage Mode Buck Regulator section.
The error amplifier outputs are clamped between a lower limit
of about 0.7 V and a higher limit of about 2.4 V. When the
COMP pins are low, the switching duty cycle goes to 0%, and
when the COMP pins are high, the switching duty cycle goes
to the maximum.
The SS and TRK pins are auxiliary positive inputs to the error
amplifiers. Whichever has the lowest voltage, SS, TRK, or the
internal 0.6 V reference controls the FB pin voltage and thus
the output. Therefore, if two or more of these inputs are close
to each other, a small offset is imposed on the error amplifier.
For example, if TRK approaches the 0.6 V reference, the FB
sees about 18 mV of negative offset at room temperature. For
this reason, the soft start pins have a built-in negative offset
and they charge to 0.8 V. If the TRK pins are not used, they
should be tied high to VREG.
SOFT START
The ADP1823 employs a programmable soft start that reduces
input current transients and prevents output overshoot. The SS1
and SS2 pins drive auxiliary positive inputs to their respective
error amplifiers, thus the voltage at these pins regulates the voltage
at their respective feedback control pins.
Program soft start by connecting capacitors from SS1 and SS2
to GND. When starting up, the capacitor charges from an
internal 90 kΩ resistor to 0.8 V. The regulator output voltage
rises with the voltage at its respective soft start pin, allowing the
output voltage to rise slowly, reducing inrush current. See the
Soft Start section in the Applications Information section for
more information.
When a controller is disabled or experiences a current fault, the
soft start capacitor discharges through an internal 6 k resistor,
so that at restart or recovery from fault, the output voltage soft
starts again.
POWER OK INDICATOR
The ADP1823 features open-drain, power OK outputs, POK1
and POK2, which sink current when their respective output
voltages drop, typically 8% below the nominal regulation
voltage. The POK pins also go low for overvoltage of typically
25%. Use this output as a logical power-good signal by connect-
ing pull-up resistors from POK1 and POK2 to VREG.
The POK1 comparator directly monitors FB1, and the threshold
is fixed at 550 mV for undervoltage and 750 mV for overvoltage.
However, the POK2 undervoltage and overvoltage comparator
input is connected to UV2 rather than FB2. For the default
thresholds at FB2, connect UV2 directly to FB2.
In a ratiometric tracking configuration, however, Channel 2 can
be configured to be a fraction of a master voltage, and thus FB2
is regulated to a voltage lower than the 0.6 V internal reference.
In this configuration, UV2 can be tied to a different tap on the
feedback divider, allowing a POK2 indication at an appropriate
output voltage threshold. See the Setting the Channel 2
Undervoltage Threshold for Ratiometric Tracking section.
TRACKING
The ADP1823 features tracking inputs, TRK1 and TRK2, which
make the output voltages track another master voltage. Voltage
tracking is especially useful in core and I/O voltage sequencing
applications where one output of the ADP1823 can be set to
track and not exceed the other, or in other multiple output
systems where specific sequencing is required.
The internal error amplifiers include three positive inputs, the
internal 0.6 V reference voltage and their respective SS and TRK
pins. The error amplifiers regulate the FB pins to the lowest of
the three inputs. To track a supply voltage, tie the TRK pin to a
resistor divider from the voltage to be tracked. See the Volt age
Tracking section.
ADP1823
Rev. D | Page 15 of 32
MOSFET DRIVERS
The DH1 and DH2 pins drive the high-side switch MOSFETs.
These boosted 5 V gate drivers are powered by bootstrap capacitor
circuits. This configuration allows the high-side, N-channel
MOSFET gate to be driven above the input voltage, allowing
full enhancement and a low voltage drop across the MOSFET.
The bootstrap capacitors are connected from the SW pins to
their respective BST pins. The bootstrap Schottky diodes from
the PV pin to the BST pins recharge the bootstrap capacitors every
time the SW nodes go low. Use a bootstrap capacitor value
greater than 100× the high-side MOSFET input capacitance.
In practice, the switch node can run up to 24 V of input voltage,
and the boost nodes can operate more than 5 V above this to
allow full gate drive. The IN pin can be run from 3.7 V to 20 V,
which can provide an advantage, for example, in the case of high
frequency operation from very high input voltage. Dissipation on
the ADP1823 can be limited by running IN from a lower voltage
rail while operating the switches from the high voltage rail.
The switching cycle is initiated by the internal clock signal. The
high-side MOSFET is turned on by the DH driver, and the SW
node goes high, pulling up on the inductor. When the internally
generated ramp signal crosses the COMP pin voltage, the switch
MOSFET is turned off and the low-side synchronous rectifier
MOSFET is turned on by the DL driver. Active break-before-
make circuitry, as well as a supplemental fixed dead time, are
used to prevent cross-conduction in the switches.
The DL1 and DL2 pins provide gate drive for the low-side
MOSFET synchronous rectifiers. Internal circuitry monitors
the external MOSFETs to ensure break-before-make switching
to prevent cross-conduction. An active dead time reduction
circuit reduces the break-before-make time of the switching to
limit the losses due to current flowing through the synchronous
rectifier body diode.
The PV pin provides power to the low-side drivers. It is limited
to 5.5 V maximum input and should have a local decoupling
capacitor.
The synchronous rectifiers are turned on for a minimum time
of about 200 ns on every switching cycle to sense the current.
This minimum on time and the nonoverlap dead times put a limit
on the maximum high-side switch duty cycle based on the selected
switching frequency. Typically, this is about 90% at 300 kHz
switching, and at 1 MHz switching, it reduces to about 70%
maximum duty cycle.
Because the two channels are 180° out of phase, if one is operating
around 50% duty cycle, it is common for it to jitter when the
other channel starts switching. The magnitude of the jitter depends
somewhat on layout, but it is difficult to avoid in practice.
When the ADP1823 is disabled, the drivers shut off the external
MOSFETs, so that the SW node becomes three-stated or
changes to high impedance.
CURRENT LIMIT
The ADP1823 employs a unique, programmable, cycle-by-cycle
lossless current-limit circuit that uses a small, ordinary, inexpensive
resistor to set the threshold. Every switching cycle, the synchro-
nous rectifier turns on for a minimum time and the voltage
drop across the MOSFET RDSON is measured during the off
cycle to determine whether the current is too high.
This measurement is done by an internal current-limit
comparator and an external current-limit set resistor. The
resistor is connected between the switch node (that is, the
drain of the rectifier MOSFET) and the CSL pin. The CSL pin,
which is the inverting input of the comparator, forces 50 A
through the resistor to create an offset voltage drop across it.
When the inductor current is flowing in the MOSFET rectifier,
its drain is forced below PGND by the voltage drop across its
RDSON. If the RDSON voltage drop exceeds the preset drop on the
external resistor, the inverting comparator input is similarly
forced below PGND and an overcurrent fault is flagged.
The normal transient ringing on the switch node is ignored for
100 ns after the synchronous rectifier turns on; therefore, the
overcurrent condition must also persist for 100 ns in order for a
fault to be flagged.
When an overcurrent event occurs, the overcurrent comparator
prevents switching cycles until the rectifier current has decayed
below the threshold. The overcurrent comparator is blanked
for the first 100 ns of the synchronous rectifier cycle to prevent
switch node ringing from falsely tripping the current limit. The
ADP1823 senses the current limit during the off cycle. When
the current-limit condition occurs, the ADP1823 resets the
internal clock until the overcurrent condition disappears. The
ADP1823 suppresses the start clock cycles until the overload
condition is removed. At the same time, the SS capacitor is
discharged through a 6 k resistor. The SS input is an auxiliary
positive input of the error amplifier, so it behaves like another
voltage reference. The lowest reference voltage wins.
Discharging the SS voltage causes the converter to use a lower
voltage reference when switching is allowed again. Therefore,
as switching cycles continue around the current limit, the output
looks roughly like a constant current source due to the rectifier
limit, and the output voltage droops as the load resistance
decreases. In the event of a short circuit, the short-circuit
output current is the current limit set by the RCL resistor and
is monitored cycle by cycle. When the overcurrent condition is
removed, operation resumes in soft start mode.
In the event of a short circuit, the ADP1823 also offers a
technique for implementing a current-limit foldback with the
use of an additional resistor. See the Setting the Current Limit
section for more information.
ADP1823
Rev. D | Page 16 of 32
APPLICATIONS INFORMATION
SELECTING THE INPUT CAPACITOR
The input current to a buck converter is a pulse waveform. It is
zero when the high-side switch is off and approximately equal
to the load current when it is on. The input capacitor carries the
input ripple current, allowing the input power source to supply
only the dc current. The input capacitor needs sufficient ripple
current rating to handle the input ripple and ESR that is low
enough to mitigate input voltage ripple. For the usual current
ranges for these converters, good practice is to use two parallel
capacitors placed close to the drains of the high-side switch
MOSFETs, one bulk capacitor of sufficiently high current rating
as calculated in Equation 1, along with a 10 F ceramic capacitor.
Select an input bulk capacitor based on its ripple current rating.
If both Channel 1 and Channel 2 maximum output load currents
are about the same, the input ripple current is less than half of
the higher of the output load currents. In this case, use an input
capacitor with a ripple current rating greater than half of the
highest load current.
2
L
RIPPLE
I
I> (1)
If the Output 1 and Output 2 load currents are significantly
different (if the smaller is less than 50% of the larger), the
procedure in Equation 1 yields a larger input capacitor than
required. In this case, the input capacitor can be chosen as in
the case of a single phase converter with only the higher load
current; therefore, first determine the duty cycle of the output
with the larger load current.
IN
OUT
V
V
D= (2)
In this case, the input capacitor ripple current is approximately
)1( DDII LRIPPLE −≈ (3)
where:
IL is the maximum inductor or load current for the channel.
D is the duty cycle.
Use this method to determine the input capacitor ripple current
rating for duty cycles between 20% and 80%.
For duty cycles less than 20% or greater than 80%, use an input
capacitor with a ripple current rating of IRIPPLE > 0.4 IL.
Selecting the Output LC Filter
The output LC filter attenuates the switching voltage, making
the output an almost dc voltage. The output LC filter characteris-
tics determine the residual output ripple voltage.
Choose an inductor value such that the inductor ripple current
is approximately 1/3 of the maximum dc output load current.
Using a larger value inductor results in a physical size larger
than is required, and using a smaller value results in increased
losses in the inductor and MOSFETs.
Choose the inductor value by
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
−
=
IN
OUT
SW
L
OUT
IN
V
V
fI
VV
L (4)
where:
L is the inductor value.
fSW is the switching frequency.
VOUT is the output voltage.
VIN is the input voltage.
ΔIL is the inductor ripple current, typically 1/3 of the maximum
dc load current.
Choose the output bulk capacitor to set the desired output voltage
ripple. The impedance of the output capacitor at the switching
frequency multiplied by the ripple current gives the output
voltage ripple. The impedance is made up of the capacitive
impedance plus the nonideal parasitic characteristics, the
equivalent series resistance (ESR), and the equivalent series
inductance (ESL). The output voltage ripple can be
approximated with
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛++Δ=Δ ESLf
Cf
ESRIV SW
OUT
SW
L
OUT 4
8
1 (5)
where:
ΔVOUT is the output ripple voltage.
ΔIL is the inductor ripple current.
ESR is the equivalent series resistance of the output capacitor
(or the parallel combination of ESR of all output capacitors).
ESL is the equivalent series inductance of the output capacitor
(or the parallel combination of ESL of all capacitors).
Note that the factors of 8 and 4 in Equation 5 would normally
be 2π for sinusoidal waveforms, but the ripple current waveform in
this application is triangular. Parallel combinations of different
types of capacitors, for example, a large aluminum electrolytic
in parallel with MLCCs, may give different results.
Usually, the impedance is dominated by ESR at the switching
frequency, as stated in the maximum ESR rating on the
capacitor data sheet, so this equation reduces to
ESRIV L
OUT
Δ
≅
Δ
(6)
Electrolytic capacitors have significant ESL also, on the order of
5 nH to 20 nH, depending on type, size, and geometry, and PCB
traces contribute some ESR and ESL as well. However, using the
maximum ESR rating from the capacitor data sheet usually
provides some margin such that measuring the ESL is not
usually required.
ADP1823
Rev. D | Page 17 of 32
In the case of output capacitors where the impedance of the
ESR and ESL are small at the switching frequency, for instance,
where the output capacitor is a bank of parallel MLCC
capacitors, the capacitive impedance dominates and the ripple
equation reduces to
SW
OUT
L
OUT fC
I
V8
≅ (7)
Make sure that the ripple current rating of the output capacitors
is greater than the maximum inductor ripple current.
During a load step transient on the output, the output capacitor
supplies the load until the control loop has a chance to ramp the
inductor current. This initial output voltage deviation due to a
change in load is dependent on the output capacitor characteristics.
Again, usually the capacitor ESR dominates this response, and
the VOUT in Equation 6 can be used with the load step current
value for IL.
SELECTING THE MOSFETs
The choice of MOSFET directly affects the dc-to-dc converter
performance. The MOSFET must have low on resistance (RDSON)
to reduce I2R losses and low gate charge to reduce switching losses.
In addition, the MOSFET must have low thermal resistance to
ensure that the power dissipated in the MOSFET does not result
in overheating.
The power switch, or high-side MOSFET, carries the load current
during the PWM on time, carries the transition loss of the
switching behavior, and requires gate charge drive to switch.
Typically, the smaller the MOSFET RDSON, the higher the gate
charge and vice versa. Therefore, it is important to choose a
high-side MOSFET that balances those two losses. The conduction
loss of the high-side MOSFET is determined by
IN
OUT
DSON
L
CV
V
RIP 2
≅ (8)
where:
PC is the conduction power loss.
RDSON is the MOSFET on resistance.
The gate charge losses are dissipated by the ADP1823 regulator
and gate drivers and affect the efficiency of the system. The gate
charge loss is approximated by
SWG
IN
GfQVP ≅ (9)
where:
PG is the gate charge power.
QG is the MOSFET total gate charge.
fSW is the converter switching frequency.
Making the conduction losses balance the gate charge losses
usually yields the most efficient choice.
Furthermore, the high-side MOSFET transition loss is
approximated by
(
)
2
SW
FRLIN
T
fttIV
P
+
≅ (10)
where tR and tF are the rise and fall times of the selected
MOSFET as stated in the MOSFET data sheet.
The total power dissipation of the high-side MOSFET is
the sum of the previous losses.
PD = PC + PG + PT (11)
where PD is the total high-side MOSFET power loss. This
dissipation heats the high-side MOSFET.
The conduction losses may need an adjustment to account
for the MOSFET RDSON variation with temperature. Note that
MOSFET RDSON increases with increasing temperature. The
MOSFET data sheet should list the thermal resistance of the
package, θJA, along with a normalized curve of the temperature
coefficient of RDSON. For the power dissipation estimated,
calculate the MOSFET junction temperature rise over the
ambient temperature of interest.
TJ = TA + θJA PD (12)
Next, calculate the new RDSON from the temperature coefficient
curve and the RDSON specification at 25°C. A typical value of the
temperature coefficient (TC) of RDSON is 0.004/°C; therefore, an
alternate method to calculate the MOSFET RDSON at a second
temperature, TJ, is
RDSON @ TJ = RDSON @ 25°C[1 + TC(TJ − 25°C)] (13)
Then the conduction losses can be recalculated and the procedure
iterated once or twice until the junction temperature calculations
are relatively consistent.
The synchronous rectifier, or low-side MOSFET, carries the
inductor current when the high-side MOSFET is off. For high
input voltage and low output voltage, the low-side MOSFET
carries the current most of the time, and therefore, to achieve
high efficiency it is critical to optimize the low-side MOSFET
for small on resistance. In cases where the power loss exceeds
the MOSFET rating, or lower resistance is required than is
available in a single MOSFET, connect multiple low-side
MOSFETs in parallel. The equation for low-side MOSFET
power loss is
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−≅
IN
OUT
DSON
L
LS V
V
RIP 1
2 (14)
where:
PLS is the low-side MOSFET on resistance.
RDSON is the parallel combination of the resistances of the low-
side MOSFETs.
Check the gate charge losses of the synchronous rectifier(s)
using the PG equation (Equation 9) to make sure they are
reasonable.
ADP1823
Rev. D | Page 18 of 32
SETTING THE CURRENT LIMIT
The current-limit comparator measures the voltage across the
low-side MOSFET to determine the load current.
The current limit is set through the current-limit resistor, RCL.
The current sense pins, CSL1 and CSL2, source 50 A through
their respective RCL. This current source creates an offset voltage of
RCL multiplied by the 50 A CSL current. When the drop across
the low-side MOSFET RDSON is equal to or greater than this
offset voltage, the ADP1823 flags a current-limit event.
Because the CSL current and the MOSFET RDSON vary over
process and temperature, the minimum current limit should be
set to ensure that the system can handle the maximum desired
load current. To do this, use the peak current in the inductor,
which is the desired current-limit level plus the ripple current,
the maximum RDSON of the MOSFET at its highest expected
temperature, and the minimum CSL current.
A44
)(MAX
DSON
LPK
CL
RI
R= (15)
where ILPK is the peak inductor current.
In addition, the ADP1823 offers a technique for implementing
a current-limit foldback in the event of a short circuit with the
use of an additional resistor, as shown in Figure 25. Resistor RLO
is largely responsible for setting the foldback current limit during a
short circuit, and RHI is mainly responsible for setting up the
normal current limit. RLO is lower than RHI. These current-limit
sense resistors can be calculated by
A44
)(
μ
=MAXDSON
PKFOLDBACK
LO
RI
R (16)
A44
)( μ−
=
LO
MAXDSON
LPK
OUT
HI
R
R
I
V
R (17)
where:
IPKFOLDBACK is the desired short-circuit peak inductor current limit.
ILPK is the peak inductor current limit during normal operation
(also used in Equation 15).
ADP1823
DH
DL
CSL
V
IN
M1
M2
R
LO
R
HI
L
C
OUT
V
OUT
05936-035
Figure 25. Short-Circuit Current Foldback Scheme
Because the buck converters are usually running fairly high
current, PCB layout and component placement may affect the
current-limit setting. An iteration of the RCL or RLO and RHI
values may be required for a particular board layout and
MOSFET selection. If alternate MOSFETs are substituted at
some point in production, these resistor values may also need
an iteration.
FEEDBACK VOLTAGE DIVIDER
The output regulation voltage is set through the feedback voltage
divider. The output voltage is reduced through the voltage divider
and drives the FB feedback input. The regulation threshold at
FB is 0.6 V. The maximum input bias current into FB is 100 nA.
For a 0.15% degradation in regulation voltage and with 100 nA
bias current, the low-side resistor, RBOT, needs to be less than 9 kΩ,
which results in 67 µA of divider current. For RBOT, use 1 k to
10 k. A larger value resistor can be used but results in a reduction
in output voltage accuracy due to the input bias current at the
FB pin, whereas lower values cause increased quiescent current
consumption. Choose RTOP to set the output voltage by using
the following equation:
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−
=
FB
FB
OUT
BOTTOP V
VV
RR (18)
where:
RTOP is the high-side voltage divider resistance.
RBOT is the low-side voltage divider resistance.
VOUT is the regulated output voltage.
VFB is the feedback regulation threshold, 0.6 V.
ADP1823
Rev. D | Page 19 of 32
COMPENSATING THE VOLTAGE MODE BUCK
REGULATOR
Assuming the LC filter design is complete, the feedback control
system can then be compensated. Good compensation is critical
to proper operation of the regulator. Calculate the quantities in
Equation 19 through Equation 47 to derive the compensation
values. The goal is to guarantee that the voltage gain of the buck
converter crosses unity at a slope that provides adequate phase
margin for stable operation. Additionally, at frequencies above
the crossover frequency, fCO, guaranteeing sufficient gain margin
and attenuation of switching noise are important secondary
goals. For initial practical designs, a good choice for the
crossover frequency is one tenth of the switching frequency.
First calculate
10
SW
CO
f
f= (19)
This gives sufficient frequency range to design a compensation
that attenuates switching artifacts, while also giving sufficient
control loop bandwidth to provide good transient response.
The output LC filter is a resonant network that inflicts two poles
upon the response at a frequency fLC, so next calculate
LCπ
fLC 2
1
= (20)
Generally, the LC corner frequency is about two orders of
magnitude below the switching frequency, and, therefore, about
one order of magnitude below crossover. To achieve sufficient
phase margin at crossover to guarantee stability, the design
must compensate for the two poles at the LC corner frequency
with two zeros to boost the system phase prior to crossover. The
two zeros require an additional pole or two above the crossover
frequency to guarantee adequate gain margin and attenuation of
switching noise at high frequencies.
Depending on component selection, one zero may already be
generated by the equivalent series resistance (ESR) of the output
capacitor. Calculate this zero corner frequency, fESR, as
OUT
ESR
ESR CRπ
f2
1
= (21)
Figure 26 shows a typical Bode plot of the LC filter by itself.
The gain of the LC filter at crossover can be linearly
approximated from Figure 26 as
AFILTER = ALC + AESR
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
×−
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
×−=
ESR
CO
LC
ESR
FILTER f
f
f
f
AlogdB20logdB40 (22)
If fESR ≈ fCO, add another 3 dB to account for the local difference
between the exact solution and the linear approximation.
0dB
GAIN
FREQUENCY
LC FILTER BODE PLOT
PHASE
f
LC
f
ESR
f
CO
f
SW
A
FILTER
–40dB/dec
Φ
FILTER
0°
–90°
–180°
–20dB/dec
05936-025
Figure 26. LC Filter Bode Plot
To compensate the control loop, the gain of the system must be
brought back up so that it is 0 dB at the desired crossover
frequency. Some gain is provided by the PWM modulation itself.
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
=
RAMP
IN
MOD V
V
Alog20 (23)
For systems using the internal oscillator, this becomes
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
=V
V
AIN
MOD 3.1
log20 (24)
Note that if the converter is being synchronized, the ramp
voltage, VRAMP, is lower than 1.3 V by the percentage of
frequency increase over the nominal setting of the FREQ pin.
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
=
SYNC
FREQ
RAMP f
f
V2
V3.1 (25)
The factor of 2 in the numerator takes into account that the
SYNC frequency is divided by 2 to generate the switching
frequency. For example, if the FREQ pin is set high for the
600 kHz range and a 2 MHz SYNC signal is applied, the ramp
voltage is 0.78 V. The gain of the modulator is increased by
4.4 dB in this example.
ADP1823
Rev. D | Page 20 of 32
The rest of the system gain is needed to reach 0 dB at crossover.
The total gain of the system, therefore, is given by
AT = AMOD + AFILTER + ACOMP (26)
where:
AMOD is the gain of the PWM modulator.
AFILTER is the gain of the LC filter including the effects of
the ESR zero.
ACOMP is the gain of the compensated error amplifier.
Additionally, the phase of the system must be brought back up
to guarantee stability. Note from the Bode plot of the filter that
the LC contributes −180° of phase shift. Additionally, because
the error amplifier is an integrator at low frequency, it contributes
an initial −90°. Therefore, before adding compensation or
accounting for the ESR zero, the system is already down −270°.
To avoid loop inversion at crossover, or −180° phase shift, a
good initial practical design is to require a phase margin of 60°,
which is therefore an overall phase loss of −120° from the initial
low frequency dc phase. The goal of the compensation is to
boost the phase back up from −270° to −120° at crossover.
Two common compensation schemes are used, which are
sometimes referred to as Type II or Type III compensation,
depending on whether the compensation design includes two
or three poles. (Dominant pole compensation, or single pole
compensation, is referred to as Type I compensation, but
unfortunately, it is not very useful for dealing successfully with
switching regulators.)
If the zero produced by the ESR of the output capacitor provides
sufficient phase boost at crossover, Type II compensation is
adequate. If the phase boost produced by the ESR of the output
capacitor is not sufficient, another zero is added to the compensa-
tion network, and thus Type III is used.
In Figure 27, the location of the ESR zero corner frequency
gives significantly different net phase at the crossover frequency.
Use the following guidelines for selecting between Type II and
Type III compensators:
If fESRZ ≤ fCO/2, use Type II compensation.
If fESRZ > fCO/2, use Type III compensation.
GAIN
FREQUENCY
PHASE
LC FILTER BODE PLOT
PHASE CONTRIBUTION AT CROSSOVER
OF VARIOUS ESR ZERO CORNERS
f
SW
f
CO
f
ESR3
f
ESR2
f
ESR1
0dB
f
LC
–40dB/dec
–20dB/dec
0°
–90°
–180°
Φ1
Φ2
Φ3
05936-026
Figure 27. LC Filter Bode Plot
The following equations were used for the calculation of the
compensation components as shown in Figure 28 and Figure 29:
I
Z
ZCR
fπ2
1
1= (27)
)(π2
1
2
FF
TOP
FF
ZRRC
f+
= (28)
HFI
HFI
Z
P
CC
CC
R
f
+
=
π2
1
1 (29)
FFFF
PCR
fπ2
1
2= (30)
where:
fZ1 is the zero produced in the Type II compensation.
fZ2 is the zero produced in the Type III compensation.
fP1 is the pole produced in the Type II compensation.
fP2 in the pole produced in the Type III compensation.
ADP1823
Rev. D | Page 21 of 32
Type II Compensator
G
(dB)
PHASE
–180°
–270°
f
Z
f
P
0V
VRAMP
C
HF
C
I
R
Z
R
TOP
R
BOT
FROM
V
OUT
VREF
EA
COMP TO PWM
0
5936-027
–1 SLOPE
–1 SLOPE
Figure 28. Type II Compensation
If the output capacitor ESR zero frequency is sufficiently low
(≤½ of the crossover frequency), use the ESR to stabilize the
regulator. In this case, use the circuit shown in Figure 28.
Calculate the compensation resistor, RZ, with the following
equation:
2
LC
IN
COESRRAMP
TOP
ZfV
ffVR
R= (31)
where:
fCO is chosen to be 1/10 of fSW.
VRAMP is 1.3 V.
Next choose the compensation capacitor to set the compensa-
tion zero, fZ1, to the lesser of ¼ of the crossover frequency or ½
of the LC resonant frequency.
I
Z
SWCO
ZCR
ff
fπ2
1
404
1=== (32)
or
I
Z
LC
ZCR
f
fπ2
1
2
1== (33)
Solving for CI in Equation 32 yields
SW
Z
IfR
Cπ
20
= (34)
Solving for CI in Equation 33 yields
LC
Z
IfR
Cπ
1
= (35)
Use the larger value of CI from Equation 34 or Equation 35.
Because of the finite output current drive of the error amplifier,
CI needs to be less than 10 nF. If it is larger than 10 nF, choose a
larger RTOP and recalculate RZ and CI until CI is less than 10 nF.
Next choose the high frequency pole fP1 to be half of fSW.
SW
Pff 2
1
1= (36)
Because CHF << CI, Equation 29 is simplified to
HF
Z
PCR
fπ2
1
1= (37)
Solving for CHF in Equation 36 and Equation 37 yields
Z
SW
HF Rf
Cπ
1
= (38)
Type III Compensator
0V
VRAMP
G
(dB)
PHASE
–270°
–90°
fZfP
C
HF
C
I
R
Z
C
FF
R
TOP
R
BOT
FROM
V
OUT
VREF
EA
COMP TO PWM
R
FF
0
5936-028
–1 SLOPE
–1 SLOPE
+1 SLOPE
Figure 29. Type III Compensation
If the output capacitor ESR zero frequency is greater than half
of the crossover frequency, use a Type III compensator as
shown in Figure 29. Set the poles and zeros as follows:
SW
PP fff 2
1
21 == (39)
I
Z
SWCO
ZZ CR
ff
ff π2
1
404
21 ==== (40)
or
I
Z
LC
ZZ CR
f
ff π
=== 2
1
2
2
1 (41)
Use the lower zero frequency from Equation 40 or Equation 41.
Calculate the compensator resistor, RZ.
2
1
LC
IN
CO
Z
RAMP
TOP
ZfV
ffVR
R= (42)
Next calculate CI.
1
π2
1
ZZ
IfR
C= (43)
ADP1823
Rev. D | Page 22 of 32
Because of the finite output current drive of the error amplifier,
CI needs to be less than 10 nF. If it is larger than 10 nF, choose a
larger RTOP and recalculate RZ and CI until CI is less than 10 nF.
Because CHF << CI, combining Equation 29 and Equation 39 yields
Z
SW
HF Rf
Cπ
1
= (44)
Next calculate the feedforward capacitor, CFF. Assuming RFF <<
RTOP, then Equation 28 is simplified to
TOP
FF
ZRC
fπ2
1
2= (45)
Solving CFF in Equation 45 yields
2
2
1
ZTOP
FF fR
Cπ
= (46)
where fZ2 is obtained from Equation 40 or Equation 41.
The feedforward resistor, RFF, can be calculated by combining
Equation 30 and Equation 39.
SW
FF
FF fC
Rπ
1
= (47)
Check that the calculated component values are reasonable. For
instance, capacitors smaller than about 10 pF should be avoided.
In addition, the ADP1823 error amplifier has finite output
current drive; therefore, RZ values less than 3 k and CI values
greater than 10 nF should be avoided. If necessary, recalculate the
compensation network with a different starting value of RTOP. If
RZ is too small and CI is too big, start with a larger value of RTOP.
This compensation technique should yield a good working
solution.
In general, aluminum electrolytic capacitors have high ESR;
therefore, a Type II compensation is adequate. However, if several
aluminum electrolytic capacitors are connected in parallel,
producing a low effective ESR, Type III compensation is needed. In
addition, ceramic capacitors have very low ESR, on the order of
a few milliohms; therefore, Type III compensation is needed for
ceramic output capacitors. Type III compensation offers better
performance than Type II in terms of more low frequency gain
and more phase margin and less high frequency gain at the
crossover frequency.
SOFT START
The ADP1823 uses an adjustable soft start to limit the output
voltage ramp-up period, thus limiting the input inrush current.
The soft start is set by selecting the capacitor, CSS, from SS1 and
SS2 to GND. The ADP1823 charges CSS to 0.8 V through an
internal 90 k resistor. The voltage on the soft start capacitor
while it is charging is
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−= SS
RC
t
CSS eV 1V8.0 (48)
The soft start period ends when the voltage on the soft start pin
reaches 0.6 V. Substituting 0.6 V for VSS and solving for the
number of RC time constants
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−= )(k90
1V8.0V6.0 SS
SS
C
t
e (49)
tSS = 1.386 RCSS (50)
Because R = 90 k,
CSS = tSS × 8 µF/s (51)
where tSS is the desired soft start time in seconds.
VOLTAGE TRACKING
The ADP1823 includes a tracking feature that prevents an
output voltage from exceeding a master voltage. This feature is
especially important when the ADP1823 is powering separate
power supply voltages on a single integrated circuit, such as the
core and I/O voltages of a DSP or microcontroller. In these
cases, improper sequencing can cause damage to the load.
The ADP1823 tracking input is an additional positive input to
the error amplifier. The feedback voltage is regulated to the lower of
the 0.6 V reference or the voltage at TRK; therefore, a lower
voltage on TRK limits the output voltage. This feature allows
implementation of two different types of tracking: coincident
tracking, where the output voltage is the same as the master
voltage until the master voltage reaches regulation, or ratiometric
tracking, where the output voltage is limited to a fraction of the
master voltage.
In all tracking configurations, the master voltage should be
higher than the slave voltage.
Note that the soft start time setting of the master voltage should
be longer than the soft start of the slave voltage. This forces the
rise time of the master voltage to be imposed on the slave voltage.
If the soft start setting of the slave voltage is longer, the slave
comes up more slowly and the tracking relationship is not seen
at the output. The slave channel should still have a soft start
capacitor to give a small but reasonable soft start time to protect
in case of a restart after a current-limit event.
COMP
FB
TRK
SS
MASTER
VOLTAGE
R
TRKT
R
TRKB
0.6V
DETAIL VIEW OF
ADP1823
ERROR
AMPLIFIER
V
OUT
R
TOP
R
BOT
05936-029
Figure 30. Voltage Tracking
ADP1823
Rev. D | Page 23 of 32
COINCIDENT TRACKING
The most common application is coincident tracking, used
in core vs. I/O voltage sequencing and similar applications.
Coincident tracking limits the slave output voltage to be the
same as the master voltage until it reaches regulation. Connect
the slave TRK input to a resistor divider from the master voltage
that is the same as the divider used on the slave FB pin. This
technique forces the slave voltage to be the same as the master
voltage.
For coincident tracking, use RTRKT = RTOP and RTRKB = RBBOT,
where RTOP and RBOT are the values chosen in the
section.
Compensating
the Voltage Mode Buck Regulator
TIME
SLAVE VOLTAGE
MASTER VOLTAGE
VOLTAGE
05936-030
Figure 31. Coincident Tracking
As the master voltage rises, the slave voltage rises identically.
Eventually, the slave voltage reaches its regulation voltage,
where the internal reference takes over the regulation while the
TRK input continues to increase and thus removes itself from
influencing the output voltage. To ensure that the output voltage
accuracy is not compromised by the TRK pin being too close in
voltage to the 0.6 V reference, make sure that the final value of
the master voltage is greater than the slave regulation voltage by
at least 10%, or 60 mV as seen at the FB node. The higher the
final value, the better the performance is. A difference of 60 mV
between TRK and the 0.6 V reference produces about 3 mV of
offset in the error amplifier, or 0.5%, at room temperature, while
100 mV between them produces only 0.6 mV or 0.1% offset.
RATIOMETRIC TRACKING
Ratiometric tracking limits the slave output voltage to a fraction
of the master voltage. For example, the termination voltage for
DDR memories, VTT, is set to half the VDD voltage.
TIME
SLAVE VOLTAGE
MASTER VOLTAGE
VOLTAGE
0
5936-031
Figure 32. Ratiometric Tracking
For ratiometric tracking, the simplest configuration is to tie the
TRK pin of the slave channel to the FB pin of the master channel.
This has the advantage of having the fewest components, but
the accuracy suffers as the TRK pin voltage becomes equal to
the internal reference voltage and an offset is imposed on the
error amplifier of about −18 mV at room temperature.
A more accurate solution is to provide a divider from the
master voltage that sets the TRK pin voltage to be something
lower than 0.6 V at regulation, for example, 0.5 V. The slave
channel can be viewed as having a 0.5 V external reference
supplied by the master voltage.
Once this is complete, the FB divider for the slave voltage is
designed as in the Compensating the Voltage Mode Buck
Regulator section, except to substitute the 0.5 V reference for
the VFB voltage. The ratio of the slave output voltage to the
master voltage is a function of the two dividers:
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛+
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛+
=
TRKB
TRKT
BOT
TOP
MASTER
OUT
R
R
R
R
V
V
1
1
(52)
Another option is to add another tap to the divider for the
master voltage. Split the RBOT resistor of the master voltage into
two pieces, with the new tap at 0.5 V when the master voltage is
in regulation. This technique saves one resistor, but be aware
that Type III compensation on the master voltage causes the
feedforward signal of the master voltage to appear at the TRK
input of the slave channel.
By selecting the resistor values in the divider carefully, Equation 52
shows that the slave voltage output can be made to have a faster
ramp rate than that of the master voltage by setting the TRK
voltage at the slave larger than 0.6 V and RTRKB greater than
RTRKT. Make sure that the master SS period is long enough (that
is, sufficiently large SS capacitor) such that the input inrush
current does not run into the current limit of the power supply
during startup.
ADP1823
Rev. D | Page 24 of 32
Setting the Channel 2 Undervoltage Threshold for
Ratiometric Tracking
THERMAL CONSIDERATIONS
The current required to drive the external MOSFETs comprises
the vast majority of the power dissipation of the ADP1823. The
on-chip LDO regulates down to 5 V, and this 5 V supplies the
drivers. The full gate drive current passes through the LDO and
is then dissipated in the gate drivers. The power dissipated on
the gate drivers on the ADP1823 is
If FB2 is regulated to a voltage lower than 0.6 V by configuring
TRK2 for ratiometric tracking, the Channel 2 undervoltage
threshold can be set appropriately by splitting the top resistor
in the voltage divider, as shown in Figure 33. RBOT is the same as
calculated for the compensation in Equation 52, and
RTOP = RA + RB (53)
PD = VIN fSW(QDH1 + QDL1 + QDH2 + QDL2) (57)
B
550mV
750mV
POK2
UV2
FB2
CHANNEL 2
OUTPUT
VOLTAGE
R
A
R
B
R
BOT
TO ERROR
AMPLIFIER
05936-032
where:
VIN is the voltage applied to IN.
fSW is the switching frequency.
Q numbers are the total gate charge specifications from the
selected MOSFET data sheets.
The power dissipation heats the ADP1823. As the switching
frequency, the input voltage, and the MOSFET size increase, the
power dissipation on the ADP1823 increases. Care must be taken
not to exceed the maximum junction temperature. To calculate
the junction temperature from the ambient temperature and
power dissipation
TJ = TA + PD θJA (58)
Figure 33. Setting the Channel 2 Undervoltage Threshold
The current in all the resistors is the same.
A
UV
OUT
B
FBUV
BOT
FB
R
VV
R
VV
R
V2
2
22
2−
=
−
= (54)
The thermal resistance, θJA, of the package is typically 40°C/W
depending on board layout, and the maximum specified junction
temperature is 125°C, which means that at a maximum ambient
temperature of 85°C without airflow, the maximum dissipation
allowed is about 1 W.
where:
VUV2 is 600 mV.
VFB2 is the feedback voltage value set during the ratiometric
tracking calculations.
VOUT2 is the Channel 2 output voltage.
A thermal shutdown protection circuit on the ADP1823 shuts
off the LDO and the controllers if the die temperature exceeds
approximately 145°C, but this is a gross fault protection only
and should not be relied upon for system reliability.
Solving for RA and RB, B
(
)
2
2
2
FB
UV
OUTA
BOT
AV
VV
RR −
= (55)
(
)
2
22
FB
FBUV
BOT
BV
VV
RR −
= (56)
ADP1823
Rev. D | Page 25 of 32
PCB LAYOUT GUIDELINES
In any switching converter, some circuit paths carry high dI/dt,
which can create spikes and noise. Other circuit paths are
sensitive to noise. Still others carry high dc current and can
produce significant IR voltage drops. The key to proper PCB
layout of a switching converter is to identify these critical paths
and arrange the components and copper area accordingly.
When designing PCB layouts, be sure to keep high current
loops small. In addition, keep compensation and feedback
components away from the switch nodes and their associated
components.
The following is a list of recommended layout practices for the
ADP1823, arranged by decreasing order of importance.
• The current waveform in the top and bottom FETs is a pulse
with very high dI/dt; therefore, the path to, through, and from
each individual FET should be as short as possible and the
two paths should be commoned as much as possible. In
designs that use a pair of D-Pak or SO-8 FETs on one side
of the PCB, it is best to counter-rotate the two so that the
switch node is on one side of the pair and the high-side
drain can be bypassed to the low-side source with a suitable
ceramic bypass capacitor, placed as close as possible to the
FETs to minimize inductance around this loop through the
FETs and capacitor. The recommended bypass ceramic
capacitor values range from 1 F to 22 F depending upon
the output current. This bypass capacitor is usually connected
to a larger value bulk filter capacitor and should be grounded to
the PGND plane.
• GND, the VREG bypass, the soft start capacitor, and the
bottom end of the output feedback divider resistors should
be tied to an (almost isolated) small AGND plane. All of
these connections should have connections from the pin
to the AGND plane that are as short as possible. No high
current or high dI/dt signals should be connected to
this AGND plane. The AGND area should be connected
through one wide trace to the negative terminal of the
output filter capacitors.
• The PGND pin handles high dI/dt gate drive current
returning from the source of the low-side MOSFET. The
voltage at this pin also establishes the 0 V reference for the
overcurrent limit protection (OCP) function and the CSL
pin. A small PGND plane should connect the PGND pin
and the PVCC bypass capacitor through a wide and direct
path to the source of the low-side MOSFET. The placement
of CIN is critical for controlling ground bounce. The negative
terminal of CIN needs to be placed very close to the source of
the low-side MOSFET.
• Avoid long traces or large copper areas at the FB and CSL
pins, which are low signal level inputs that are sensitive to
capacitive and inductive noise pickup. It is best to position
any series resistors and capacitors as close as possible to
these pins. Avoid running these traces close and parallel to
high dI/dt traces.
• The switch node is the noisiest place in the switcher circuit
with large ac and dc voltage and current. This node should
be wide to minimize resistive voltage drop. However, to
minimize the generation of capacitively coupled noise, the
total area should be small. Place the FETs and inductor all
close together on a small copper plane to minimize series
resistance and keep the copper area small.
• Gate drive traces (DH and DL) handle high dI/dt and,
therefore, they tend to produce noise and ringing. They
should be as short and direct as possible. If possible, avoid
using feedthrough vias in the gate drive traces. If vias are
needed, it is best to use two relatively large ones in parallel
to reduce the peak current density and the current in each
via. If the overall PCB layout is less than optimal, slowing
down the gate drive slightly can be very helpful to reduce
noise and ringing. It is occasionally helpful to place small
value resistors (such as 5 or 10 Ω) in series with the gate
leads, mainly DH traces to the high-side FET gates. These
can be populated with 0 resistors if resistance is not
needed. Note that the added gate resistance increases the
switching rise and fall times and that in turn increases the
switching power loss in the MOSFET.
• The negative terminal of output filter capacitors should be
tied closely to the source of the low-side FET. Doing this
helps to minimize voltage difference between GND and
PGND at the ADP1823.
• Generally, be sure that all traces are sized according to the
current to be handled as well as their sensitivity in the
circuit. Standard PCB layout guidelines mainly address
heating effects of current in a copper conductor. These are
completely valid, but they do not fully cover other concerns,
such as stray inductance or dc voltage drop. Any dc voltage
differential in connections between ADP1823 GND and the
converter power output ground can cause a significant
output voltage error, because it affects converter output
voltage according to the ratio with the 600 mV feedback
reference. For example, a 6 mV offset between ground on the
ADP1823 and the converter power output causes a 1% error
in the converter output voltage.
ADP1823
Rev. D | Page 26 of 32
LFCSP CONSIDERATIONS
The LFCSP has an exposed die paddle on the bottom that
efficiently conducts heat to the PCB. To achieve the optimum
performance from the LFCSP, give special consideration to the
layout of the PCB. Use the following layout guidelines for the
LFCSP:
• The pad pattern is given in Figure 36. The pad dimension
should be followed closely for reliable solder joints while
maintaining reasonable clearances to prevent solder
bridging.
• The thermal pad of the LFCSP provides a low thermal
impedance path to the PCB. Therefore, the PCB must be
properly designed to effectively conduct the heat away from
the package. This is achieved by adding thermal vias to the
PCB, which provide a thermal path to the inner or bottom
layers. See Figure 36 for the recommended via pattern. Note
that the via diameter is small, which prevents the solder
from flowing through the via and leaving voids in the
thermal pad solder joint.
Note that the thermal pad is attached to the die substrate;
therefore, the planes that the thermal pad is connected to
must be electrically isolated or connected to GND.
• The solder mask opening should be about 120 microns
(4.7 mils) larger than the pad size, resulting in a minimum
60 microns (2.4 mils) clearance between the pad and the
solder mask.
• The paste mask opening is typically designed to match the
pad size used on the peripheral pads of the LFCSP. This
technique should provide a reliable solder joint as long as
the stencil thickness is about 0.125 mm.
• The paste mask for the thermal pad needs to be designed for
the maximum coverage to effectively remove the heat from
the package. However, due to the presence of thermal vias
and the large size of the thermal pad, eliminating voids may
not be possible. In addition, if the solder paste coverage is
too large, solder joint defects may occur. Therefore, it is
recommended to use multiple small openings over a single
big opening in designing the paste mask. The recommended
paste mask pattern is given in Figure 36. This pattern results
in about 80% coverage, which should not degrade the
thermal performance of the package significantly.
• The recommended paste mask stencil thickness is 0.125 mm.
A laser cut stainless steel stencil with trapezoidal walls
should be used.
• A no-clean, Type 3 solder paste should be used for mounting
the LFCSP. In addition, a nitrogen purge during the reflow
process is recommended.
• The package manufacturer recommends that the reflow
temperature should not exceed 220°C and the time above
liquid is less than 75 seconds. The preheat ramp should be
3°C per second or lower. The actual temperature profile
depends on the board density; the assembly house must
determine what works best.
ADP1823
Rev. D | Page 27 of 32
APPLICATION CIRCUITS
The ADP1823 controller can be configured to regulate outputs
with loads of more than 20 A if the power components, such as
the inductor, MOSFETs, and the bulk capacitors, are chosen
carefully to meet the power requirement. The maximum load
and power dissipation are limited by the powertrain components.
Figure 1 shows a typical application circuit that can drive an
output load of 8 A.
Figure 34 shows an application circuit that can drive 20 A loads.
Note that two low-side MOSFETs are needed to deliver the 20 A
load. The bulk input and output capacitors used in this example
are Sanyo OS-CON capacitors, which have low ESR and high
current ripple rating. An alternative to the OS-CON capacitors
are the polymer aluminum capacitors that are available from
other manufacturers, such as United Chemi-Con. Aluminum
electrolytic capacitors, such as the Rubycon ZLG low-ESR series,
can also be paralleled up at the input or output to meet the
ripple current requirement. Because the aluminum electrolytic
capacitors have higher ESR and much larger variation in
capacitance over the operating temperature range, a larger bulk
input and output capacitance is needed to reduce the effective
ESR and suppress the current ripple. Figure 34 shows that the
polymer aluminum or the aluminum electrolytic capacitors can
be used at the outputs.
f
OSC
= 300kHz
M1, M4: IRLR7807Z
L1, L2: TOKO, FDA1254-1ROM
C
OUT1
: SANYO, 2R5SEPC820M
D1, D2: VISHAY, BAT54
M2, M3, M5, M6: IRFR3709Z
C
IN1
, C
IN2
: SANYO, 20SP180M
C
OUT2
: RUBYCON, 6.3ZLG1200M10×16
05936-033
1µF
C
OUT2
820µF
25V
×2
47kΩ
6.8nF
1.5nF
120nF
10kΩ
4.7nF
1µF C
OUT1
1200µF
6.3V
×3
10nF
200Ω
2kΩ
M2
2kΩ
1kΩ
390Ω2kΩ
M5
5600pF
1.2V, 20A
L2
1µH
L1
1µH
M4 M1
0.47µF
0.47µF
1µF
C
IN1
180µF
20V
1µF
C
IN2
180µF
20V
D1
D2
1µF
IN = 5.5
V
TO 20V
1.8V, 20A
PV IN
TRK1
TRK2
VREG
BST1
DH1
SW1
CSL1
DL1
PGND1
FB1
COMP1
EN1
EN2
BST2
DH2
SW2
CSL2
DL2
PGND2
FB2
COMP2
FREQ
LDOSD
SYNC
GND
AGND
PGND
ADP1823
2kΩ
2kΩ
M6 M3
Figure 34. Application Circuit with 20 A Output Loads
ADP1823
Rev. D | Page 28 of 32
The ADP1823 can also be configured to drive an output load of
less than 1 A. Figure 35 shows a typical application circuit that
drives 1.5 A and 3 A loads in all multilayer ceramic capacitor
(MLCC) solutions. Notice that the two MOSFETs used in this
example are dual-channel MOSFETs in a PowerPAK® SO-8
package, which reduces cost and saves layout space. An alternative
to using the dual-channel SO-8 package is using two single
MOSFETs in SOT-23 or TSOP-6 packages, which are low cost
and small in size.
1µF
C
OUT2
47µF
C
OUT3
100µF
6.65kΩ
1.5nF
120nF120nF
6.65kΩ
1.5nF
1µF 10µF C
OUT1
100µF
8.2nF
84.5Ω
2kΩ
M2
1.4kΩ
1kΩ
IN
84.5Ω1.33kΩ
M4
8.2nF
1.0V, 3A
L2
2.2µH
L1
2.5µH
M3 M1
0.22µF
0.22µF
1µF 10µF
×2
1µF
10µF
×2 D1
D2
1µF
IN = 3.7
V
TO 5V
1.8V, 1.5A
PV IN
TRK1
TRK2
VREG
BST1
DH1
SW1
CSL1
DL1
PGND1
FB1
COMP1
EN1
EN2
BST2
DH2
SW2
CSL2
DL2
PGND2
FB2
COMP2
FREQ
LDOSD
SYNC
GND
AGND
PGND
fOSC
= 600kHz
M1 TO M4: DUAL-CHANNEL SO-8 IRF7331
L2: TOKO, FDV0602-2R2M
C
OUT2
: MURATA, GRM31CR60J476M
L1: SUMIDA, CDRH5D28-2R5NC
C
OUT1
, C
OUT3
: MURATA, GRM31CR60J107M
D1, D2: VISHAY BAT54
ADP1823
05936-034
2kΩ
4.12kΩ
Figure 35. Application Circuit with All Multilayer Ceramic Capacitors (MLCC)
ADP1823
Rev. D | Page 29 of 32
OUTLINE DIMENSIONS
COMPLIANT TO JEDEC STANDARDS MO-220-VHHD-2
0.30
0.23
0.18
0.20 REF
0.80 MAX
0.65 TYP
0.05 MAX
0.02 NOM
12° MAX
1.00
0.85
0.80 SEATING
PLANE
COPLANARITY
0.08
1
32
8
9
25
24
16
17
0.50
0.40
0.30
3.50 REF
0.50
BSC
PIN 1
INDICATOR
TOP
VIEW
5.00
BSC SQ
4.75
BSC SQ
3.25
3.10 SQ
2.95
PIN 1
INDICATOR
0.60 MAX
0.60 MAX
0.25 MIN
EXPOSED
PAD
(BOTTOM VIEW)
Figure 36. 32-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
5 mm × 5 mm Body, Very Thin Quad
(CP-32-2)
Dimensions shown in millimeters
ORDERING GUIDE
Model Temperature Range1Package Description Package Option
ADP1823ACPZ-R72−40°C to +85°C 32-Lead Lead Frame Chip Scale Package [LFCSP_VQ] CP-32-2
ADP1823-EVAL Evaluation Board
1 Operating junction temperature is −40°C to +125°C.
2 Z = RoHS Compliant Part.
ADP1823
Rev. D | Page 30 of 32
NOTES
ADP1823
Rev. D | Page 31 of 32
NOTES
ADP1823
Rev. D | Page 32 of 32
NOTES
©2006–2007 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D05936-0-10/07(D)
