A6265KLPTR-T - Allegro Microsystems

Description

The A6265 is a DC-to-DC converter controller, providing a

programmable constant current output for driving high power

LEDs in series. Driving the LEDs in series ensures identical

currents and uniform brightness. For automotive applications,

optimum performance is achieved when driving between

2 and 10 LEDs at currents up to 1 A.

The A6265 provides a cost-effective solution using an external

logic-level MOSFET and minimum additional external

components. The maximum LED current is set with a single

external sense resistor and can be modified using a current

reference input. Direct PWM control is possible via the Enable

input, which also provides a shutdown mode.

This DC-DC converter can be configured as a ground-referenced

boost converter or as a supply-referenced boost converter

providing buck-boost capability. The buck-boost topology used

ensures that there is no leakage path through the LEDs when

in shutdown and no inrush current at power-up.

Integrated diagnostics and two fault outputs give indication of

undervoltage, chip overtemperature, output open circuit, LED

short circuit and LED undercurrent, and can be configured to

provide short to supply and short to ground protection for the

LED connections. A unique feature is the ability to detect one

or more shorted LEDs.

The device is provided in a 16-pin TSSOP package with exposed

thermal pad (suffix LP). It is lead (Pb) free, with 100% matte

tin leadframe plating.

A6265-DS, Rev. 5

Features and Benefits

 AEC Q100 Grade 0 Automotive Qualified

 Constant current LED drive

 6.5 to 50 V supply

 Boost or buck-boost modes

 Drives between 2 and 10 LEDs in series

 Programmable switching frequency 100 to 700 kHz

 Open LED overvoltage indication and protection

 Single and multiple LED short indication

 LED short to ground and supply protection

 PWM dimming control

 10 A shutdown current including LED leakage

Automotive High Current LED Controller

Package: 16-pin TSSOP with exposed

thermal pad (suffix LP)

Applications:

Typical Application Diagrams

Not to scale

A6265

Buck-Boost Mode

(Supply-referenced boost)

Boost Mode

 Automotive high power LED lighting systems

 Fog lights, reversing lights, daytime running lights

 Headlights

GND LA

Fault

Flags

VBAT 12 V or 24 V

Power net

(50 V max)

VREG

FF1

FF2

IREF

CKOUT

OSC

Enable

6265

VIN

GND LA

VBAT 12 V or 24 V (50 V max)

Fault

Flags

Power net

FF1

FF2

IREF

CKOUT

OSC

Enable

6265

VREG

VIN

Boost Mode

VBAT(min)

(V)

Maximum

Quantity of

LEDs

712

813

915

Vf of each LED = 3.5 V,

D(max) = 85%

Buck Boost Mode

VBAT(min)

(V)

Maximum

Quantity of

LEDs

710

811

913

Vf of each LED = 3.5 V,

D(max) = 85%

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

Absolute Maximum Ratings With respect to GND at TA = 25°C, unless otherwise specified

Characteristic Symbol Notes Rating Unit

Load Supply Voltage VIN –0.3 to 50 V

Pins FF1, EN –0.3 to 50 V

Pins FF2, CKOUT –0.3 to 6.5 V

Pin OSC –0.3 to 6.5 V

Pin SG –0.3 to 6.5 V

Pins LA, LN –0.3 to 50 V

Pin LF With respect to LA –6 to 6 V

Pin LP With respect to LN –6 to 6 V

Pin SP, SN –0.3 to 5 V

Pin VREG –0.3 to 7 V

Pin IREF –0.3 to 7 V

Junction Temperature TJ(max) 150 °C

Storage Temperature Range Tstg –55 to 150 °C

Operating Temperature Range TA Range K –40 to 150 °C

Selection Guide

Part Number Packing Package

A6265KLPTR-T 4000 pieces per 13-in. reel 16-pin TSSOP with exposed thermal pad

Thermal Characteristics may require derating at maximum conditions, see application information

Characteristic Symbol Test Conditions* Value Unit

Package Thermal Resistance

(Junction to Ambient) RJA

On 4-layer PCB based on JEDEC standard 34 ºC/W

On 2-layer PCB with 3.8 in.2 of copper area each side 43 ºC/W

Package Thermal Resistance

(Junction to Exposed Pad) RJP 2 ºC/W

*Additional thermal information available on the Allegro website

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

Pin-out Diagram

FF1

FF2

CKOUT

IREF

OSC

VIN

VREG

GND

PAD

Terminal List Table

Number Name Function

1 EN Enable chip

2 FF1 Fault flag

3 FF2 Fault flag

4 CKOUT Oscillator output, with phase shift

5 IREF Current reference

6 OSC Oscillator input/frequency set

7 SN Switch current sense –ve input

8 SP Switch current sense +ve input

9 SG Switch gate drive

10 GND Ground

11 VREG Internal regulator capacitor

12 VIN Main supply

13 LP Load current sense +ve input

14 LN Load current sense –ve input

15 LA LED string voltage sense

16 LF Reference LED voltage sense

– PAD Exposed thermal pad

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

Functional Block Diagrams

VIN

GND

VREG

Slope

Gen

OSC



Osc

FF1

VREG

Fault

Detect

LN LP LA

Shut-

down

Control

Logic

RSL

RSS

ROSC

VBAT

LED Open and

Short Detect

IREF

Temp

Monitor

IREF CKOUT

FF2

Buck-Boost

Boost

VIN

GND

Slope

Gen

OSC



Osc

FF1

VREG

Fault

Detect

LN LP LA

Shut-

down

Control

Logic RSS

ROSC



VBAT

LED Open and

Short Detect

Temp

Monitor

IREF

See Functional Description section.

CKOUT

FF2

RSL

IREF

VREG

AL AE AC AS



Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

Supply and Reference

VIN Functional Operating Range2 6.5 – 50 V

VIN Quiescent Current IINQ SG open circuit – – 8 mA

IINS EN = GND – 4 10 A

VREG Output Voltage VREG I

REG = 0 to 2 mA 4.75 5 5.25 V

Gate Output Drive

Turn-On Time trCLOAD = 1 nF, 20% to 80% – 35 – ns

Turn-Off Time tfCLOAD = 1 nF, 80% to 20% – 35 – ns

Maximum Duty Cycle D tON × fOSC 80 85 – %

Pull-Up On Resistance RDS(on)UP TJ = 25°C, IGHx = –100 mA – 1.7 – 

TJ = 150°C, IGHx = –100 mA – – 3.5 

Pull-Down On Resistance RDS(on)DN TJ = 25°C, IGLx = 100 mA – 1 – 

TJ = 150°C, IGLx = 100 mA – – 2 

Output High Voltage VSGH I

SG = –100 A VREG –

0.1 – V

REG V

Output Low Voltage VSGL I

SG = 100 A – – 0.1 V

Logic Inputs and Outputs

Fault Output (Open Drain) VOL I

OL = 1 mA, fault not asserted – – 0.4 V

Fault Output FF1 Sink Current1 I

OH(snk) 0.4 V < VO < 50 V, fault not asserted – 1.3 – mA

Fault Output FF1 Leakage Current1IOH1(lkg) VO = 12 V, fault asserted –1 – 1 A

Fault Output FF2 Leakage Current1 I

OH2(lkg) VO = 5 V, fault asserted –5 – 5 A

Input Low Voltage VIL – – 0.8 V

Input High Voltage VIH 2 – – V

Input Hysteresis VIhys 225 330 – mV

Enable Input Internal Clamp Voltage VENC – 8.4 – V

Enable Input Current Limit Resistor REN Between EN and internal clamp – 200 – k

Boost Mode Select Voltage VLNB Defined by VLN – – 1.2 V

Buck-Boost Mode Select Voltage VLNBB Defined by VLN 4.0 – – V

Disable Time tDIS fOSC = 350 kHz – 94 – ms

Oscillator

Oscillator Frequency fOSC

ROSC = 43 k– 500 – kHz

ROSC = 62 k315 350 385 kHz

ROSC = VREG – 350 – kHz

OSC Pin Voltage VOSC R

OSC = 62 k 1.15 1.2 1.25 V

CKOUT Output Delay tDC OSC input rise to CKOUT rise – 150 – ns

OSC Input Low Voltage VOIL – – 0.8 V

OSC Input High Voltage VOIH 3.5 – – V

OSC Input Hysteresis VOihys 300 600 – mV

ELECTRICAL CHARACTERISTICS1 Valid at TJ = –40°C to 150°C, VIN = 8 to 40 V; unless otherwise noted

Characteristics Symbol Test Conditions Min. Typ. Max. Unit

Continued on the next page…

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

ELECTRICAL CHARACTERISTICS1 (continued) Valid at TJ = –40°C to 150°C, VIN = 8 to 40 V; unless otherwise noted

Characteristics Symbol Test Conditions Min. Typ. Max. Unit

Oscillator (continued)

OSC Watchdog Period tOSWD Between successive rising edges 7 – – s

CKOUT Output High Voltage VCOH I

OH = –1 mA VREG – 1 – VREG V

CKOUT Output Low Voltage VCOL I

OL = 1 mA – – 0.4 V

LED Current Sense

Input Bias Current LN (BB mode)3 I

LN LP = LN = VIN – 6 – A

Input Bias Current LP (BB mode)3 I

LP LP = LN = VIN – 220 – A

Input Bias Current LN (B mode)3 I

LN LP = LN = 0 V – –2 – A

Input Bias Current LP (B mode)3 I

LP LP = LN = 0 V – –25 – A

Differential Input Voltage (Active) VIDL EN = High, VIDL = VLP – VLN – 100 – mV

Differential Input Voltage (PWM off) VIDLO EN = Low, VIDL = VLP – VLN – 8 – mV

Input Common-Mode Range (BB mode)3 VCMLH VLP = VLN VIN –V

IN + 1 V

Input Common-Mode Range (B mode)3 VCMLL VLP = VLN 0 – 1 V

Current Error EISL [(10 × ILED × RSL) – 1] × 100 –5 – 5 %

Switch Current Sense

Input Bias Current IBIASS SP = SN = 0 to 2 V –30 – – A

Maximum Differential Input Voltage4 V

IDS VIDS = VSP – VSN with D = 50% 330 410 490 mV

Input Source Current IINS V

IDS = 400 mV – 400 – A

Input Common-Mode Range VCMS VSP = VSN 0 – 2 V

Diagnostics and Protection

Fault Blank Timer5tFB Start-up – 3 – ms

VREG Undervoltage Turn-Off VREGUV Decreasing VREG 4.1 4.3 4.5 V

VREG Undervoltage Hysteresis VREGUVhys 130 150 160 mV

LED String Short Voltage VSCL 430 505 580 mV

Non-Reference LED Short Offset

Voltage VSCO 160 200 240 mV

Reference LED Short Offset Voltage VSCOR 430 505 580 mV

LED Open Voltage VOCL 5 5.5 6 V

LF Bias Current (BB mode)3 I

LF LF = LA = VIN + 1.7 V – 46 – A

LA Bias Current (BB mode)3 I

LA LF = LA = VIN + 1.7 V – 170 – A

LF Bias Current (B mode)3ILF LF = 1.7 V – 8 – A

LA Bias Current (B mode)3 I

LA LA = 1.7 V – 34 – A

LED Undercurrent Voltage Difference6 VUCL – 1 – mV

Open Fault Time-Out tOTO fOSC = 350 kHz – 94 – ms

Overtemperature Warning Threshold TJF Temperature increasing – 170 – ºC

Overtemperature Hysteresis TJhys Recovery = TJF – TJhys –15– ºC

1For input and output current specifications, negative current is defined as coming out of (sourcing) the specified device pin.

2Function is correct but parameters are not guaranteed below the general limit (8 V).

3BB mode = buck-boost (supply-referenced) mode, B mode = boost (ground-referenced) mode.

4Parameters guaranteed by design.

5Fault Blank timer not enabled for open-LED condition.

6Undercurrent when VSENSEL < VIDL– VUCL

, where VSENSEL is the voltage across the LED current sense resistor RSL.

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

Functional Description

The A6265 is a DC-DC converter controller that is designed to

drive series-connected high power LEDs in automotive applica-

tions. It provides programmable constant current output at load

voltages and currents limited only by the external components.

For automotive applications optimum performance is achieved

when driving between two and ten LEDs at currents up to 1 A.

The A6265 can be configured as a standard boost converter or

as a supply referenced boost converter. In the supply referenced

configuration the load voltage is the difference between the boost

voltage and the supply voltage. This difference can be greater

than, equal to, or less than the supply voltage, effectively provid-

ing a buck-boost capability. This configuration provides seam-

less, uninterrupted operation over the wide supply voltage range

possible in automotive applications and, because the output is ref-

erenced to the positive supply, there is no load current to ground.

This ensures that there is no leakage path through the LEDs when

in shutdown and no inrush current at power-up.

The A6265 integrates all necessary control elements to pro-

vide a cost-effective solution using a single external logic-level

MOSFET and minimum additional external passive components.

The LED current is set by selecting an appropriate value for the

sense resistor value and using the EN input to provide simple

on-off control or for PWM brightness control using a suitable

externally generated PWM signal. The LED current can be

reduced in a single step by reducing the voltage between the

IREF pin and GND to less than 1 V.

The pin functions and circuit operation are described in detail in

the following sections.

Pin Functions

VIN Supply to the control circuit. A bypass capacitor must be

connected between this pin and GND.

GND Ground reference connection. This pin should be connected

directly to the negative supply.

EN Logic input to enable operation. Can be used as direct PWM

input. Chip enters low power sleep mode when low for longer

than the disable time, tDIS.

FF1 Fault Flag output and isolation control. Open drain current

sink output, when high impedance indicates detection of a critical

circuit fault. An external pull-up resistor should be connected to a

suitable logic supply for simple logic fault flag operation or to the

source of the PMOS FET used to isolate the load from the supply.

Table 1 defines when FF1 is active. If FF1 is pulled low when

an output short fault is indicated then the output disable will be

overridden.

FF2 Fault Flag output. Open drain output, when high impedance

indicates detection of a circuit fault. An external pull-up resistor

should be connected to a suitable logic supply. If VREG is not

used, then the logic supply should not be pulled 300 mV above

VREG. Table 1 defines when FF2 is active. If FF2 is pulled low

when an open LED fault is indicated then the output disable will

be overridden.

OSC Resistor to ground to set the internal oscillator or clock

input from external oscillator. When connected to VREG the

oscillator runs at typically 350 kHz. Higher accuracy in the

frequency is possible by connecting a resistor from this pin to

ground or by driving this pin with an external precision oscillator.

CKOUT Logic output at the oscillator frequency with phase

shift. Used to drive succeeding controllers to interleave switching

instants.

IREF LED current reference modifier. A voltage input that can be

used to reduce the LED current sense voltage. When connected to

VREG, the current sense voltage, VIDL, and the value of the sense

resistor, RSL , define the maximum LED current.

SG Gate drive for external logic-level MOSFET low-side switch

that connects the inductor to ground.

SP, SN Sense amplifier connections for switch current limit

sense resistor, RSS .

LP Positive sense amplifier connection for LED current limit

sense resistor, RSL . This pin is also the bias supply for the LED

current sense amplifier.

LN Negative sense amplifier connection for LED current limit

sense resistor, RSL . The voltage at LN also determines whether

the boost or buck-boost mode is configured.

VREG Compensation capacitor for internal 5 V regulator.

LA Anode reference connection to LEDs. Using an external resis-

tor divider with the same ratio as the number of LEDs provides

a measurement of the voltage across all LEDs in the load. This

is compared to the voltage on the LF pin to provide shorted LED

detection. In addition, it is compared against voltage references to

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

provide open circuit or shorted LED string detection.

LF Single diode forward voltage reference input. Measures the

forward voltage of the first LED. This value is used as a reference

against the voltage on the LA pin to detect possible shorted LEDs

in the LED string.

Circuit Operation

Converter A constant frequency, current mode control scheme

is used to regulate the current through the LEDs. There are two

control loops within the regulator. The inner loop formed by the

amplifier, AS (see the Functional Block Diagram for AS, AC, AE,

and AL), comparator, AC, and the RS bistable, controls the induc-

tor current as measured through the switch by the switch sense

resistor, RSS .

The outer loop including the amplifier, AL, and the integrating

error amplifier, AE, controls the average LED current by provid-

ing a setpoint reference for the inner loop.

The LED current is measured by the LED sense resistor, RSL ,

and compared to the internal reference current to produce an

integrated error signal at the output of AE. This error signal sets

the average amount of energy required from the inductor by the

LEDs. The average inductor energy transferred to the LEDs is

defined by the average inductor current as determined by the

inner control loop.

The inner loop establishes the average inductor current by

controlling the peak switch current on a cycle-by-cycle basis.

Because the relationship between peak current and average cur-

rent is non-linear, depending on the duty cycle, the reference

level for the peak switch current is modified by a slope generator.

This compensation reduces the peak switch current measurement

by a small amount as the duty cycle increases (refer to figure 1).

The slope compensation also removes the instability inherent in a

fixed frequency current control scheme.

The control loops work together as follows: the switch current,

sensed by the switch current sense resistor, RSS , is compared

to the LED current error signal. As the LED current increases

the output of AE will reduce, reducing the peak switch current

and thus the current delivered to the LEDs. As the LED current

decreases the output of AE increases, increasing the peak switch

current and thus increasing the current delivered to the LEDs.

Under some conditions, especially when the LED current is set to

a low value, the energy required in the inductor may result in the

inductor current dropping to zero for part of each cycle. This is

known as discontinuous mode operation, and results in some low

frequency ripple. The average LED current, however, remains

regulated down to zero. In discontinuous mode, when the induc-

tor current drops to zero, the voltage at the drain of the external

MOSFET rings, due to the resonant LC circuit formed by the

inductor, and the switch and diode capacitance. This ringing is

low frequency and is not harmful.

Switch Current Limit The switch current is measured by the

switch sense resistor, RSS , and the switch sense amplifier, AS

(see the Functional Block Diagram). The input limit of the sense

amplifier, VIDS

, and the maximum switch current, ISMAX , define

the maximum value of the sense resistor as:

SS = VIDS / ISMAX (1)

This defines the maximum measurable value of the switch (and

inductor) current.

The maximum switch current is modulated by the on-time of the

switch. An internal slope compensation signal is subtracted from

the voltage sense signal to produce a peak sense voltage which

effectively defines the current limit. This signal is applied at a

rate of –50 mV / s starting with no contribution (t = 0 s) at the

beginning of each switching cycle. Figure 1 illustrates how the

peak sense voltage (typical values) changes over a period of 3 s.

For example, the maximum current (typical) through the switch

at t = 1.5 s (D = 50%) would be 410 mV/RSS

, however, if the

switch remained on for a further 1 s, the maximum current

through the switch would be 360 mV/RSS .

500

450

400

350

300

250

200

150

100

001

Period (s)

Peak Sense Votlage (mV)

Figure 1. Slope compensation for peak switch current control.

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

LED Current Level The LED current is determined by a

combination of the LED sense resistor, RSL , the LED current

threshold voltage, VIDL , and the voltage between the IREF pin

and GND ( VIREF

The 100% current level, when the IREF pin is connected to

VREG, is defined as:

ILED(max) = VIDL / RSL (2)

If VIREF is less than 1 V then the 100% current level is defined as:

ILED(max) = VIREF

/ (10 × RSL

) (3)

This feature provides direct analog dimming using a voltage from

0 to 1 V. This can be used to provide intensity-matching between

modules or groups of LEDs in critical display or backlighting

applications. It can also be used to provide a soft start, by con-

necting a capacitor from IREF to GND and a resistor from IREF

to VREG, or one-step dimming by use of a single logic control.

LED Brightness: PWM Dimming LED brightness can

be controlled by changing the current, which affects the light

intensity. However in some applications, for example with amber

LEDs, this will have some effect on the color of the LEDs.

In these cases it is more desirable to control the brightness by

switching the fixed LED current with a pulse width modulated

signal. This allows the LED brightness to be set with little effect

on the LED color and intensity and allows direct digital control

of the LED brightness.

A PWM signal can be applied to the EN input to enable PWM

dimming. The period of this signal should be less than the

minimum disable time, tDIS . During PWM dimming, the A6265

switches the LED current between 100% and typically 8% of

the full current. This ensures that the voltage change across the

LED string is limited to a few volts, depending on the number

of LEDs. This limits the stress on the load capacitor (across

the string of LEDs) due to large changes in voltage. If the load

capacitor is a multilayer ceramic type, then this will reduce any

audible noise due to the piezoelectric effect of the capacitor.

The rate of change of the LED current is also limited, to reduce

any large variations in the input current.

Sleep Mode If EN is held low for longer than the disable time,

tDIS , then the A6265 will shut down and put all sections into a

low-power sleep mode. In this mode the bias current is typically

less than 4 A. In the buck-boost configuration the only leakage

path remaining will be the path through the MOSFET.

Provided this is low, then the complete circuit may remain con-

nected to the power supply under all conditions. Note that the

disable time is derived from the oscillator period by a ratio of

32,768, so any variation in the oscillator frequency will change

the disable time.

Oscillator The main oscillator may be configured as a clock

source or it may be driven by an external clock signal. The oscil-

lator is designed to run between 100 and 700 kHz.

When the oscillator is configured as a clock source, the frequency

is controlled by a single external resistor, ROSC (k), between the

OSC pin and the GND pin. The oscillator frequency is approxi-

mately:

fOSC = 21700 / ROSC (kHz) (4)

Figure 2 shows the resulting fOSC for various values of ROSC.

If the OSC pin is connected to VREG or GND, the oscillator

frequency will be set internally to approximately 350 kHz.

When an external clock source is used to drive the OSC pin, it

can synchronize a number of A6265s operating together. This

ensures that only a single fundamental frequency is detectable

on the supply line, thus simplifying the design of any required

EMC filter. The disadvantage of using a single external clock

source is that all controllers will be switching current from the

supply at the same time. However, this effect may be reduced,

and the EMC performance may be further enhanced, by using

the CKOUT pin of another A6265 as the external clock source.

In this case the switching point of each subsequent A6265 in the

chain will be delayed from that of the previous A6265, and the

current pulses will be spread across the oscillator period.

700

600

500

400

300

200

100

30 50 70 11090 130 150 170 190 210

External Resistor Value, ROSC (k)

Oscillator Frequency, fOSC (kHz)

Figure 2. Internal oscillator frequency when set by ROSC

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

Diagnostics

The circuit includes several diagnostic and safety functions to

assist in ensuring safe operation of the LEDs, the A6265, and the

external components. When any fault is detected, one or both of

the fault flag outputs, FF1 and FF2, will be inactive (high imped-

ance, open drain) until the fault is removed. The action taken by

the A6265 when a fault occurs is defined in table 1. To be able to

monitor the state of FF1 and FF2, add a suitable external pull-up

resistor.

The A6265 will continue to drive the LEDs under most fault con-

ditions and will only disable the drive to the LEDs when a high

voltage hazard is present or the external components are likely to

be over-stressed. For output short circuits or open LED condi-

tions, the fault status is latched until EN is taken low or a power

cycle occurs. For output short circuits or a shorted LED string,

the fault status is latched until either EN is taken low for a period

greater than the disable time, or a power cycle occurs.

At start-up, a Fault Blank period, tFB , occurs before the fault

detection circuitry becomes active. This period allows steady

state conditions to be established before fault monitoring takes

place.

Note that no fault blanking is applied to open LED faults. This

is generally not an issue because the charging of the output filter

capacitor provides a degree of filtering. In addition, extremely

high voltages are prevented from causing potential device break-

down, for example in the external switching MOSFET.

VREG Undervoltage If the voltage at VREG, VREG , drops

below the specified turnoff voltage, VREGUV , the gate drive

output, SG, will be driven low and both fault flags, FF1 and

FF2, will be high impedance. VREG must rise above the turn-on

threshold, VREGUV + VREGUV

, before the output circuits are

activated. This ensures that the external FET is operating in its

fully enhanced state and avoids permanent damage to the FET,

caused by overheating.

LED Undercurrent Under some circuit conditions, particularly

during a low input voltage condition, it is possible that there

could be insufficient drive to maintain the current to the LEDs

at the required level. If the voltage across the LED current sense

resistor, RSS , falls below the target sense voltage, VIDL , by an

amount that is more than the LED undercurrent voltage differ-

ence, VUCL

, the A6265 will indicate an LED undercurrent condi-

tion by setting FF2 to high impedance. However, the A6265 will

continue to drive the output. When the output again reaches the

required current level, FF2 will go low.

Overtemperature Warning If the chip temperature exceeds

the overtemperature threshold, TJF , fault flag FF2 will be high

impedance. No action will be taken by the A6265 to limit the

chip temperature. An external control circuit must take action

to avoid permanent damage to the A6265 and/or the LEDs. The

temperature will continue to be monitored and the fault flags will

be deactivated when the temperature drops below the recovery

threshold provided by the hysteresis, TJhys .

LED Diagnostics The status of the LEDs in the load can be

determined by monitoring the voltage with respect to ground at

the three pins LP, LF, and LA, namely VLP , VLF , and VLA . These

voltages provide two differential voltage measurements:

• the voltage across a single reference LED:

LED = VLF – VLP (5)

• the ratio of the voltage across all LEDs in a single string:

VSTR = VLA – VLP (6)

The voltage, VSTR , is derived from the voltage across all LEDs in

the string, by an external resistor divider with a ratio equal to the

quantity of LEDs in the string. To minimize the effects of the bias

currents introducing an offset voltage, it is recommended that the

resistor between LP and LA should be approximately 560 .

So for example, if eight LEDs were used, the ratio required

would be an eighth, therefore the resistor connected between LA

and the anode end of the LED string would be 3.9 k;

560 / [560 + 3900] = 1/8 .

Table 1. Fault Table

Fault Pin Action Latched

FF1 FF2

No Fault L L No Action –

VREG Undervoltage Z Z Disable* No

Output Short Z L Disable* Yes

LED Undercurrent L Z No Action No

Overtemperature L Z No Action No

Open LED L Z Disable* Yes

Shorted LED L Z No Action No

Shorted LED String Z L Disable* Yes

* SG low, MOSFET off

L = active pull-down, Z = inactive, open drain

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

These measurements are used to determine if there is an open

circuit, if one or more LEDs are shorted, if the output is shorted,

or if there is a short across the LED string. Each condition is

described in turn in the following sections.

Open LED–An open circuit is evaluated when:

VSTR > VOCL (7)

where VOCL is the LED open circuit voltage defined in the Elec-

trical Characteristics table.

Because the output is current-controlled it is possible for an open

circuit on the output to cause extremely high voltages to be pres-

ent. Therefore, to prevent any hazardous voltages or damage to

the circuits, the gate drive output, SG, is immediately driven low

when an open circuit is detected. After an open circuit fault has

been detected, FF2 will become high impedance, and the open

circuit fault state will remain until the open fault time-out period

toto, expires. When the gate drive output is re-enabled at the end

of the open fault time-out period, the output is again monitored

for an open circuit. If the open circuit is still present, then the

fault will again be flagged and the switch drive disabled. This

cycle will continue, as long as the open circuit condition is pres-

ent.

Note that the Fault Blank timer is not used when an open LED

fault occurs. This is to avoid potentially damaging voltages

appearing in the power circuitry.

Shorted LED – A short circuit on one or more LEDs is detected

when:

• for the first (reference) LED:

VSTR > VLED + VSCOR (8)

• for other than the first (reference) LED:

VLED > VSTR + VSCO (9)

where VSTR and VLED are as defined above, VSCO is the nonrefer-

ence LED short offset voltage, and VSCOR is the reference LED

short offset voltage. VSCO and VSCOR are defined in the Electrical

Characteristics table.

When a short is present, the fault flag FF2 is high impedance, but

the regulator continues to operate and drives the remaining LEDs

with the correct regulated current. FF2 will remain high imped-

ance while the short circuit condition is present.

A short circuit on one or more LEDs will not cause a hazard

because the output is current-controlled. If one LED fails and

becomes a short circuit, then the remaining LEDs will continue

to be lit with the same current through, and voltage across,

each LED.

Note—Accuracy: The output status monitor relies on all the

LEDs in the load having a similar forward voltage drop. Where

possible all the LEDs forming the load for a single controller

should be taken from the same voltage bin. With only two or

three LEDs a wider variation in forward voltage is acceptable,

but the selection of LEDs from the same bin is more critical when

higher numbers of LEDs are used in a single string.

Shorted LED String or Output Short – A short circuit across

the LED string, is detected when:

VSTR < VSCL (10)

An output short can consist of the LP, LN, or LF terminals of

the LED string being shorted, either to the battery terminal or

to ground. Either a shorted LED string or output short will be

latched and will only be cleared by pulling EN low for a period

greater than the disable time, or by cycling the power.

If either a shorted LED string, or an output short is detected, the

A6265 will stop the switching action by pulling SG low. The

fault flag FF1 will go high impedance and should be pulled up to

the supply with suitable external pull-up resistors to indicate the

fault. Either a shorted LED string or output short will be latched

and will only be cleared by pulling EN low for a period greater

than the disable time, or by cycling the power.

The FF1 output can also be used with pull-up resistors and a

P-channel MOSFET in the supply, to isolate the switching ele-

ments and the load from the supply. This MOSFET should be

connected, as shown in figure 3, with the source connected to the

supply and the drain connected to the inductor of the converter.

To FF1

VBAT

To VIN

Figure 3. Example of a supply isolation MOSFET

Automotive High Current LED Controller

A6265

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115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

Two pull-up resistors are used to limit the voltage across the gate-

source junction during high input voltages or load dump condi-

tions. If the battery voltage is restricted, one resistor across the

gate-source junction can be used. The FF1 provides a sink current

of up to 1.3 mA.

This circuit can be used to avoid most hazardous conditions and

protect the circuit components from over-stress. Note that under

extreme cases, the circuit cannot protect against certain fault con-

ditions. For example, when any of the following occurs:

• In boost mode:

 If the cathode end of the LED string is shorted to VBAT, the

LED sense resistor effectively appears between VBAT and

ground. Depending on the current limit of the source supply or

the input fuse rating, the fault current may damage the resistor.

 If the LF node is shorted to VBAT, the reference LED and

the LED sense resistor effectively appear between VBAT and

ground. Depending on the current limit of the source supply or

the input fuse rating, the fault current may damage the resistor

and /or the reference LED.

 If the cathode end of the LED string is shorted to ground. A

fault current determined by the impedance of the shorting link

(now effectively the LED sense resistor) flows through the

power circuit. The fault current will either: be limited by the

maximum switch current sense, VIDS

, or if the source supply

cannot maintain this current, be limited by the source supply.

(The source supply will either: fold back, or if the current

exceeds the input fuse rating, the fuse will blow, creating an

open circuit).

• In buck-boost mode:

 If the cathode end of the LED string is shorted to ground, the

LED sense resistor effectively appears between VBAT and

ground. Depending on the current limit of the source supply or

the input fuse rating, the fault current may damage the resistor.

 If the LF node is shorted to ground, the reference LED and

the LED sense resistor effectively appear between VBAT and

ground. The reference LED will be reversed biased and will

probably be damaged.

 If the cathode end of the LED string (LP) is shorted to VBAT,

the impedance of the short appears in parallel with the series

combination of the series protection MOSFET and the LED

sense resistor. This will tend to reduce the effective impedance

of the LED sense resistor and correspondingly increase the

LED current.

To ensure the A6265 inputs (LP and LN) are not damaged dur-

ing any of the above faults, it is necessary to add a differential

resistor in series with the LN connection (between the sense

resistor and the LN pin). This resistor value should be approxi-

mately 150 .

If an output short is detected but it is necessary to keep the output

active, the FF1 output can be pulled low. This will override the

output disable but will not clear the fault.

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

Application Information

Component Selection

External component selection is critical to the successful appli-

cation of the LED driver. Although the inductor, the switching

MOSFET, and the output capacitor are the most critical elements,

the specification of the rectifying diode and sense resistors should

also be carefully considered.

The starting point for component selection is to define the maxi-

mum LED current, the voltage across the LEDs, and the input

operating voltage range. This then allows the average inductor

current under worst case conditions to be calculated. The induc-

tor value is then selected based on the acceptable inductor ripple

current. The amount of ripple current will then determine the

maximum inductor current under worst case conditions. From

this current the switch current sense resistor can be calculated.

LED Current Sense Resistor (RLS) If the voltage at the

IREF pin, VIREF

, is greater than 1V, or if IREF is tied to VREG,

then the value of the LED current sense resistor, RLS , can be

calculated from:

RLS = VIDL / ILED(max) (11)

where VIDL is the differential voltage across the LED current

sense amplifier and ILED(max) is the maximum LED current.

If VIREF is less than 1 V, then the value of the LED current sense

resistor can be calculated from:

RLS = VIREF / (10 × ILED(max) ) (12)

The typical value for VIDL is 100 mV. Examples of various sense

resistor values are given in table 2.

In boost mode, the power loss in the current sense resistor is

worse at the lowest input voltage:

PLOSS = (VLED / VIN(min)

) × RLS × I 2

LED (13)

In buck-boost mode, the power loss in the current sense resistor is

worse at the lowest input voltage:

LOSS = ( [ VIN + VLED

] / VIN ) × RLS × I 2

LED (14)

The resistors should be of a low inductance construction. Surface

mount chip resistors are usually the most suitable, however, axial

or radial leaded resistors can be used provided that the lead length

is kept to a minimum.

Inductor Selection Selecting the correct inductance is a

balance between choosing a value that is small enough to help

reduce size and cost, but high enough to ensure that the inductor

current ripple is kept to an acceptable level. A reasonable target

for the ripple current is 20% of the maximum average current.

The inductor current equations differ slightly depending on

whether the A6265 is configured as a boost or as a buck-boost

converter.

• In a boost converter configuration:

 The maximum average inductor current is approximately:

IL(av)(max) = ILED(max) × VLED / VIN(min) (15)

 The inductor current ripple is approximately:

ILRIP = VIN × (VLED – VIN

) / (fOSC × L × VLED

) (16)

 The inductor value is therefore:

L = VIN × ( VLED –VIN

) / (fOSC × ILRIP × VLED ) (17)

• In a buck-boost configuration:

 The maximum average inductor current is approximately:

IL(av)(max) = ILED(max) × ( VIN(min) + VLED) / VIN(min) (18)

 The inductor current ripple is approximately:

ILRIP = VIN × VLED / (fOSC × L × [VIN + VLED

] ) (19)

 The inductor value is therefore:

L = VIN × VLED / (fOSC × ILRIP × [ VIN

+ VLED ] ) (20)

where:

VLED is the voltage across the LED string,

VIN is the supply voltage,

VIN(min) is the minimum supply voltage,

L is the inductor value, and

fOSC is the oscillator frequency.

With an internal oscillator frequency of 350 kHz, the value of

the inductor for most cases will be between 20 and 50 H. The

maximum inductor current can then be calculated as:

IL(PK) = IL(av)(max) + (IRIP / 2) (21)

Table 2. Sense Resistor Values

ILED(max)

(mA)

RLS

(mΩ)

350 286

700 143

1000 100

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

This defines the minimum peak switch current as set by the

switch current sense resistor.

The current rating for the inductor should be greater, by some

margin, than the peak value above, IL(PK)

. When selecting an

inductor from manufacturers datasheets, there are two current

levels usually defined, the smallest value being the figure to work

with:

• Saturation level, where the inductance value typically drops by

10%, or

• Temperature rise, where the part experiences a certain rise in

temperature at full rated current. This parameter can be defined

between a 20°C and 50°C rise in temperature. It is important to

understand how manufacturers define the maximum operating

temperature, because this can often incorporate the self-heating

temperature rise.

In most cases the limiting current is usually the saturation value.

To improve efficiency, the inductor should also have low winding

resistance, typically < 50 m, and the core material will usually

be ferrite, with low losses at the oscillator frequency.

Recommended inductor manufacturers/series are:

• Coilcraft/ MSS1278T

• TDK/ SLF12575 type H

Diode The diode should have a low forward voltage, to reduce

conduction losses, and a low capacitance, to reduce switching

losses. Schottky diodes can provide both these features if care-

fully selected. The forward voltage drop is a natural advantage

for Schottky diodes and reduces as the current rating increases.

However, as the current rating increases, the diode capacitance

also increases so the optimum selection is usually the lowest cur-

rent rating above the required maximum, in this case IL(PK).

Switch Current Sense Resistor (RSS) Neither the absolute

value of the switch current nor the accuracy of the measurement

is important, because the regulator will continuously adjust the

switch current, within a closed loop, to provide sufficient energy

for the output. For maximum accuracy the switch sense resistor

value should be chosen to maximize the differential signal seen

by the sense amplifier. The input limit of the sense amplifier,

VIDS

, and the maximum switch current, IS(max), therefore define

the maximum value of the sense resistor as:

RSS = VIDS / IS(max) (22)

Where IS(max) is the maximum switch current and should be set

above the maximum inductor current, IL(PK) .

This represents the maximum measurable value of the switch

(and inductor) current; however, the peak switch current will

always be less than this, set by the control circuit, depending on

the input voltage and the required load conditions. Because the

switch current control is within a closed loop, it is possible to

reduce the value of the sense resistor to reduce its power dissipa-

tion. However this will reduce the accuracy of the regulated LED

current.

In Boost mode, the power loss in the switch sense resistor is

worse at the lowest input voltage:

PLOSS = (VLED [VLED – VIN(min)] / VIN(min)2) × RSS × I 2

LED (23)

In Buck Boost mode, the power loss in the switch sense resistor

is worse at the lowest input voltage:

PLOSS = (VLED / VIN(min))(VLED + VIN(min)) × RSS × I 2

LED (24)

External Switch MOSFET A logic-level N-channel MOSFET

is used as the switch for the DC-to-DC converter. In the boost

configuration the voltage at the drain of the MOSFET is equal

to the maximum voltage across the string of LEDs. In the

buck-boost configuration the output voltage is referenced to

the positive supply. This means that the voltage at the drain of

the MOSFET will reach a voltage equal to the sum of the LED

voltage and the supply voltage. Under load dump conditions,

up to 90 V may be present on this node. In this case the external

MOSFET should therefore be rated at greater than 100 V.

The peak switch current is defined by the maximum inductor cur-

rent, IL(PK) . However in most cases the MOSFET will be chosen

by selecting low on-resistance, which usually results in a current

rating of several times the required peak current.

In addition to minimizing cost, the choice of MOSFET should

consider both the on-resistance and the total gate charge. The

total gate charge will determine the average current required from

the internal regulator and thus the power dissipation.

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

Output Capacitor There are several points to consider when

selecting the output capacitor.

Unlike some switch-mode regulators, the value of the output

capacitor in this case is not critical for output stability. The

capacitor value is only limited by the required maximum ripple

voltage.

Due to the switching topology used, the ripple current for this

circuit is high because the output capacitor provides the LED

current when the switch is active. The capacitor is then recharged

each time the inductor passes energy to the output. The ripple

current on the output capacitor will be equal to the peak inductor

current.

Normally this large ripple current, in conjunction with the

requirement for a larger capacitance value for stability, would

dictate the use of large electrolytic capacitors. However in this

case stability is not a consideration, and the capacitor value can

be low, allowing the use of ceramic capacitors.

To minimize self-heating effects and voltage ripple, the equiva-

lent series resistance (ESR), and the equivalent series inductance

(ESL) should be kept as low as possible. This can be achieved by

multilayer ceramic chip (MLCC) capacitors. To reduce perfor-

mance variation over temperature, low drift types such as X7R

and X5R should be used.

The value of the output capacitor will typically be about 10 F

and it should be rated above the maximum voltage defined by the

series output LEDs.

Reverse Supply Protection Protection for the A6265 is

provided by an external low current diode between the supply

and the VIN pin, as shown in the Functional Block Diagrams

section. The isolation MOSFET shown in figure 3 is only able to

provide isolation when the supply polarity is correct. However,

with an additional P-channel MOSFET, it is also possible to

provide reverse battery protection to the switching elements and

the LEDs. The additional FET should be connected, as shown in

figure 4, with the drain to the supply and the source to the source

connection of the original isolation MOSFET.

In the complete circuit, consideration should be given to limiting

the maximum gate-source voltage of the FET. If the supply volt-

age is likely to exceed 20 V, then either: a Zener clamp must be

added in parallel with the gate-source resistor to prevent damage

to the FET, or a second resistor added as shown in figure 3.

To FF1

VBAT

To VIN

Figure 4. Example of a supply isolation MOSFET

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

Package LP 16-Pin TSSOP with Exposed Thermal Pad

1.20 MAX

0.15

0.00

0.30

0.19

0.20

0.09

8º

0º

0.60 ±0.15

1.00 REF

SEATING

PLANE

C0.10

16X

0.65 BSC

0.25 BSC

5.00±0.10

4.40±0.10 6.40±0.20

GAUGE PLANE

SEATING PLANE

ATerminal #1 mark area

For Reference Only; not for tooling use (reference MO-153 ABT)

Dimensions in millimeters

Dimensions exclusive of mold flash, gate burrs, and dambar protrusions

Exact case and lead configuration at supplier discretion within limits shown

Exposed thermal pad (bottom surface); dimensions may vary with device

6.10

0.65

0.45

1.70

3.00

Reference land pattern layout (reference IPC7351

SOP65P640X110-17M);

All pads a minimum of 0.20 mm from all adjacent pads; adjust as

necessary to meet application process requirements and PCB layout

tolerances; when mounting on a multilayer PCB, thermal vias at the

exposed thermal pad land can improve thermal dissipation (reference

EIA/JEDEC Standard JESD51-5)

PCB Layout Reference View

Branded Face

3 NOM

Automotive High Current LED Controller

A6265

Allegro MicroSystems, Inc.

115 Northeast Cutoff

Worcester, Massachusetts 01615-0036 U.S.A.

1.508.853.5000; www.allegromicro.com

For the latest version of this document, visit our website:

www.allegromicro.com

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information being relied upon is current.

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failure of that life support device or system, or to affect the safety or effectiveness of that device or system.

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Revision History

Revision Revision Date Description of Revision

Rev. 5 July 26, 2012 Update application information